CN115334716B - LED drive controller and system - Google Patents
LED drive controller and system Download PDFInfo
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- CN115334716B CN115334716B CN202211082807.5A CN202211082807A CN115334716B CN 115334716 B CN115334716 B CN 115334716B CN 202211082807 A CN202211082807 A CN 202211082807A CN 115334716 B CN115334716 B CN 115334716B
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- H—ELECTRICITY
- H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
- H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
- H05B45/00—Circuit arrangements for operating light-emitting diodes [LED]
- H05B45/30—Driver circuits
- H05B45/345—Current stabilisation; Maintaining constant current
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- H—ELECTRICITY
- H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
- H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
- H05B45/00—Circuit arrangements for operating light-emitting diodes [LED]
- H05B45/10—Controlling the intensity of the light
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- H—ELECTRICITY
- H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
- H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
- H05B45/00—Circuit arrangements for operating light-emitting diodes [LED]
- H05B45/50—Circuit arrangements for operating light-emitting diodes [LED] responsive to malfunctions or undesirable behaviour of LEDs; responsive to LED life; Protective circuits
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Abstract
The application discloses an LED driving controller and a system, wherein the LED driving controller comprises a zero-crossing detection circuit, a reference generation circuit, a constant current control circuit and a driving circuit, wherein the zero-crossing detection circuit is used for detecting inductance current in a driving system to generate a mode control signal and a zero-crossing detection signal, the reference generation circuit is used for generating peak reference voltage and average reference voltage according to a dimming signal, the constant current control circuit is used for linearly compensating the current sampling signal according to the mode control signal and the driving signal to generate compensation voltage and generating a conduction control signal according to the compensation voltage, the zero-crossing detection signal, the mode control signal, the driving signal and the average reference voltage, and the driving circuit is used for generating the driving signal according to the conduction control signal and a turn-off control signal and is used for controlling a power switch. The dimming is performed in a hysteresis-like control mode, a state that no current exists in the inductor for a long time is avoided, the output current can be kept constant, and the better screen flicker coefficient and current ripple rate are achieved.
Description
Technical Field
The invention relates to the technical field of LED driving, in particular to an LED driving controller and an LED driving system.
Background
The LED (LIGHT EMITTING Diode) has the characteristics of energy conservation, environmental protection, high efficiency, long service life and the like, and is widely applied to the fields of illumination, display and the like. In order to obtain better display visual effect, the accuracy requirement on LED dimming and toning is also higher and higher. Since the forward voltage drop of the LED is affected by the discrete process, the constant voltage driving method may cause the difference in brightness of the LED, so that the constant current driving method is generally used to drive the LED, and the brightness of the LED is determined by the average current passing through the LED.
In the current phase-cut dimming mode, a PWM1 signal with lower frequency (200 Hz-2 kHz) is input into a dimming control pin, and a PWM2 signal with higher frequency (200 kHz) is used for controlling a power tube switch to realize constant current. When PWM1 is low, the internal drive is closed, the inductance current is reduced, when PWM1 is high, the internal PWM2 signal controls the drive switch to work, therefore, when the dimming operation is carried out and the brightness of the LED is weakened, when the duty ratio of PWM1 is gradually reduced, the time for which the PWM1 signal is low is prolonged, the inductance current is reduced to 0, the maintenance time is longer and longer, the output current is reduced slowly, and larger output ripple is generated, and the screen flash coefficient exceeds the standard.
As the demand for health and intelligence in the lighting market continues to increase, intelligent lighting has become a future development. Generally, a stable constant current is provided to achieve the required luminous intensity and dimming precision of the LED, so it is very important to provide an LED driving controller capable of controlling the main circuit to output a stable current for the development of the LED.
Disclosure of Invention
In view of the above problems, an object of the present invention is to provide an LED driving controller and an LED driving system, so as to meet the requirements of a screen flicker coefficient, and achieve the purpose of deep dimming and color mixing.
According to an aspect of the present invention, there is provided an LED driving controller including:
The zero-crossing detection circuit is used for detecting the inductance current in the driving system to generate a mode control signal and a zero-crossing detection signal;
A reference generation circuit for generating a peak reference voltage and an average reference voltage according to the dimming signal;
the constant current control circuit is used for carrying out linear compensation on the current sampling signal according to the mode control signal and the driving signal to generate a compensation voltage, and generating a conduction control signal according to the compensation voltage, the zero-crossing detection signal, the mode control signal, the driving signal and the average value reference voltage;
and the driving circuit generates a driving signal according to the on control signal and the off control signal, wherein the driving signal is used for controlling the power switch.
Optionally, the constant current control circuit further generates the turn-off control signal based on the compensation voltage and the peak reference voltage.
Optionally, the current sampling signal characterizes a current flowing through a power switch in the drive system.
Optionally, when the mode control signal is at an active level, the inductor in the driving system is in an intermittent mode, and when the mode control signal is at an inactive level, the inductor in the driving system is in a continuous mode.
Optionally, the constant current control circuit includes:
And the compensation module is used for providing different compensation amounts according to the working mode of the inductor in the driving system and generating the compensation voltage according to the current sampling signal and the corresponding compensation amount.
Optionally, the compensation module provides a smaller amount of compensation when the inductance is in the continuous mode than when the inductance is in the discontinuous mode.
Optionally, the compensation module is configured to obtain a first current sampling signal and a second current sampling signal of the current sampling signal at different moments according to the driving signal, and generate a compensation current according to the first current sampling signal and the second current sampling signal.
Optionally, the compensation current is applied to resistors having different effective resistance values according to an operation mode of an inductor in the driving system to obtain different compensation amounts, and the compensation amounts are superimposed with the current sampling signal to generate the compensation voltage.
Optionally, the compensation module includes:
a logic processor that receives the drive signal and generates a first pulse signal after a first time delay of a rising edge of the drive signal and a second pulse signal after a second time delay of a falling edge of the first pulse signal, wherein the second pulse signal is generated before the falling edge of the drive signal arrives;
The first sampling and holding unit is connected with the logic processor and is used for sampling and holding the current sampling signal in the pulse time of the first pulse signal so as to generate the first current sampling signal;
The second sampling and holding unit is connected with the logic processor and is used for sampling and holding the current sampling signal in the pulse time of the second pulse signal so as to generate a second current sampling signal;
the first transconductance operational amplifier has a first input end receiving the first current sampling signal, a second input end receiving the second current sampling signal, and an output end providing the compensation current;
The first compensation resistor is connected with the output end of the first transconductance operational amplifier;
a second compensation resistor having a first end receiving the current sampling signal, and
And the first end of the first switch is connected with the second end of the second compensation resistor, the second end of the first switch is connected with the second end of the first compensation resistor, and the control end receives the mode control signal.
Optionally, the constant current control circuit further includes a conduction control module, including:
A reference voltage generation unit that provides a reference voltage;
An analog unit selecting one of the compensation voltage, the ground voltage, and the reference voltage as an analog voltage at different periods of the entire control period according to level states of the driving signal and the zero-crossing detection signal, and
And the conduction control unit generates the conduction control signal according to the analog voltage, the average value reference voltage and the driving signal.
Optionally, the analog unit includes:
the first end of the second switch receives the compensation voltage, and the control end receives the driving signal;
the first end of the third switch is grounded, and the control end receives the zero-crossing detection signal;
a NAND gate, a first end receives the driving signal and a second end receives the zero crossing detection signal;
a fourth switch, the first end receives the reference voltage, the control end is connected with the output end of the NAND gate,
The second end of the second switch is connected with the second end of the third switch and the second end of the fourth switch to output the analog voltage.
Optionally, the reference voltage generating unit is configured to set the reference voltage to the average value reference voltage when the inductor is in a continuous mode, and set the reference voltage to one half of the peak voltage of the compensation voltage when the inductor is in an intermittent mode.
Optionally, the reference voltage generating unit includes:
A third sample-hold unit that samples and holds a peak value of the compensation voltage and outputs the peak value voltage;
the voltage processing unit is connected with the third sample and hold unit and generates half of peak voltage;
A fifth switch, the first end of which is connected with the voltage processing unit, and the control end of which receives a mode control signal;
A sixth switch having a first end receiving the average reference voltage, a control end receiving a mode control signal, a second end connected to the second end of the fifth switch and outputting the reference voltage, and
A first capacitor having a first end connected to the second end of the sixth switch and the second end of the fifth switch, a second end grounded,
The fifth switch is turned on when the mode control signal is in a first level state, and the sixth switch is turned on when the mode control signal is in a second level state.
Optionally, the on control unit includes:
A second transconductance operational amplifier, the first end receiving the average reference voltage and the second end receiving the analog voltage;
the first end of the second capacitor is connected with the output end of the second transconductance operational amplifier, and the second end of the second capacitor is grounded;
The first end of the third transconductance operational amplifier is connected with the output end of the second transconductance operational amplifier, and the second end of the third transconductance operational amplifier is grounded;
the first end of the third capacitor is connected with the output end of the third transconductance operational amplifier, and the second end of the third capacitor is grounded;
A seventh switch, the first end of which is connected with the first end of the third capacitor, the second end of which is connected with the second end of the third capacitor, and the control end of which receives the driving signal;
and the first end of the first comparator is connected with the output end of the third transconductance operational amplifier, the second end of the first comparator receives a set threshold voltage, and the output end of the first comparator outputs the conduction control signal.
Optionally, the reference voltage generating unit obtains an intermediate voltage according to the average value reference voltage, and generates the reference voltage according to the average value reference voltage and the intermediate voltage.
Optionally, the reference voltage generating unit includes:
a scaling unit for scaling the average value reference voltage;
A subtracting unit connected with the scaling unit for subtracting the scaled average reference voltage from the intermediate reference voltage;
A clamping unit connected to the subtracting unit and clamping the subtraction result;
and the adding unit is respectively connected with the clamping unit and the average value reference voltage to generate the reference voltage.
Optionally, the constant current control circuit further includes a shutdown control module, including:
A front edge blanking unit for receiving the compensation voltage and outputting after a third time, and
And the first end of the second comparator is connected with the leading edge blanking unit, the second end of the second comparator receives the peak reference voltage, and the output end of the second comparator outputs the turn-off control signal.
Optionally, the peak reference voltage is in direct proportion to the duty ratio of the dimming signal, and the duty ratio of the dimming signal is lower than the slope between the peak reference voltage and the duty ratio of the dimming signal when the duty ratio of the dimming signal is lower than a set value, and higher than the slope between the peak reference voltage and the duty ratio of the dimming signal when the duty ratio of the dimming signal is higher than the set value.
Optionally, the method further comprises:
and the protection circuit is used for detecting overcurrent and/or overtemperature and outputting a protection signal to the driving circuit, and the driving circuit generates the driving signal according to the protection signal so as to control the power switch of the driving system to be turned off.
According to a second aspect of the present invention, there is provided an LED driving system comprising:
A diode having a first end receiving an input voltage and a second end connected to the first node;
The first end of the inductor is connected with the first node, and a load is connected between the second end of the inductor and the first end of the diode;
a power switch with a first end connected to the first node and a second end grounded, a control end receiving a driving signal and being turned on to charge the inductor and turned off to discharge the inductor in response to the driving signal, and
The LED drive controller as described above.
Optionally, the current sampling signal is connected to the second terminal of the power switch via a first resistor.
Optionally, the detection voltage is connected to the first end of the inductor via a fourth capacitor.
According to the LED driving controller and the LED driving system, the constant current control circuit in the LED driving controller is used for obtaining the compensation voltage through linear compensation of the current sampling signal of the main circuit, generating the on control signal according to the compensation voltage, the zero-crossing detection signal, the mode control signal, the driving signal and the average value reference voltage, and generating the off control signal based on the compensation voltage and the peak value reference voltage. And the driving circuit controls the power switch in the dimming process based on the driving signals generated by the turn-off control signals and the turn-on control signals, so that a long-time inductance no-current state can not occur, the current ripple rate is reduced, and a relatively excellent screen flicker coefficient is achieved.
Further, the compensation module in the constant current control circuit obtains the first current sampling signal and the second current sampling signal by sampling the current sampling signals at different moments, and generates corresponding compensation current based on the first current sampling signal and the second current sampling signal, so as to obtain corresponding compensation quantity compensation current sampling signals, keep the output current stable, and improve the load and the linear adjustment rate.
Furthermore, the reference voltage generating unit and the analog unit in the constant current control circuit output analog voltage to the conduction control unit, wherein the analog voltage represents inductance current in a continuous mode and an intermittent mode, and is equivalent to current sampling in the full period of on and off of the power switch, so that constant current output in the continuous mode and the intermittent mode is realized, and deep dimming under the condition of meeting a screen flash coefficient can be realized.
Further, the peak reference voltage generated by the reference generating circuit has inflection points when the dimming brightness is low (for example, when the duty ratio of the dimming signal is lower than a set value), so that the working frequency of the LED driving system is prevented from being too low to enter audio frequency, and audio noise is prevented from occurring.
Drawings
The above and other objects, features and advantages of the present invention will become more apparent from the following description of embodiments of the present invention with reference to the accompanying drawings, in which:
FIG. 1 shows a schematic diagram of an LED drive system provided in accordance with an embodiment of the present invention;
fig. 2 shows a schematic structural diagram of an LED driving controller according to an embodiment of the present invention;
fig. 3 shows a schematic structural diagram of a constant current control circuit in an LED driving controller according to an embodiment of the present invention;
Fig. 4 shows a schematic circuit diagram of a constant current control circuit in an LED driving controller according to an embodiment of the present invention;
fig. 5 shows a timing diagram of a constant current control circuit in an LED driving controller according to an embodiment of the present invention;
Fig. 6 shows a schematic diagram of a dimming reference curve of a reference generating circuit in an LED driving controller according to an embodiment of the present invention;
fig. 7 is a schematic diagram showing another structure of a reference voltage generating unit in a constant current control circuit of an LED driving controller according to an embodiment of the present invention;
fig. 8 shows a waveform schematic diagram of the reference voltage generating unit in fig. 7.
Detailed Description
Various embodiments of the present invention will be described in more detail below with reference to the accompanying drawings. The same reference numbers will be used throughout the drawings to refer to the same or like parts. For clarity, the various features of the drawings are not drawn to scale.
The current sampling of the traditional step-down constant current driving system can be placed at the low end, so that the sampling mode is simple and accurate, and the efficiency is higher than that of the high-end sampling. The common mode voltage of low-end current detection is low, a low-cost common operational amplifier can be used, but the detection resistor can introduce low-level interference, and the larger the sampling circuit is, the more obvious the ground potential interference is, and even the load can be influenced. And the phase-cut dimming mode matched with dimming cannot meet the requirements of the screen flicker coefficient.
Fig. 1 shows a schematic diagram of an LED driving system provided according to an embodiment of the present invention. Fig. 2 shows a schematic structural diagram of an LED driving controller according to an embodiment of the present invention. Fig. 3 shows a schematic structural diagram of a constant current control circuit in an LED driving controller according to an embodiment of the present invention. Fig. 4 shows a schematic circuit diagram of a constant current control circuit in an LED driving controller according to an embodiment of the present invention. Fig. 5 shows a timing diagram of a constant current control circuit in an LED driving controller according to an embodiment of the present invention.
As shown in fig. 1, the LED driving system 1000 includes a main circuit 1100 and an LED driving controller 1200.
The main circuit 1100 is, for example, a BUCK topology, and includes a diode D1, an inductor Lm, and a power switch Q1. A first terminal (e.g., cathode) of the diode D1 receives the input voltage VIN, and a second terminal (e.g., anode) of the diode D1 is connected to the first node D. A first terminal of the inductor Lm is connected to the first node D, and a load is connected between a second terminal of the inductor Lm and the first terminal of the diode D1. The first end of the power switch Q1 is connected to the first node d, the second end of the power switch Q1 is connected to the ground, and the control end of the power switch Q1 receives a driving signal and is turned on to charge the inductor Lm and turned off to discharge the inductor Lm in response to the driving signal.
Further, the main circuit 1100 further includes an output capacitor Cout, a resistor R3, and an input module 1110. The output capacitor Cout is connected between the first terminal of the diode D1 and the second terminal of the inductance Lm and is connected in parallel with the load. Wherein the load is for example a plurality of light emitting diodes LEDs connected in series. Resistor R3 is connected between the second terminal of power switch Q1 and ground. The input module 1110 includes a DC power DC, a positive terminal of which outputs an input voltage VIN, and an input filter capacitor Cin. The input filter capacitor Cin is connected in parallel with the direct current power supply DC.
LED drive controller 1200 includes zero crossing detect pin zcd, drive pin dr, sample pin cs, ground gnd, power pin vdd, temperature detect pin otp, dim pin dim, input pin vin. The LED driving system 1000 also includes some external components. Specifically, the zero crossing detection pin zcd of the LED drive controller 1200 is connected to the first node d in the main circuit 1100, for example, via a capacitor C4, to receive a detection voltage, which is used to characterize the inductance Lm current. The driving pin dr of the LED driving controller 1200 is connected to the control terminal of the power switch Q1, for example, via a resistor R2, to transmit a driving signal. The sampling pin cs of the LED driving controller 1200 is connected to the second terminal of the power switch Q1, for example, via a resistor R1, to obtain a current sampling signal. The power supply pin vdd of the LED drive controller 1200 is grounded, for example, via a capacitor C5. The temperature detection pin otp of the LED drive controller 1200 is grounded, for example, via a resistor R4. The dimming pin DIM of the LED driving controller 1200 receives the dimming signal DIM, for example, via a filter network comprising a resistor Rdim and a capacitor Cdim, a first terminal of the resistor Rdim receiving the dimming signal DIM, and a second terminal of the resistor Rdim connected to the dimming pin DIM. The capacitor Cdim is connected between the second terminal of the resistor Rdim and ground. The ground gnd of the LED driving controller 1200 is connected to, for example, one end of the resistor R3 and grounded. The input pin VIN of the LED driving controller 1200 is connected to, for example, a positive terminal of a direct current source DC in the main circuit 110 to receive the input voltage VIN.
It should be noted that the above-mentioned LED driving system is only one example, and the implementation of the LED driving system in the present application is not limited thereto. The main circuit 1100 in the LED driving system also includes other prior art shutdown circuits.
As shown in fig. 2, the LED driving controller 1200 includes at least a constant current driving circuit 1210, a reference generating circuit 1220, and a zero-crossing detecting circuit 1250, wherein the constant current driving circuit 1210 includes a constant current control circuit 1230 and a driving circuit 1240.
The zero-crossing detection circuit 1250 is configured to perform zero-crossing detection on the detection voltage Vzcd of the main circuit to generate a mode control signal vg_dcm and a zero-crossing detection signal vg_zcd, where the detection voltage characterizes the current of the inductor in the main circuit 1100. Further, the mode control signal vg_dcm characterizes that the inductor works in the discontinuous mode or the continuous mode, when the mode control signal vg_dcm is at an active level, the inductor in the main circuit 1100 is in the discontinuous mode, and when the mode control signal vg_dcm is at an inactive level, the inductor in the main circuit 1100 is in the continuous mode. The zero-crossing detection signal vg_zcd characterizes whether the inductor current crosses zero. Further, a zero-crossing detection circuit 1250 is connected to the zero-crossing detection pin zcd to receive the detection voltage Vzcd.
The reference generating circuit 1220 generates a peak reference voltage vpk_ref and an average reference voltage vav_ref according to the dimming signal DIM. Further, the reference generating circuit 1220 is connected to the dimming pin DIM to receive the dimming signal DIM.
The driving circuit 1240 is configured to provide a driving signal DR to control the power switch Q1 of the main circuit 1100 to be turned on or off, so that the main circuit 1100 provides a constant output current to the load. Further, the driving circuit 1240 is connected to the driving pin DR to provide the driving signal DR.
The constant current control circuit 1230 is used for linearly compensating the current sampling signal Vcs of the main circuit 1100 to obtain a compensation voltage, and is connected to the reference generation circuit 1220, the zero-crossing detection circuit 1250, and the driving circuit 1240 to generate the turn-on control signal vg_s based on the compensation voltage, the mode control signal vg_dcm, the zero-crossing detection signal vg_zcd, the driving signal DR, and the average reference voltage vav_ref. Further, the constant current control circuit 1230 also generates the off control signal vg_r based on the compensation voltage, the peak reference voltage vpk_ref. Further, the constant current control circuit 1230 is coupled to the dimming pin dim to receive a current sampling signal Vcs, which characterizes the current flowing through the power switch Q1, reflecting the relationship between the input voltage output voltage and the inductance in the main circuit 1100. The driving circuit also generates a driving signal DR according to the on control signal vg_s and the off control signal vg_r.
Further, referring to fig. 3, the constant current control circuit 1230 includes a compensation module 1231 for providing different compensation amounts according to the operation mode of the inductor in the main circuit 1100, and generating the compensation voltage Vcsc according to the current sampling signal Vcs and the corresponding compensation amounts. To compensate the output current by sampling the variation information of the current, thereby ensuring excellent adjustment rate at wide voltage input and wide voltage output. Further, the compensation module 1231 provides a smaller amount of compensation when the inductor is in the continuous mode than when the inductor is in the discontinuous mode. Further, the compensation module 1231 is configured to obtain a first current sampling signal v_h1 and a second current sampling signal v_h2 of the current sampling signal Vcs at different time according to the driving signal DR, generate a compensation current according to the first current sampling signal v_h1 and the second current sampling signal v_h2, apply the compensation current to a resistor having different effective resistance values according to an operation mode of an inductor in the main circuit 1100 to obtain different compensation amounts, and superimpose the compensation amounts on the current sampling signal Vcs to generate the compensation voltage Vcsc.
Specifically, referring to fig. 4, the compensation module 1231 includes a logic processor 12311, a first sample-and-hold unit 12312, a second sample-and-hold unit 12313, a first transconductance operational amplifier Gm1, a first compensation resistor Rc1, a second compensation resistor Rc2, and a first switch K1. The logic processor 12311 receives the driving signal DR output from the driving circuit 1240 and generates a first pulse signal vg_sh1 after a rising edge of the driving signal DR delays for a first time, and generates a second pulse signal vg_sh2 after a falling edge of the first pulse signal vg_sh1 delays for a second time, wherein the second pulse signal vg_sh2 is generated before the falling edge of the driving signal DR arrives. The first sample-and-hold unit 12312 is connected to the logic processor 12311 to sample and hold the current sampling signal Vcs during the pulse time of the first pulse signal vg_sh1 to generate the first current sampling signal v_h1. The second sample-and-hold unit 12313 is connected to the logic processor 12311 to sample and hold the current sampling signal Vcs during the pulse time of the second pulse signal vg_sh2 to generate the second current sampling signal v_h2. The first transconductance amplifier Gm1 has a first input terminal receiving a first current sampling signal v_h1, a second input terminal receiving a second current sampling signal v_h2, and an output terminal providing a compensation current. The first compensation resistor Rc1 has a first terminal receiving the current sampling signal Vcs and a second terminal connected to the output terminal of the first transconductance amplifier Gm 1. The first terminal of the second compensation resistor Rc2 receives the current sampling signal Vcs. The first end of the first switch K1 is connected to the second end of the second compensation resistor Rc2, the second end is connected to the second end of the first compensation resistor Rc1, and the control end receives the mode control signal vg_dcm. The first switch K1 is, for example, a low-turn-on switch, and when the mode control signal vg_dcm is low (an inactive level, indicating that the inductor is operating in the continuous mode), the first switch K1 is turned on. That is, the first switch K1 is turned on in the continuous mode, so that the compensation amount in the continuous mode is smaller than that in the discontinuous mode. The compensation current gm1 (v_h1-v_h2) is generated by the first current sampling signal v_h1 and the second current sampling signal v_h2 at different moments in the sampling current sampling signal Vcs, so that the corresponding compensation amount is obtained, and the load and the linear adjustment rate can be improved.
Further, referring to fig. 3, the constant current control circuit 1230 further includes a turn-on control module 1236 including a reference voltage generating unit 1232, an analog unit 1233, and a turn-on control unit 1234. The reference voltage generating unit 1232 is used to provide the reference voltage Voff. The analog unit 1233 selects one of the compensation voltage Vcsc, the ground voltage, and the reference voltage Voff as the analog voltage VL at different periods of the entire control period according to the level states of the driving signal DR and the zero-crossing detection signal vg_zcd. The turn-on control unit 1234 generates the turn-on control signal vg_s according to the analog voltage VL, the average reference voltage vav_ref, and the driving signal DR.
Specifically, referring to fig. 4, the analog unit 1233 includes a second switch K2, a third switch K3, a fourth switch K4, and a nand gate U1. The first terminal of the second switch K2 receives the compensation voltage Vcsc, and the control terminal receives the driving signal DR. The first end of the third switch K3 is grounded, and the control end receives the zero-crossing detection signal vg_zcd. The first end of the nand gate U1 receives the driving signal DR, and the second end receives the zero crossing detection signal vg_zcd. The first end of the fourth switch K4 receives the reference voltage Voff, and the control end is connected to the output end of the nand gate U1. The second terminal of the second switch K2 is connected to the second terminal of the third switch K3 and the second terminal of the fourth switch K4 to output the analog voltage VL. further, fig. 4 shows a reference voltage generating unit 1232 in which the reference voltage Voff is the average value reference voltage vav_ref in the inductor in the continuous mode and the reference voltage Voff is one half of the peak voltage vpk_h of the compensation voltage in the discontinuous mode. Specifically, the reference voltage generating unit 1232 includes a third sample-and-hold unit 12321, a voltage processing unit 12322, a fifth switch K5, a sixth switch K6, and a capacitor C1. The third sample-and-hold unit 12321 is configured to sample and hold the peak value of the compensation voltage Vcsc and output the peak voltage vpk_h. The voltage processing unit 12322 is connected to the third sample-and-hold unit 12321 for generating one-half of the peak voltage. The first terminal of the fifth switch K5 is connected to the voltage processing unit 12322, and the control terminal receives the mode control signal vg_dcm. The sixth switch K6 has a first terminal receiving the average reference voltage vav_ref, a control terminal receiving the mode control signal vg_dcm, and a second terminal connected to the second terminal of the fifth switch K5 and outputting the reference voltage Voff. The first terminal of the first capacitor C1 is connected to the second terminal of the sixth switch K6 and the second terminal of the fifth switch K5, and the second terminal is grounded to output the reference voltage Voff. The fifth switch K5 is turned on when the mode control signal vg_dcm is in the first level state, and the sixth switch K6 is turned on when the mode control signal vg_dcm is in the second level state. For example, when the inductor is in the discontinuous mode, the fifth switch K5 is turned on, the reference voltage Voff has a value vpk_h 1/2, and is regulated and output through the capacitor C1. When the inductor is in the continuous mode, the fifth switch K6 is turned on, the reference voltage Voff is the average value of the reference voltage vav_ref, and the reference voltage is regulated and output through the capacitor C1. Further, for example, when the driving signal DR is at a high level, the second switch K2 is turned on, and the analog voltage VL is the compensation voltage Vcsc. When the driving signal DR is at a low level and the zero crossing detection signal vg_zcd is at a low level, the fourth switch K4 is turned on, further, when the mode control signal vg_dcm is at a low level (the inductor is in a continuous mode), the sixth switch K6 is turned on, the analog voltage VL is the average reference voltage vav_ref, and when the mode control signal vg_dcm is at a high level (the inductor is in a discontinuous mode), the fifth switch K5 is turned on, the analog voltage VL is vpk_h 1/2. When the driving signal DR is at a low level and the inductor current drops to zero in the intermittent mode, the zero-crossing detection signal vg_zcd is at a high level, and the third switch is turned on, the analog voltage VL is a ground voltage. Further, the on control unit 1234 includes a second transconductance operational amplifier Gm2, a third transconductance operational amplifier Gm3, a capacitor C3, a seventh switch K7, and a first comparator CMP1. The first terminal of the second transconductance operational amplifier Gm2 receives the average reference voltage vav_ref and the second terminal receives the analog voltage VL. The first terminal of the capacitor C2 is connected to the output terminal of the second transconductance operational amplifier Gm2 and provides the voltage Vcomp, and the second terminal is grounded. The first end of the third transconductance operational amplifier Gm3 is connected to the output end of the second transconductance operational amplifier Gm2, and the second end is grounded. The first end of the capacitor C3 is connected to the output end of the third transconductance operational amplifier Gm3 and provides the voltage Vramp, and the second end is grounded. the first end of the seventh switch K7 is connected to the first end of the capacitor C3, the second end is connected to the second end of the capacitor C3, and the control end receives the driving signal DR. The first end of the first comparator CMP1 is connected to the output end of the third transconductance operational amplifier Gm3, the second end receives the set threshold voltage vr_1, and the output end outputs the on control signal vg_s. Further, the on control unit 1234 is configured to calculate the off time of the corresponding power switch Q1 according to the analog voltage VL and the average reference voltage vav_ref of the whole period, so as to control the stabilization of the output current. Further, when the driving signal DR is at a low level, the current output by the third transconductance operational amplifier Gm3 charges the capacitor C3, and when the voltage Vramp on the capacitor C3 is greater than the set threshold voltage vr_1, the on control signal vg_s is generated.
Under deep dimming, the working frequency of the switching tube is reduced, the peak reference and the average reference are also reduced, the inductance current enters an intermittent mode, and if a traditional control mode is adopted, only the current signal of the switching tube in the conduction stage is sampled, so that the requirement of the stroboscopic coefficient cannot be met.
In order to meet the requirements of the screen flicker coefficient under deep dimming, the current signal of the power switch in the turn-off stage must also be sampled. The invention designs the analog unit to output the analog voltage VL, the analog voltage characterizes the inductance current in the continuous mode and the discontinuous mode, and the current sampling is equivalent to the current sampling in the full period of the on and off of the power switch. The conduction control unit outputs a conduction control signal according to the analog voltage, the average value reference voltage and the driving signal, so that the conduction and the disconnection of the power switch can be controlled, and constant current output in a continuous mode and an intermittent mode is realized.
Referring to fig. 3 and 4, the constant current control circuit 1230 further includes a turn-off control module 1235 for generating a turn-off control signal vg_r according to the compensation voltage Vcsc and the peak reference voltage vpk_ref. Further, the turn-off control module 1235 includes a leading edge blanking unit 12351 and a second comparator CMP2. The leading edge blanking unit 12351 receives the compensation voltage Vcsc and outputs it after a third time. The second comparator CMP2 has a first terminal connected to the leading edge blanking unit 12351, a second terminal receiving the peak reference voltage vpk_ref, and an output terminal outputting the off control signal vg_r. Specifically, for example, when the compensation voltage Vcsc is greater than the peak reference voltage vpk_ref after the third time, the off control signal vg_r is generated.
In other embodiments, the LED drive controller 1200 also includes a power supply circuit 1260 therein. The power supply circuit 1260 is connected to the input pin VIN and receives the input voltage VIN. The power supply circuit 1260 provides a power supply voltage to the power supply pin vdd and the drive circuit 1240 to power the LED drive controller 1200.
In other embodiments, a protection circuit 1270 is also included in the LED drive controller 1200. The protection circuit 1270 performs over-current detection and/or over-temperature detection, and outputs a protection signal vg_shutdown to the driving circuit 1240, and the driving circuit 1240 generates a driving signal DR according to the protection signal vg_shutdown to control the power switch Q1 of the main circuit 1100 to be turned off. Wherein the resistance value of the resistor R4 varies with temperature.
Further, fig. 5 is incorporated. At time t0, the driving signal DR changes from low to high, the power switch Q1 is turned on, the inductor current IL starts to rise slowly, and the voltage value of the current sampling signal Vcs rises gradually. After the first time, at time t1, the logic processor 12311 in the compensation module 1231 generates a first pulse signal vg_sh1 with a certain pulse width, and the first sample-and-hold unit 12312 samples and holds the current sampling signal Vcs at time t1 to generate a first current sampling signal v_h1. At time t2 when the falling edge of the first pulse signal vg_sh1 arrives, the logic processor 12311 in the compensation module 1231 generates a second pulse signal vg_sh2 with a certain pulse width, and the second sample-and-hold unit 12313 samples and holds the current sample signal Vcs at time t2 to generate a second current sample signal v_h2. At time t3, the driving signal DR is changed from high to low, the power switch Q1 is turned off, the inductance current IL is reduced, the voltage Vramp starts to charge from 0, and at time t4, the driving signal DR is changed from low to high, and the voltage Vramp is cleared after reaching the set threshold voltage Vr_1. In the period t0-t4, the zero crossing detection signal vg_zcd is always low, the mode control signal vg_dcm is always low, and the above is a complete period, describing the operation timing of the constant current control circuit 1230 when the inductor operates in the continuous mode. And between t4 and t5, the inductor still works in the continuous mode, the value of the analog voltage VL changes along with the states of the mode control signal vg_dcm, the zero crossing detection signal vg_zcd and the driving signal DR, and particularly, how to take the value can be described in relation to the analog unit 1233 in fig. 3.
Then at time t5, the driving signal DR is changed from low to high, the power switch Q1 is turned on, the inductance current IL starts to rise slowly from 0, the voltage value of the current sampling signal Vcs gradually rises, at time t6, the driving signal DR is changed from high to low, the power switch Q1 is turned off, the inductance current IL is lowered, the voltage Vramp starts to charge, at time t7, the inductance current IL is lowered to 0, the zero crossing detection signal Vg_ZCD signal is changed from low to high, at time t8, the driving signal DR is changed from low to high, the zero crossing detection signal Vg_ZCD signal is changed from high to low, and the voltage Vramp reaches the set threshold voltage Vr_1. That is, during t5-t8, the mode control signal vg_dcm is always high level, describing the operation timing of the constant current control circuit 1230 when the inductor is operated in the discontinuous mode.
Fig. 6 shows a schematic diagram of a dimming reference curve of a reference generating circuit in an LED driving controller according to an embodiment of the present invention.
As shown in fig. 6, the reference generating circuit 1220 generates the corresponding peak reference voltage vpk_ref and the average reference voltage vav_ref according to the dimming signal DIM of different duty ratios D. Further, the average reference voltage vav_ref increases in proportion to an increase in the duty ratio D. The peak reference voltage vpk_ref is proportional to the duty ratio D, and the duty ratio D of the dimming signal DIM is lower than the slope between the peak reference voltage vpk_ref and the duty ratio D of the dimming signal DIM when the duty ratio D of the dimming signal DIM is higher than the set value. That is, the peak reference voltage vpk_ref may be turned around when the dimming brightness is low (e.g., when the duty ratio D of the dimming signal DIM is lower than a set value), and the operating frequency of the LED driving system may be prevented from being too low to enter audio and thus audio noise may occur.
Fig. 7 is a schematic diagram showing another structure of the reference voltage generating unit in the constant current control circuit of the LED driving controller according to the embodiment of the present invention. Fig. 7 shows a waveform schematic diagram of the reference voltage generating unit in fig. 8.
In an alternative embodiment, as shown in fig. 7, another reference voltage generating unit 2232 in the constant current control circuit 1230 obtains an intermediate voltage according to the average value reference voltage vav_ref and generates the reference voltage v_3 according to the average value reference voltage vav_ref and the intermediate voltage. Specifically, the reference voltage generating unit 2232 includes a scaling unit 22321, a subtracting unit 22322, a clamping unit 22323, and an adding unit 22324. The scaling unit 22321 performs scaling processing on the average reference voltage vav_ref. The subtracting unit 22322 is connected to the scaling unit 22321, and performs a subtraction process on the scaled average reference voltage and the intermediate reference voltage vref_4. The clamping unit 22323 is connected to the subtracting unit 22322, and clamps the subtraction result. The adding unit 22324 is connected to the clamping unit 22323 and the average reference voltage vav_ref, respectively, to generate the reference voltage v_3.
Referring to fig. 8, when the duty ratio D of the dimming signal DIM is higher than the set value, the reference voltage v_3 provided by the reference voltage generating unit 2232 is the average value reference voltage vav_ref. When the duty ratio D of the dimming signal DIM is lower than the set value, the reference voltage v_3 provided by the reference voltage generating unit 2232 has a value between the average reference voltage vav_ref and the peak reference voltage vpk_ref.
According to the LED driving system comprising the LED driving controller, dimming is performed in a hysteresis-like control mode, a state that inductance is free of current for a long time is avoided, output current can be kept constant, and therefore excellent screen flashing coefficient and current ripple rate are achieved. Furthermore, the application realizes higher load and linear adjustment rate by compensating the current sampling signal and the compensation quantity thereof is related to the input and output voltage.
Furthermore, the reference voltage generating unit and the analog unit in the constant current control circuit output analog voltage to the conduction control unit, wherein the analog voltage represents inductance current in a continuous mode and an intermittent mode, and is equivalent to current sampling in the full period of on and off of the power switch, so that constant current output in the continuous mode and the intermittent mode is realized, and deep dimming under the condition of meeting a screen flash coefficient can be realized.
Furthermore, the application adjusts the slope between the peak reference voltage and the duty ratio under the dimming condition of low duty ratio, so that the working frequency of the LED driving system is still controlled above the audio frequency, and the audio noise is avoided.
Embodiments in accordance with the present invention, as described above, are not intended to be exhaustive or to limit the invention to the precise embodiments disclosed. Obviously, many modifications and variations are possible in light of the above teaching. The embodiments were chosen and described in order to best explain the principles of the invention and the practical application, to thereby enable others skilled in the art to best utilize the invention and various modifications as are suited to the particular use contemplated. The invention is limited only by the claims and the full scope and equivalents thereof.
Claims (21)
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| CN102273327A (en) * | 2008-12-10 | 2011-12-07 | 凌力尔特有限公司 | Dimming Controlled LEDs Using Flyback Converters with High Power Factor |
| CN103533703A (en) * | 2012-06-28 | 2014-01-22 | 三星电机株式会社 | Circuit and method for driving LED light |
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| US7276861B1 (en) * | 2004-09-21 | 2007-10-02 | Exclara, Inc. | System and method for driving LED |
| CN102076149B (en) * | 2010-11-15 | 2012-01-04 | 凹凸电子（武汉）有限公司 | Light source drive circuit, controller and method for controlling light source brightness |
| CN112512170B (en) * | 2020-11-05 | 2023-06-06 | 杭州士兰微电子股份有限公司 | LED control circuit, LED driving device and driving control method |
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| Publication number | Priority date | Publication date | Assignee | Title |
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| CN102273327A (en) * | 2008-12-10 | 2011-12-07 | 凌力尔特有限公司 | Dimming Controlled LEDs Using Flyback Converters with High Power Factor |
| CN103533703A (en) * | 2012-06-28 | 2014-01-22 | 三星电机株式会社 | Circuit and method for driving LED light |
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