CN118249827A

CN118249827A - Method for demodulating RF signals in the presence of in-band harmonic spurs

Info

Publication number: CN118249827A
Application number: CN202311761190.4A
Authority: CN
Inventors: D·哈那; F·斯特勒
Original assignee: STMicroelectronics Grand Ouest SAS
Current assignee: STMicroelectronics Grand Ouest SAS
Priority date: 2022-12-22
Filing date: 2023-12-20
Publication date: 2024-06-25

Abstract

The present disclosure relates to a method for demodulating a Radio Frequency (RF) signal, the method comprising the steps of: determining the nearest harmonic of the clock signal according to the center frequency (Frx) of the reception band; and IF the nearest harmonic is within the frequency band, setting an Intermediate Frequency (IF) of a Near Zero Intermediate Frequency (NZIF) receiver to a difference (SpurOffset) between the center frequency (Frx) and the nearest harmonic.

Description

Method for demodulating RF signals in the presence of in-band harmonic spurs

Cross Reference to Related Applications

The present application claims priority from french patent application EP 22306989 entitled "Method fordemodulating A RF SIGNAL IN THE PRESENCE of inband harmonic spurs (method of demodulating RF signals in the presence of in-band harmonic spurs)" filed on 12 months 22 of 2022, which is incorporated herein by reference to the maximum extent allowed by law.

Technical Field

The present disclosure relates generally to methods for demodulating Radio Frequency (RF) signals and RF circuitry for implementing such methods.

Background

RF receivers can be divided into two main categories: a Zero Intermediate Frequency (ZIF) architecture based receiver and a Near Zero Intermediate Frequency (NZIF) architecture based receiver.

The NZIF receiver converts the received radio signal to an intermediate frequency, the carrier frequency of which is in the order of the baseband signal bandwidth, but significantly lower than the radio carrier frequency to be demodulated. Digital clock circuits used in systems carrying NZIF receivers may generate harmonic spurs whose frequencies tend to fall within the frequency band of the RF received signal to be demodulated.

Disclosure of Invention

It is desirable to provide a method for demodulating an RF received signal that reduces the effects of clock harmonic spurs.

One or more embodiments address all or part of the shortcomings of known methods for demodulating RF signals.

One or more embodiments provide a method for demodulating an RF signal, the method comprising the steps of: determining the nearest harmonic wave of the clock signal according to the center frequency of the receiving frequency band; and if the nearest harmonic is within the frequency band, setting an intermediate frequency of a near zero intermediate frequency receiver to be a difference between the center frequency and the nearest harmonic.

One or more embodiments provide an RF signal demodulation circuit configured to: determining the nearest harmonic wave of the clock signal according to the center frequency of the receiving frequency band; and if the nearest harmonic is within the frequency band, setting an intermediate frequency of a near zero intermediate frequency receiver to be the difference between the center frequency and the nearest harmonic.

According to one embodiment, the intermediate frequency is set to a nominal NZIF value if the nearest harmonic is outside the frequency band.

According to one embodiment, determining the nearest harmonic comprises: a rounded value (rounding) of the ratio between the center frequency of the receive band and the frequency value of the clock signal is determined.

According to an embodiment, the nearest harmonic is equal to the product of the rounding value times the frequency value of the clock signal.

According to one embodiment, the nearest harmonic is within the frequency band if the absolute value of the difference between the center frequency and the nearest harmonic is less than half-bandwidth value of the RF signal, otherwise the nearest harmonic is outside the frequency band.

According to one embodiment, a method or circuit includes: the RF signal is amplified and the amplified signal is split into a first path and a second path.

According to one embodiment, a method or circuit includes: the amplified RF signal is mixed in a first path with an in-phase signal of a local oscillator frequency corresponding to the sum of the center frequency and the intermediate frequency, and the amplified RF signal is mixed in a second path with a quadrature signal of the local oscillator frequency.

According to one embodiment, a method or circuit includes: the high frequencies of the mixed signals of the first and second paths are filtered and the filtered signals are amplified.

According to one embodiment, a method or circuit includes: the amplified filtered signal is converted to a digital signal.

According to one embodiment, a method or circuit includes: the digitized signal is mixed with a third signal having an intermediate frequency IF.

According to one embodiment, a method or circuit includes: the high frequencies of the digitally mixed signal are filtered.

According to one embodiment, a method or circuit includes: a decimation operation is performed on the filtered and digitally mixed signal.

Drawings

The above features and advantages and other features and advantages will be described in detail in the following description of the particular embodiments, given by way of example and not limitation, with reference to the accompanying drawings, in which:

Fig. 1 is a graph of signal level as a function of frequency, illustrating an example of conventional operation of an RF receiver;

FIG. 2 illustrates an example of a communication channel subject to harmonic spurious interference at various frequency offsets;

FIG. 3 illustrates the conventional operation of NZIF demodulation and the resulting harmonic spurs in baseband;

fig. 4 illustrates an embodiment of an RF signal demodulation method;

fig. 5 illustrates an embodiment of an RF signal demodulation method;

FIG. 6 illustrates an embodiment of an RF circuit;

FIG. 7A illustrates frequency sensitivity measurements made by the method of FIG. 3; and

Fig. 7B illustrates frequency sensitivity measurements obtained by the methods of fig. 4 and 5.

Detailed Description

Like features have been designated by like reference numerals throughout the various drawings. In particular, structural and/or functional features common between the various embodiments may have the same reference numerals and may be provided with the same structure, dimensions, and material properties.

For clarity, only the operations and elements useful for understanding the embodiments described herein have been illustrated and described in detail.

Unless otherwise indicated, when referring to two elements being connected together, this means that there is no direct connection of any intermediate element other than a conductor, and when referring to two elements being coupled together, this means that the two elements may be connected or they may be coupled via one or more other elements.

In the following disclosure, unless otherwise indicated, when reference is made to absolute positional qualifiers such as the terms "front", "rear", "top", "bottom", "left", "right", etc., or relative positional qualifiers such as the terms "above", "below", "higher", "lower", etc., or orientation qualifiers such as "horizontal", "vertical", etc., reference is made to the orientation shown in the figures.

Unless otherwise indicated, the expressions "about", "approximately" and "on the order of …" mean within 10%, and preferably within 5%.

Fig. 1 is a graph of signal level (strength (a.u)) as a function of frequency (f (MHz)), illustrating a conventional operational example of an RF receiver.

In RF communication, frequencies are divided into frequency bands that are respectively allocated to different standards and/or operators. Fig. 1 shows two example frequency bands 120 for cellular communication, the two example frequency bands 120 extending from 758MHz to 803MHz (3 GPP band 28) and from 925MHz to 960MHz (3 GPP band 8), respectively. The frequency band is divided into channels (which are not illustrated in fig. 1) having a bandwidth typically comprised between 10kHz and 10 MHz.

The RF circuits use different frequencies, all based on the master clock 100 (master clock) present in the circuit. In practice, the clock signal is not perfect and harmonics 110 (the master clock harmonics) are generated. These harmonics are present as odd/even multiples of the frequency value Fref of the master clock 100. The intensity of the harmonic 110 may vary according to its level and decrease as the level increases. The frequency value Fref of the master clock is much lower (at least 10 orders of magnitude) than the frequency of the RF cellular band. Some of the harmonics will therefore fall within the cellular frequency band.

In the illustrated example, four harmonics fall within the frequency band formed from 758MHz to 803MHz and three harmonics fall within the frequency band formed from 930MHz to 950 MHz. These particular harmonic spurs will fall in channels that include their respective frequencies, which causes a hearing loss (deaf) channel in signal reception, i.e., a channel with lower performance sensitivity.

Fig. 2 illustrates an example of a communication channel subject to harmonic spurious interference at different frequency offsets. More precisely, the example of fig. 2 illustrates different channels of the 3GPP NBIOT standard (narrowband internet of things). The NBIOT standard defines adjacent channels each having a bandwidth of 200kHz and a center frequency. In fig. 2, four channels 210, 212, 214, 216 are shown.

Channel allocation in RF communications, particularly the NBIOT standard, may use raster shifting in the presence of cellular bands around the NBIOT channel. The example of fig. 2 shows two independent channels 202 and 204 (no raster offset, i.e. no surrounding cellular band), which independent channels 202 and 204 are aligned with a 200kHz raster of a frequency band centered at a frequency of 940.800MHz and with a 100kHz raster, respectively (center frequency 940.900 MHz). Fig. 2 also shows a channel 206 having a center frequency of 940.8075MHz and a channel 208 having a center frequency of 940.8925MHz, the channel 206 corresponding to a "guard band" or "in-band" mode in which the received signal is aligned on a 200kHz grating, the grating offset being +7.5kHz, the channel 208 corresponding to a "guard band" or "in-band" mode in which the received signal is aligned on a 100kHz grating, the grating offset being-7.5 kHz.

In the illustrated example, a clock signal generated in an RF receiver and having a frequency of 19.2MHz may generate a clock harmonic spur 110 of the order 49 at 940.800 MHz. The harmonic spur 110 will fall into the received signal channels of configurations 202, 206 and 208.

Fig. 3 illustrates the normal operation of NZIF demodulation and the harmonic spurs generated in baseband. More specifically, fig. 3 illustrates an NZIF demodulation method applied to a received NBIOT signal 200, the received NBIOT signal 200 having a center frequency Frx and having an exemplary harmonic spurious 110, the harmonic spurious 110 being present at frx+50kHz and falling into the signal channel.

In step a), the intermediate frequency IF is set to-480 kHz and a first analog de-rotation is applied to the received signal by mixing the received signal with a signal having a local oscillator frequency LO. The frequency LO is set equal to the intermediate frequency IF added to the center frequency Frx of the received signal. At the end of step a), the center frequency of the signal is shifted from the frequency LO by an intermediate frequency IF value, i.e., -480kHz.

In step B), the signal obtained at the end of step a) is digitally processed to perform another de-rotation at 480kHz, thereby down-converting the signal to baseband.

The resulting down-converted signal is represented in C). Clock harmonic spurs remain in the demodulated signal.

The demodulation method described in fig. 3 cannot efficiently handle the effects of harmonic spurs, since clock harmonic spurs at 50kHz still exist. To reduce the effects of harmonic spurs 110, additional methods may be implemented using digital signal processing techniques such as band pass filters or Fast Fourier Transforms (FFTs) or Indirect Fast Fourier Transforms (IFFTs), but such efforts may result in heavy computation and the signals may be affected.

Fig. 4 illustrates an embodiment of an RF signal demodulation method. More specifically, similar to the example of fig. 3, fig. 4 illustrates an embodiment of a demodulation method applied to the received NBIOT signal 200 and in the case of an exemplary harmonic spur 110, the exemplary harmonic spur 110 being present at frx+50kHz and falling into the signal channel.

In step a'), the intermediate frequency IF is set equal to the clock harmonic spurious 110 frequency, which falls in the corresponding signal channel, i.e. here 50kHz. In one or more embodiments, a first analog de-rotation is applied to the received signal, for example, by mixing the received signal with a signal having a local oscillator frequency LO. The frequency LO is set equal to the intermediate frequency IF added to the center frequency Frx of the received signal. In one or more embodiments, at the end of step a'), the center frequency of the mixed signal is shifted from the frequency LO by an intermediate frequency IF value, i.e., 50kHz.

In step B '), the signal obtained during step a') is subjected to a second digital de-rotation, for example by digitally processing the signal for another de-rotation of the value of the intermediate frequency IF, thereby down-converting the signal to baseband. In one or more embodiments, during the second de-rotation, since the intermediate frequency IF is set to the harmonic spurious 110 frequency value, the harmonic spurious is now concentrated in the middle of the frequency band and will therefore reduce its effect by filtering the down-processing of the signal.

In one or more embodiments, clock harmonic spurs 110 are almost completely suppressed in the resulting down-converted signal represented in C). In one or more embodiments, the method presented in fig. 4 improves the performance of a so-called quench channel that is affected by harmonic clock spurs emanating from the digital baseband.

Fig. 5 illustrates an embodiment of an RF signal demodulation method. More specifically, fig. 5 illustrates steps of an embodiment for determining the frequency LO to be used in the method of fig. 4.

In step 502 (RxIF algorithm entry), the algorithm to determine the frequency LO begins. In one or more embodiments, the algorithm runs, for example, prior to the demodulation method of fig. 4.

In step 504 (HarmRank =round (Frx/Fref)), a rounded value of the ratio between the center frequency Frx of the received signal and the frequency Fref of the clock signal is calculated. In one or more embodiments, the rounding value gives a level HarmRank of clock harmonics, which is closest to the center frequency Frx of the received signal. By way of example, in the case of signal 202 of FIG. 2 at 940.8MHz, the reference clock signal is at 19.2MHz, and the nearest harmonic would be level 49 (940.8/19.2). In one or more embodiments, the most recent harmonic (= HarmRank ×fref) of the clock signal is allowed to be determined from the center frequency Frx of the reception band.

In step 506 (SpurOffset =frx-HarmRank ×fref), the difference Frx-HarmRank ×fref is performed and the result SpurOffset is stored.

In step 508 (| SpurOffset | < half bandwidth. In the example of fig. 2, the half bandwidth value is 200 kHz/2=100 kHz. In one or more embodiments, step 510 (if= SpurOffset) is performed to fix the intermediate frequency at the SpurOffset value IF the absolute value of SpurOffset is less than half the bandwidth of the received signal (branch Y), i.e., IF the most recent clock harmonic spurious falls in the received signal channel. In one or more embodiments, step 512 (if=std NZIF value) is performed IF SpurOffset absolute is greater than half-bandwidth (branch N), i.e., IF the nearest clock harmonic spurs are outside the received signal channel. In other words, in one or more embodiments, the intermediate frequency IF is set to a value that is not dependent on harmonic spurious frequencies, such as the value used in the NZIF common method, such as ±480kHz.

In one or more embodiments, where the RX frequency is an integer multiple of the reference clock signal Fref, then step 510 is implemented with SpurOffset =if=0 Hz, which corresponds to a direct conversion.

In step 514 (lo=frx+if), the result of step 510 or step 512 (depending on the result of step 508) is implemented to calculate the local oscillator frequency lo=frx+if.

In one or more embodiments, the algorithm then exits in step 516 (exit RxIF selection). In an example, the method of fig. 4 may then be performed using the calculated lo=frx+if value.

In one or more embodiments, the algorithm of fig. 5 is performed, for example, whenever the center frequency of the received signal changes or in the case of a clock signal reference frequency change.

Fig. 6 illustrates an embodiment of an RF circuit 500. More specifically, in one or more embodiments, RF circuit 500 may be used to implement the method of fig. 4 and the method of fig. 5.

In one or more embodiments, RF circuit 600 includes, for example, a clock circuit 601 and a Near Zero Intermediate Frequency (NZIF) receiver 604, clock circuit 601 configured to generate a clock signal having a reference frequency Fref. Alternatively, in one or more embodiments, the clock circuit 601 is disposed outside of the RF circuit, or in an NZIF receiver.

In the illustrated example, the NZIF receiver comprises a first module RFE (RF front end) coupled or preferably connected to a second module BB (baseband). In one or more embodiments, the second module BB is coupled or preferably connected to a third module ADC (analog-to-digital converter), which is coupled or preferably connected to a fourth module DFE (digital front end).

In one or more embodiments, the first module RFE includes an amplifier 610, such as a low noise amplifier LNA (low noise amplifier), the amplifier 610 being configured to amplify the received RF signal Frx. In one or more embodiments, the amplified signal is then split into two different paths 611, 613. In one or more embodiments, the first module includes one mixer 612, 614 per path. In one or more embodiments, each of the mixers 612, 614 is configured to mix the received RF signal Frx of the corresponding path with a signal having a local oscillator frequency LO. In the disclosed embodiment, the frequency LO corresponds to the sum of the center frequency Frx of the received signal and the intermediate frequency IF determined, for example, based on the algorithm of fig. 5. In one or more embodiments, the local oscillator frequency LO of path 611 corresponds to the in-phase signal LO-I and the local oscillator frequency LO of path 613 corresponds to the quadrature signal LO-Q. In one or more embodiments, the in-phase LO-I signal and the quadrature LO-Q signal are the real and imaginary parts, respectively, of the signal having the LO frequency and supplied by the local oscillator 603. In one or more embodiments, the de-rotation of step a' of fig. 4 is performed by mixers 612 and 614, respectively. In one or more embodiments, the mixed signals at the outputs of mixers 612 and 614 have in-phase and quadrature intermediate frequencies IF, respectively.

In one or more embodiments, the mixing signals at the outputs of the mixers 612, 614 are coupled or preferably connected, respectively, to different frequency filters 622, 623 of the second module BB, which are configured to filter out frequencies that are e.g. two or three times higher than the NZIF frequency. In one or more embodiments, the filtering performs attenuation of the band signal, which relaxes the requirements of the other components of the receiver.

In one or more embodiments, the outputs of the filters 622, 623 are coupled or preferably connected, respectively, to different series of amplifiers (626, 628, 629 for the first path 611 and 624, 625, 627 for the second path 613), which are programmable gain amplifiers of, for example, the second module BB. The number of amplifiers may depend on the application.

In one or more embodiments, the outputs of the amplifiers 629 and 627 are coupled or preferably connected to different analog-to-digital converters (ADCs) 631, 632, respectively, of the third module to convert the filtered and amplified signal of the second module 632 to a digital signal.

In one or more embodiments, the fourth module DFE includes optional DC offset cancellation circuitry 633, 634, coupling the output of the analog-to-digital converter 631 to the mixer 640 for the first path 611 and the output of the analog-to-digital converter 632 to the other mixer 641 for the second path 613. In one or more embodiments, the DC offset cancellation circuits 633, 634 are configured to cancel unwanted DC offsets that may originate from the received signal Frx or from the ADC circuit to improve system performance degradation and bit error rate.

In the example shown, an oscillator 646 (NCO) of the fourth module DFE, which is for example a digitally controlled oscillator, supplies a signal nco_if with an intermediate frequency IF to the mixers 640 and 641.

The de-rotation of step B' in fig. 4 is implemented by mixers 640, 641 for the signal of the first path and the signal of the second path, respectively.

In one or more embodiments, a low pass filter 642 (LPF) of the first path couples the output of the mixer 640 to a first decimator 644 (decimator)Select 8) and another low pass filter 643 (LPF) of the second path couples the output of the mixer 641 to a second decimator 645 (decimatorSelection 8). In one or more embodiments, the decimator is configured to reduce the data rate by removing samples from the data stream without affecting the signal. In the illustrated example, the decimator is configured as an eighth decimator. Other configurations are also possible, such as one-half decimation. In one or more embodiments, the half decimation function corresponds to a clocked (clocked) data converter at half the original rate, with the analog anti-aliasing filter at half the original nyquist bandwidth. In one or more embodiments, decimation is filtered by eliminating unwanted signal images. It also eliminates half of the noise power. In one or more embodiments, there is an overall signal-to-noise ratio (SNR) improvement since the desired signal remains unchanged and the noise power is reduced by half. In one or more embodiments, the SNR is improved by 10 x log (D) for any decimation factor D.

In one or more embodiments, the outputs of decimators 644 and 645 are coupled or preferably connected to Digital Signal Processor (DSP) inputs 648 (I path) and 647 (Q path), respectively, to analyze the demodulated received signal.

Fig. 7A illustrates frequency sensitivity measurements obtained by the method of fig. 3.

In the example of fig. 7A, the minimum signal sensitivity 702 required by the NBIOT standard is about-107.5 dB. In one or more embodiments, clock harmonic spurs produce sensitivity losses at frequencies around 1824MHz (95X 19.2 MHz), 1843MHz (96X 19.2 MHz), and 1862MHz (97X 19.2 MHz), which may be near acceptable limits.

In the example of fig. 7B, once the algorithm of fig. 5 is implemented at frequencies around 1824MHz, 1843MHz, and 1862MHz, the sensitivity loss due to clock harmonic spurs is significantly reduced.

Various embodiments and modifications have been described. Those skilled in the art will appreciate that certain features of the embodiments may be combined and that other modifications will readily occur to those skilled in the art. Specifically, the example of fig. 4 only considers a single reference clock signal. In one or more embodiments, if multiple clocks with non-integer ratios have a risk of causing channel hearing loss, the algorithm may be applied to all clocks in a sequence.

Finally, in one or more embodiments, practical implementations of the embodiments and variations described herein are within the ability of one skilled in the art based on the functional description provided above. In particular, in one or more embodiments, the proposed methods and circuits may be used for different RF signal bands, such as 4G, 5G, ioT or non-cellular RF standards BT, zigBee or other industry standards.

Claims

1. A method for demodulating a radio frequency, RF, signal, the method comprising:

determining the nearest harmonic wave of the clock signal according to the center frequency of the receiving frequency band; and

When the nearest harmonic is within the receive band, an intermediate frequency of a near-zero intermediate frequency NZIF receiver is set to a difference between the center frequency and the nearest harmonic.

2. The method of claim 1, wherein the intermediate frequency is set to a nominal NZIF value when the nearest harmonic is outside the receive frequency band.

3. The method of claim 1, wherein determining the nearest harmonic comprises: a rounded value of a ratio between the center frequency of the receive band and a frequency value of the clock signal is determined.

4. The method of claim 3, wherein the nearest harmonic is equal to a product of the rounded value multiplied by the frequency value of the clock signal.

5. The method according to claim 4, wherein:

When the absolute value of the difference between the center frequency and the nearest harmonic is less than a half-bandwidth value of the RF signal, the nearest harmonic is within the receive band, and

The nearest harmonic is outside the receive band when the absolute value of the difference between the center frequency and the nearest harmonic is not less than the half-bandwidth value of the RF signal.

6. The method of claim 1, further comprising:

Amplifying the RF signal; and

The amplified RF signal is split into a first path and a second path to obtain a first path signal and a second path signal.

7. The method of claim 6, further comprising:

mixing the first path signal with an in-phase signal of a local oscillator frequency, which corresponds to the sum of the center frequency and the intermediate frequency, to obtain a first mixed signal; and

The second path signal is mixed with a quadrature signal of the local oscillator frequency to obtain a second mixed signal.

8. The method of claim 7, further comprising:

Filtering a first high frequency of the first mixed signal of the first path and a first high frequency of the second mixed signal of the second path to obtain a first filtered signal corresponding to the first mixed signal and a second filtered signal corresponding to the second mixed signal; and

Amplifying the first filtered signal and the second filtered signal to obtain a first amplified filtered signal corresponding to the first filtered signal and a second amplified filtered signal corresponding to the second filtered signal.

9. The method of claim 8, further comprising: the first amplified filtered signal and the second amplified filtered signal are converted to digital signals.

10. The method of claim 9, further comprising: the digital signal is mixed with a third signal having the intermediate frequency to obtain a digital mixed signal.

11. The method of claim 10, further comprising: the second high frequency of the digital mixed signal is filtered.

12. The method of claim 11, comprising: and performing the decimation operation of the digital mixed signal.

13. An apparatus for demodulating a radio frequency, RF, signal, comprising:

an RF signal demodulation circuit configured to:

14. The apparatus of claim 13, wherein the intermediate frequency is set to a nominal NZIF value when the nearest harmonic is outside the receive frequency band.

15. The apparatus of claim 13, wherein determining the nearest harmonic comprises: a rounded value of a ratio between the center frequency of the receive band and a frequency value of the clock signal is determined.

16. The device of claim 15, wherein the nearest harmonic is equal to a product of the rounded value multiplied by the frequency value of the clock signal.

17. The apparatus of claim 16, wherein:

18. An apparatus for demodulating a radio frequency, RF, signal, comprising:

an RF signal demodulation circuit configured to:

determining the nearest harmonic wave of the clock signal according to the center frequency of the receiving frequency band;

Setting an intermediate frequency of a near-zero intermediate frequency NZIF receiver to be the difference between the center frequency and the nearest harmonic when the nearest harmonic is within the receive band;

Amplifying the RF signal; and

19. The apparatus of claim 18, wherein the RF signal demodulation circuit is further configured to:

Mixing the first path signal with an in-phase signal of a local oscillator frequency corresponding to the sum of the center frequency and the intermediate frequency to obtain a first mixed signal, and

20. The apparatus of claim 19, wherein the RF signal demodulation circuit is further configured to:

21. The apparatus of claim 20, wherein the RF signal demodulation circuit is further configured to convert the first amplified filtered signal and the second amplified filtered signal to digital signals.

22. The apparatus of claim 21, wherein the RF signal demodulation circuit is further configured to mix the digital signal with a third signal having the intermediate frequency to obtain a digital mixed signal.

23. The apparatus of claim 22, wherein the RF signal demodulation circuit is further configured to filter the second high frequency of the digital mixed signal.

24. The apparatus of claim 23, wherein the RF signal demodulation circuit is further configured to perform a decimation operation of the digital mixing signal.