CN112398398A - Method and device for field weakening control of dual-phase permanent magnet synchronous motor - Google Patents
Method and device for field weakening control of dual-phase permanent magnet synchronous motor Download PDFInfo
- Publication number
- CN112398398A CN112398398A CN202011412252.7A CN202011412252A CN112398398A CN 112398398 A CN112398398 A CN 112398398A CN 202011412252 A CN202011412252 A CN 202011412252A CN 112398398 A CN112398398 A CN 112398398A
- Authority
- CN
- China
- Prior art keywords
- current
- given
- voltage
- sub
- coordinate system
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Pending
Links
- 230000003313 weakening effect Effects 0.000 title claims abstract description 81
- 230000001360 synchronised effect Effects 0.000 title claims abstract description 76
- 238000000034 method Methods 0.000 title claims abstract description 42
- 230000004907 flux Effects 0.000 claims description 43
- 230000009466 transformation Effects 0.000 claims description 38
- 239000011159 matrix material Substances 0.000 claims description 16
- 230000001131 transforming effect Effects 0.000 claims description 8
- 238000004804 winding Methods 0.000 description 18
- 238000010586 diagram Methods 0.000 description 13
- 230000009977 dual effect Effects 0.000 description 9
- 230000001276 controlling effect Effects 0.000 description 6
- 238000006243 chemical reaction Methods 0.000 description 3
- 230000007935 neutral effect Effects 0.000 description 3
- 230000000452 restraining effect Effects 0.000 description 2
- 238000004458 analytical method Methods 0.000 description 1
- 238000004364 calculation method Methods 0.000 description 1
- 230000007547 defect Effects 0.000 description 1
- 238000005516 engineering process Methods 0.000 description 1
- 230000005389 magnetism Effects 0.000 description 1
- 230000001105 regulatory effect Effects 0.000 description 1
- 230000002441 reversible effect Effects 0.000 description 1
Images
Classifications
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/0085—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for high speeds, e.g. above nominal speed
- H02P21/0089—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for high speeds, e.g. above nominal speed using field weakening
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/22—Current control, e.g. using a current control loop
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
- H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
- H02P25/022—Synchronous motors
- H02P25/024—Synchronous motors controlled by supply frequency
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
- H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
- H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using DC to AC converters or inverters
- H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using DC to AC converters or inverters with pulse width modulation
Landscapes
- Engineering & Computer Science (AREA)
- Power Engineering (AREA)
- Control Of Ac Motors In General (AREA)
Abstract
本发明涉及电机技术领域，具体地涉及一种双三相永磁同步电机弱磁控制的方法及装置。采集双三相永磁同步电机的六相定子电流；将六相定子电流通过空间解耦变换到三个子平面，变换为子平面电流；将子平面电流进行旋转变换得到坐标系反馈电流；将坐标系反馈电流和给定电流通过控制器调节得到给定电压，将得到的电压通过矢量控制变换得到电机需要的电压给定值参数；弱磁电流即d轴电流通过弱磁控制器给定，采用第一子平面即α‑β子平面的输出电压幅值作为弱磁控制器的电压负反馈。本发明可以解决传统双三相永磁同步电机在弱磁控制时弱磁电流六次谐波问题和弱磁电流的不平衡问题，在降低双三相电机电流不平衡和谐波电流方面优于传统的弱磁控制。
The invention relates to the technical field of electric motors, in particular to a method and device for field weakening control of a dual-phase permanent magnet synchronous motor. Collect the six-phase stator current of the dual-three-phase permanent magnet synchronous motor; transform the six-phase stator current into three sub-planes through spatial decoupling, and transform into sub-plane currents; rotate the sub-plane currents to obtain the coordinate system feedback current; The feedback current and the given current are adjusted by the controller to obtain the given voltage, and the obtained voltage is transformed by the vector control to obtain the voltage given value parameter required by the motor; the field weakening current, that is, the d-axis current, is given by the field weakening controller, using The output voltage amplitude of the first sub-plane, that is, the α-β sub-plane, is used as the voltage negative feedback of the field weakening controller. The invention can solve the problem of the sixth harmonic of the field-weakening current and the unbalance of the field-weakening current during the field-weakening control of the traditional dual-phase permanent magnet synchronous motor, and is superior in reducing the current imbalance and harmonic current of the dual-phase motor. Traditional field weakening control.
Description
Technical Field
The invention relates to the technical field of motors, in particular to a method and a device for controlling field weakening of a double three-phase permanent magnet synchronous motor.
Background
When the traditional double three-phase permanent magnet synchronous motor is subjected to flux weakening control, because each set of three-phase winding adopts a separate flux weakening controller, flux weakening currents, namely J-axis currents, are given differently, and the currents of the two sets of three-phase windings are unbalanced; and due to the nonlinearity of the inverter and the fifth harmonic voltage and the seventh harmonic voltage generated by the non-sinusoidal back electromotive force of the motor, the voltage feedback in the traditional flux weakening control contains the sixth harmonic voltage, and further, the sixth harmonic current is generated in a dq coordinate system, so that the current THD and the power loss are deteriorated.
The invention can eliminate the sixth harmonic voltage in the feedback voltage of the double three-phase permanent magnet synchronous motor during the flux weakening control, thereby eliminating the flux weakening current sixth harmonic. The current-balancing control method is superior to the traditional flux-weakening control in reducing the current unbalance and harmonic current.
Disclosure of Invention
The embodiment of the invention aims to provide a method and a device for controlling the field weakening of a double three-phase permanent magnet synchronous motor. In order to achieve the above object, a first aspect of the present invention provides a method for flux weakening control of a dual three-phase permanent magnet synchronous motor, including:
collecting six-phase stator current of a double three-phase permanent magnet synchronous motor;
transforming the six-phase stator current into three sub-planes through spatial decoupling to transform into sub-plane currents;
performing rotation transformation on the sub-plane current to obtain a first coordinate system feedback current and a second coordinate system feedback current;
inputting the difference value of the given current of the first coordinate system and the feedback current of the first coordinate system into a first regulator to calculate a first given voltage and a second given voltage, and inputting the difference value of the given current of the second coordinate system and the feedback current of the second coordinate system into a second regulator to calculate a third given voltage and a fourth given voltage;
the first coordinate system given current comprises a first coordinate system d-axis given current id *And a first coordinate system q-axis given current iq *Wherein the first coordinate system d-axis gives the current id *Given by a flux weakening controller, the feedback voltage v of the flux weakening controllermThe voltage is obtained through a first given voltage and a second given voltage;
subjecting the first given voltage and the second given voltage to vector control TparkInversely transforming to obtain fifth given voltage, sixth given voltage, third given voltage and fourth given voltage through vector control TdqzPerforming inverse transformation to obtain a seventh given voltage and an eighth given voltage;
carrying out spatial decoupling inverse transformation on the fifth given voltage, the sixth given voltage, the seventh given voltage and the eighth given voltage to obtain given value parameters of each phase voltage of the double three-phase permanent magnet synchronous motor;
and driving a switching device of the inverter by using PWM (pulse-width modulation) according to the given value parameter of each phase voltage of the double three-phase permanent magnet synchronous motor.
Optionally, the six-phase stator currents are transformed to the three sub-planes by spatial decoupling, including: the spatial decoupling transformation matrix is formula (1);
wherein [ T6]Is a spatial decoupling transformation matrix.
Optionally, the three sub-planes are a fundamental sub-plane, a harmonic sub-plane and a second harmonic sub-plane, respectively; the sub-plane current comprises a fundamental sub-plane current iα、iβHarmonic sub-plane current iz1、iz2And a second harmonic sub-plane current io1、io2。
Optionally, the q axis of the first coordinate system gives the current iq *From torque current command iqcmd *Obtained by means of a current limiter.
Optionally, the feedback voltage v of the field weakening controllermObtained according to the following equation (2):
wherein v ismIs the feedback voltage of the field weakening controller, vd *Is a first given voltage, vq *Is a second given voltage.
Optionally, the second coordinate system gives a given current value idz *And iqz *Are all 0.
Optionally, a fundamental sub-plane current iα、iβAnd harmonic sub-plane current iz1、iz2And performing rotation transformation to obtain a coordinate system feedback current, wherein: fundamental wave sub-plane current iα、iβT defined by formula (3)parkConverted into a first coordinate system feedback current id、iq(ii) a Harmonic sub-plane current iz1、iz2T defined by equation (4)dqzConverted into a second coordinate system feedback current idz、iqz；
Wherein, thetaeIs the rotor electrical angle, Fd、FqIs a component in a fundamental sub-plane in a first coordinate system of the double three-phase permanent magnet synchronous motor, Fdz、FqzIs a component in a harmonic sub-plane under a second coordinate system of the double three-phase permanent magnet synchronous motor, Fα、FβIs a component in the fundamental sub-plane of the double three-phase permanent magnet synchronous motor, Fz1And Fz2Are components in the harmonic sub-plane.
Optionally, calculating, by the first regulator, a given first voltage and a given second voltage by subtracting the given current of the first coordinate system from the feedback current of the first coordinate system, and calculating, by the second regulator, a third given voltage and a fourth given voltage by subtracting the given current of the second coordinate system from the feedback current of the second coordinate system includes:
adjusting the difference between the given current of the second coordinate system and the feedback current of the second coordinate system through a second regulator defined by formula (5) to obtain a third given voltage and a fourth given voltage;
wherein G isPR6(s) is a second regulator, KpAnd KIProportional and harmonic coefficients, ω, of the second regulator, respectivelycTo cut-off frequency, ω6Is the resonance frequency, which is six times the electrical frequency of the fundamental wave of the motor.
Optionally, the performing spatial decoupling inverse transformation on a fifth given voltage, a sixth given voltage, a seventh given voltage, and an eighth given voltage to obtain given value parameters of voltages of each phase of the dual three-phase permanent magnet synchronous motor, further includes:
the given voltage provided by the second harmonic sub-plane is a zero-sequence voltage with a reference value of 0.
The invention provides a device for controlling the field weakening of a double three-phase permanent magnet synchronous motor, which comprises:
configured to perform any of the above-described methods for flux weakening control of a dual three-phase permanent magnet synchronous motor.
The invention can eliminate the sixth harmonic voltage in the feedback voltage of the double three-phase permanent magnet synchronous motor during the flux weakening control, thereby eliminating the flux weakening current sixth harmonic. Compared with the traditional double three-phase weak magnetic control, the double three-phase permanent magnet synchronous motor control method is superior to the traditional weak magnetic control in the aspects of current balance and harmonic current of the double three-phase permanent magnet synchronous motor.
With the above technical solutions, other features and advantages of the embodiments of the present invention will be described in detail in the following detailed description.
Drawings
The accompanying drawings, which are included to provide a further understanding of the embodiments of the invention and are incorporated in and constitute a part of this specification, illustrate embodiments of the invention and together with the description serve to explain the embodiments of the invention without limiting the embodiments of the invention. In the drawings:
fig. 1 schematically shows a flow chart of a method for flux weakening control of a double three-phase permanent magnet synchronous motor according to an embodiment of the invention;
fig. 2 schematically shows a flow chart of a method for flux weakening control of a double three-phase permanent magnet synchronous motor according to an embodiment of the invention;
fig. 3 is a block diagram schematically illustrating a structure of a device for flux weakening control of a double three-phase permanent magnet synchronous motor according to an embodiment of the present invention;
FIG. 4a is a schematic diagram showing the sixth harmonic voltage content in the voltage feedback in the conventional field weakening control and in the voltage feedback in two sets of dq coordinate systems in the conventional field weakening control;
FIG. 4b is a schematic diagram showing the d-axis current feedback under the conventional field weakening control and the 6 th harmonic current content in the d-axis current feedback under the conventional field weakening control;
FIG. 4c is a schematic diagram showing the q-axis current feedback in the conventional field weakening control and the 6 th harmonic current content in the q-axis current feedback in the conventional field weakening control;
FIG. 5a is a schematic diagram showing the voltage feedback and the six-order harmonic voltage content of the weak magnetic control method in the voltage feedback in two sets of dq coordinate systems during weak magnetic control;
FIG. 5b is a schematic diagram showing the d-axis current feedback of the flux weakening control method and the 6 th harmonic current content in the J-axis current feedback of the flux weakening control method;
fig. 5c schematically shows a q-axis current feedback of the field weakening control method and a schematic diagram of the 6 th harmonic current content of the field weakening control method in the q-axis current feedback.
Detailed Description
The following detailed description of embodiments of the invention refers to the accompanying drawings. It should be understood that the detailed description and specific examples, while indicating embodiments of the invention, are given by way of illustration and explanation only, not limitation.
Fig. 1 schematically shows a schematic diagram of a method for flux weakening control of a dual three-phase permanent magnet synchronous motor according to an embodiment of the invention. As shown in fig. 1, in an embodiment of the present invention, a method for flux weakening control of a dual three-phase permanent magnet synchronous motor is provided, including the following steps:
102, transforming the six-phase stator current into three sub-planes through spatial decoupling so as to transform the six-phase stator current into a sub-plane current;
six-phase stator currents of the double three-phase permanent magnet synchronous motor are collected, and the six-phase stator currents of the double three-phase permanent magnet synchronous motor are converted into three sub-planes through a space decoupling transformation matrix and converted into sub-plane currents.
In one embodiment, the six-phase stator currents are transformed to three sub-planes by spatial decoupling, including: the spatial decoupling transformation matrix is formula (1);
wherein [ T6]Is a spatial decoupling transformation matrix.
The six-phase stator current of the double three-phase permanent magnet synchronous motor is converted into three sub-planes through the formula (1) so as to be converted into sub-plane current.
In one embodiment, the three sub-planes are a fundamental sub-plane, a harmonic sub-plane, and a second harmonic sub-plane, respectively; the sub-plane current comprises a fundamental sub-plane current iα、iβHarmonic sub-plane current iz1、iz2And a second harmonic sub-plane current io1、io2。
The fundamental sub-plane is alpha-beta sub-plane, and the harmonic sub-plane is z1-z2Sub-plane, the second harmonic sub-plane being o1-o2A sub-plane. Six-phase stator current of the double three-phase permanent magnet synchronous motor is collected, and the six-phase stator current of the double three-phase permanent magnet synchronous motor is converted to an alpha-beta sub-plane and z through a space decoupling transformation matrix formula (1)1-z2Sub-planes and o1-o2Three sub-planes to form a sub-plane current, wherein the alpha-beta sub-plane current is iα、iβ，z1-z2A sub-plane current of iz1、iz2，o1-o2A sub-plane current of io1、io2。
Six-phase stator current of the double three-phase permanent magnet synchronous motor is transformed to three planes through a space decoupling transformation matrix formula (1), and the six-phase stator current is respectively alpha-beta sub-plane current: i.e. iα、iβ；z1-z2Sub-plane current: i.e. iz1、iz2(ii) a Because two sets of windings of the double three-phase permanent magnet synchronous motor are isolated from neutral points, the motor has a structure of1-o2The sub-planes have no current.
103, carrying out rotation conversion on the sub-plane current to obtain a first coordinate system feedback current and a second coordinate system feedback current;
six-phase stator current of double three-phase permanent magnet synchronous motor is transformed through spatial decouplingMatrix equation (1), transformed to three planes, is the α - β sub-plane current: i.e. iα、iβ；z1-z2Sub-plane current: i.e. iz1、iz2Converting the sub-plane current iα、iβRotating and transforming to obtain feedback current of the first coordinate system and converting the current of the sub-plane iz1、iz2The rotation transformation obtains a second coordinate system feedback current, the first coordinate system is a dq coordinate system, and the second coordinate system is dqz coordinate system. And performing rotation transformation on the sub-plane current to obtain dq coordinate system feedback current and dqz coordinate system feedback current.
In one embodiment, the fundamental sub-plane current iα、iβAnd harmonic sub-plane current iz1、iz2And performing rotation transformation to obtain a coordinate system feedback current, wherein: fundamental wave sub-plane current iα、iβT defined by formula (3)parkConverted into a first coordinate system feedback current id、iq(ii) a Harmonic sub-plane current iz1、iz2T defined by equation (4)dqzConverted into a second coordinate system feedback current idz、iqz；
Wherein theta iseIs the rotor electrical angle, Fd、FqIs a component in a fundamental sub-plane in a first coordinate system of the double three-phase permanent magnet synchronous motor, Fdz、FqzIs a component in a harmonic sub-plane under a second coordinate system of the double three-phase permanent magnet synchronous motor, Fα、FβIs a component in the fundamental sub-plane of the double three-phase permanent magnet synchronous motor, Fz1And Fz2Are components in the harmonic sub-plane.
Alpha-beta sub-plane current iα、iβThrough the formula (3) TparkConversion to dq coordinate system feedback current id、iq，z1-z2Sub-plane current iz1、iz2Through the formula (4) TdqzTransformed into dqz coordinate system feedback current idz、iqz. Theta in the formula (3)eIs the rotor electrical angle, Fd、FqIs a component in an alpha-beta sub-plane under a dq coordinate system of the double three-phase permanent magnet synchronous motor, Fα、FβIs a component in an alpha-beta sub-plane of the double three-phase permanent magnet synchronous motor, theta in formula (4)eIs the rotor electrical angle, Fdz、FqzIs z under a double three-phase permanent magnet synchronous motor dqz coordinate system1-z2Component of the sub-plane, Fz1And Fz2Is z1-z2Component in the sub-plane.
And 104, inputting the difference value of the given current of the first coordinate system and the feedback current of the first coordinate system into a first regulator to calculate a first given voltage and a second given voltage, and inputting the difference value of the given current of the second coordinate system and the feedback current of the second coordinate system into a second regulator to calculate a third given voltage and a fourth given voltage.
The first regulator may be a proportional integral regulator and the second regulator may be a proportional resonant regulator. And calculating the difference between the dq coordinate system given current and the dq coordinate system feedback current through a proportional integral regulator to obtain a first given voltage and a second given voltage. Wherein the first given voltage is
The second given voltage is
dqz the difference between the given current of the coordinate system and the feedback current of the dqz coordinate system is calculated by a proportional resonant regulator to obtain a third given voltage and a fourth given voltage, wherein the third given voltage is
A fourth given voltage of
In one embodiment, calculating the difference between the first coordinate system set current and the first coordinate system feedback current by the first regulator to obtain the given first voltage and the given second voltage, and calculating the difference between the second coordinate system set current and the second coordinate system feedback current by the second regulator to obtain the third set voltage and the fourth set voltage comprises: regulating the given current of the second coordinate system and the feedback current of the second coordinate system through a second regulator defined by a formula (5) to obtain a third given voltage and a fourth given voltage;
wherein G isPR6(s) is a second regulator, KpAnd KIProportional and harmonic coefficients, ω, of the second regulator, respectivelycTo cut-off frequency, ω6Is the resonance frequency, which is six times the electrical frequency of the fundamental wave of the motor.
The difference between the given current of the dq coordinate system and the feedback current of the dq coordinate system is calculated by a proportional-integral regulator to obtain a first given voltage
And a second given voltage
dqz coordinate system given current and dqz coordinate system feedback current are differed to obtain third given voltage through calculation of a proportional resonant regulator
And a fourth given voltage
Wherein the proportional resonant controller is GPR6(s), the expression is:
wherein KpAnd KIProportional coefficient and resonance coefficient, omega, of the proportional resonance controller, respectivelycTo cut-off frequency, ω6The proportional resonant controller has maximum gain at the resonant point and no phase lag at the resonant frequency which is equal to the six-time multiplication of the fundamental wave electric frequency of the motor, and the resonant point can be adjusted in real time according to the motor frequency.
105, the first coordinate system given current comprises a first coordinate system d-axis given current id *And a first coordinate system q-axis given current iq *Wherein the first coordinate system d-axis gives the current id *Given by a flux weakening controller, the feedback voltage v of the flux weakening controllermObtained by a first given voltage and a second given voltage.
In one embodiment, the first coordinate system q-axis gives the current iq *From torque current command iqcmd *Obtained by means of a current limiter.
In one embodiment, the feedback voltage v of the field weakening controllermObtained according to the following equation (2):
wherein v ismIs the feedback voltage of the field weakening controller, vd *Is a first given voltage, vq *Is a second given voltage.
In one embodiment, the second coordinate system gives the given current value idz *And iqz *Are all 0.
The dq coordinate system given current comprises a d-axis given current id *And q-axis set current iq *Q-axis given current iq *Commanded by torque current iqcmd *Obtained by means of a current limiter. d-axis given current id *Obtained by means of a flux weakening control, the voltage in the flux weakening controlThe feedback being a first given voltage using the alpha-beta sub-plane
And a second given voltage
Feedback voltage vmBy applying a first given voltage to the alpha-beta sub-plane
And a second given voltage
Obtained by the formula (2). dqz coordinate system given current idz *And iqz *All the current values are 0, in the traditional double three-phase permanent magnet synchronous motor flux weakening control method, each set of three-phase winding adopts an independent flux weakening controller, and voltage feedback of an alpha-beta coordinate system of each three-phase winding is adopted, flux weakening currents can be different and comprise fifth harmonic voltage and seventh harmonic voltage, and meanwhile, the same torque current instruction iqcmd *Different q-axis current settings may eventually be generated by two different current limiters. Therefore, the traditional method for controlling the field weakening of the double three-phase permanent magnet synchronous motor has the defects of current imbalance and current harmonic.
In one embodiment, the first coordinate system gives the current d-axis given value id *The current i is given to the q axis of the first coordinate system by weak magnetic controlq *From torque current command iqcmd *Obtained by a current limiter; given current given value i of second coordinate systemdz *And iqz *Are all 0.
In the traditional double three-phase permanent magnet synchronous motor flux weakening control method, flux weakening currents given by dq coordinate systems of two sets of three-phase motor windings are respectively obtained through two independent flux weakening controllers, but in the embodiment of the invention, the flux weakening currents given by the dq coordinate systems of the two sets of three-phase motor windings are obtained through flux weakening control under an alpha-beta sub-plane dq coordinate system.
106, performing vector control on the first given voltage, the second given voltage, the third given voltage and the fourth given voltage to reverse TparkTransformation and inverse TdqzAnd converting to obtain a fifth given voltage, a sixth given voltage, a seventh given voltage and an eighth given voltage.
A first given voltage
And a second given voltage
By vector control of inverse TparkConverted to obtain a fifth given voltage
And a sixth given voltage
A third given voltage
And a fourth given voltage
By reversing TdqzConverted to obtain a seventh given voltage
And an eighth given voltage
a fifth given voltage
A sixth given voltage
A seventh given voltage
And an eighth given voltage
And carrying out spatial decoupling inverse transformation to obtain given value parameters of each phase voltage of the double three-phase permanent magnet synchronous motor. And the space decoupling inverse transformation is the inverse matrix transformation of the space decoupling transformation formula (1) matrix.
In one embodiment, the spatial decoupling inverse transformation is performed on a fifth given voltage, a sixth given voltage, a seventh given voltage and an eighth given voltage to obtain given value parameters of voltages of each phase of the dual three-phase permanent magnet synchronous motor, and the method further includes:
the given voltage provided by the second harmonic sub-plane is a zero-sequence voltage with a reference value of 0.
A fifth given voltage
A sixth given voltage
A seventh given voltage
And an eighth given voltage
Carrying out space decoupling inverse transformation to obtain given value parameters of each phase voltage of the double three-phase permanent magnet synchronous motor, carrying out space decoupling inverse transformation to obtain inverse matrix transformation of a space decoupling transformation formula (1) matrix, wherein the given voltage further comprises o1-o2Given voltage of the sub-plane due to o1-o2The given voltage of the sub-plane is zero sequence voltage, so o1-o2 Voltage reference value 0, v of the sub-planeo1 *＝vo2 *＝0。
And step 108, driving the switching device of the inverter by the given value parameter of each phase voltage of the double three-phase permanent magnet synchronous motor through a PWM (pulse width modulation) technology.
The given value parameter of each phase voltage of the double three-phase permanent magnet synchronous motor is respectively
In one embodiment, as shown in fig. 2, a flow chart of the flux weakening control of the dual three-phase permanent magnet synchronous motor based on space vector decoupling is shown.
Based on the vector space decoupling theory, the double three-phase permanent magnet synchronous motor is mapped to a six-dimensional plane. Respectively alpha-beta sub-plane, z1-z2Sub-planes and o1-o2A sub-plane. Six-phase stator current i of double three-phase permanent magnet synchronous motor is collecteda、ib、ic、ix、iy、izThrough formula (1) space decoupling transformation matrix [ T ]6]The six-phase stator current ia、ib、ic、ix、iy、izTransformed to three sub-planes for conversion to sub-plane currents, respectively alpha-beta sub-plane currents iα、iβ，z1-z2Sub-plane current iz1、iz2、o1-o2Sub-plane current io1、io2Wherein two sets of windings of the double three-phase permanent magnet synchronous motor are isolated from each other by neutral points, so1-o2The sub-planes have no current. Wherein, formula (1) is:
fundamental waves and 12k +/-1 (k is 1, 2, 3 …) subharmonics in the stator current of the motor are distributed on an alpha-beta sub-plane, and 6k +/-1 (k is 1, 3, 5 …) subharmonics are distributed on a z-beta sub-plane1-z2Sub-plane, 6 k. + -.3(k-1, 3, 5 …) subharmonic distribution at o1-o2A sub-plane. Since the fifth and seventh harmonics are mapped to z1-z2The sub-plane is not the alpha-beta sub-plane, and the alpha-beta sub-plane voltage can be used for voltage feedback in the field weakening control of the double three-phase permanent magnet motor according to the characteristic.
Alpha-beta sub-plane current iα、iβBy the formula (3) TparkTransforming to dq coordinate system to obtain feedback current i of dq coordinate systemd、iq，z1-z2Sub-plane current iz1、iz2By the formula (4) TdqzTransforming to dqz coordinate system to obtain dqz coordinate system feedback current idz、iqz. Wherein, the formula (3) and the formula (4) are:
in particular where thetaeIs the rotor electrical angle, Fd、FqIs a component in an alpha-beta sub-plane under a dq coordinate system of the double three-phase permanent magnet synchronous motor, Fdz、FqzIs z under a double three-phase permanent magnet synchronous motor dqz coordinate system1-z2Component in the sub-plane, Fα、FβIs a component in an alpha-beta sub-plane of the double three-phase permanent magnet synchronous motor, Fz1And Fz2Is z1-z2Component in the sub-plane. F may be the stator resistance RsStator voltage v, stator current i, stator flux linkage ΨsOr a permanent magnetic linkage Ψf。
dq coordinate system given current id *Is obtained by weak magnetic control, and the q axis of the first coordinate system gives a current iq *From torque current command iqcmd *Obtained by means of a current limiter, dqz coordinate system gives a current idz *And iqz *Are all 0. The dq coordinate system is given a current id *And iq *Feedback current i with dq coordinate systemd、iqCalculating the difference by a proportional-integral regulator to obtain a given voltage
And a given voltage
The difference between the dqz coordinate system given current and the dqz coordinate system feedback current is calculated by a formula (5) proportional resonant regulator to obtain a given voltage
And a given voltage
Wherein the proportional resonant regulator of formula (5) is GPR6(s), the expression of which is:
wherein KpAnd KIProportional coefficient and resonance coefficient, omega, of the proportional resonance controller, respectivelycTo cut-off frequency, ω6The proportional resonant controller has maximum gain at the resonant point and no phase lag at the resonant frequency which is equal to the six-time multiplication of the fundamental wave electric frequency of the motor, and the resonant point can be adjusted in real time according to the motor frequency.
FIG. 2 shows on the left part a field weakening controller, with dq coordinate system giving current id *Is obtained by weak magnetic control, and a dq coordinate system gives a current iq *From torque current command iqcmd *Obtained by means of a current limiter. Output voltage v of alpha-beta sub-plane under dq coordinate systemd *、vq *The amplitude feedback voltage has an amplitude of
d-axis current set value id *Limited by voltage amplitude vm *And the output voltage vmIs obtained by adjusting a proportional-integral controller when the output voltage v ismLess than vm *When i isd *0; when the output voltage v ismGreater than vm *When i isd *Is less than 0. Given value of q-axis current iq *From torque current command iqcmd *Obtained by a current limiter whose value is defined by a current maximum value ImaxSquare of minus d-axis current set value id *The square root number of (a). In particular to
A given voltage to be obtained
And a given voltage
By the formula (3) TparkInverse transformation of (a) to obtain a given voltage
And a given voltage
A given voltage to be obtained
And a given voltage
By the formula (4) TdqzInverse transformation of (a) to obtain a given voltage
And a given voltage
Because two sets of windings of the double three-phase permanent magnet synchronous motor are isolated from neutral points, the motor has the advantages of high efficiency, low cost and high efficiency1-o2Voltage reference value of the sub-plane is vo1 *＝vo2 *＝0。
A given voltage to be obtained
vo1 *、vo2 *A space decoupling transformation matrix [ T ] through a formula (1)6]Obtaining the reference value of each phase voltage of the double three-phase permanent magnet synchronous motor by inverse transformation
The switching devices of the inverter are driven through a PWM modulation technique.
Fig. 4a is a schematic diagram showing the sixth harmonic voltage content in the voltage feedback of the two sets of dq coordinate system in the conventional field weakening control and the conventional field weakening control.
Fig. 4b schematically shows a diagram of the current content of the 6 th harmonic in the d-axis current feedback in the conventional field weakening control and the d-axis current feedback in the conventional field weakening control.
Fig. 4c schematically shows a diagram of the current content of the 6 th harmonic in the q-axis current feedback in the conventional field weakening control and the q-axis current feedback in the conventional field weakening control.
FIG. 5a is a schematic diagram showing the voltage feedback and the six-order harmonic voltage content of the weak magnetic control method in the voltage feedback in two sets of dq coordinate systems during weak magnetic control.
Fig. 5b schematically shows a d-axis current feedback of the field weakening control method and a 6 th harmonic current content in the d-axis current feedback of the field weakening control method.
Fig. 5c schematically shows a q-axis current feedback of the field weakening control method and a schematic diagram of the 6 th harmonic current content of the field weakening control method in the q-axis current feedback.
Compared with the prior art, the embodiment of the invention has the following advantages:
the first advantage is that: in order to compare the advantages and disadvantages of the conventional flux weakening control method and the proposed flux weakening control method, the experimental results are shown in fig. 4 and 5, respectively. The magnitude of the output feedback voltage is shown in fig. 4(a) and 5 (a). The results show that the sixth harmonic voltage in fig. 5(a) is lower than that in fig. 4 (a). Therefore, in consideration of harmonic voltage and inverter nonlinearity, the margin of the reserve voltage can be smaller in the linear PWM operation, thereby improving the maximum output voltage reference value and power capacity.
The second advantage is that: FIGS. 4(b) and 5(b) show d-axis currents and corresponding FFT analysis, comparing fig. 4(b) to i in FIG. 5(b)d1And id2The 6 th harmonic in (a) is negligible. i.e. id1And id2Is the same, and i of FIG. 4(b)d1And id2The average values are not equal, and the phenomenon of current imbalance occurs. FIG. 5(c) shows iq1And iq2The two windings are basically the same, so the flux weakening control decoupled according to the space vector provided by the invention has good current balance degree for the two sets of three-phase windings.
Therefore, the field weakening control method provided by the invention is superior to the traditional field weakening control of the double three-phase permanent magnet synchronous motor in the aspects of restraining the current unbalance degree and restraining the dq axis six-order harmonic current.
In one embodiment, as shown in fig. 3, there is provided an apparatus for flux weakening control of a dual three-phase permanent magnet synchronous motor, configured to perform the method for flux weakening control of a dual three-phase permanent magnet synchronous motor in any one of the above embodiments.
Specifically, the device for controlling the field weakening of the double three-phase permanent magnet synchronous motor is a double three-phase permanent magnet synchronous motor. The rotor of the double three-phase permanent magnet synchronous motor is a permanent magnet, and the stator winding is connected by two sets of Y-shaped windings which are respectively A, B, C phases and X, Y, Z phases;
two sets of three-phase windings of the stator winding have a spatial difference of 30 degrees in electrical angle, each set of windings is powered by a set of three-phase two-level inverter, and the inverter switching device is an IGBT.
Since the fundamental component is projected onto the α - β sub-plane, the output voltage of the inverter is mostly derived from the fundamental component in the α - β sub-plane, and can be used for voltage feedback in the weak magnetic control. Because the alpha-beta sub-plane has no 5 th harmonic and 7 th harmonic, the weak magnetic reference current is more favorably generated. Meanwhile, q-axis currents generated by the same weak magnetic currents of the two sets of windings are also the same, so that the currents of the two sets of three-phase windings of the double three-phase permanent magnet synchronous motor are naturally balanced.
Claims (10)
1. A method for flux weakening control of a double three-phase permanent magnet synchronous motor is characterized by comprising the following steps:
collecting six-phase stator current of a double three-phase permanent magnet synchronous motor;
transforming the six-phase stator currents to three sub-planes through spatial decoupling to transform into sub-plane currents;
performing rotation transformation on the sub-plane current to obtain a first coordinate system feedback current and a second coordinate system feedback current;
inputting the difference value of the given current of the first coordinate system and the feedback current of the first coordinate system into a first regulator to calculate a first given voltage and a second given voltage, and inputting the difference value of the given current of the second coordinate system and the feedback current of the second coordinate system into a second regulator to calculate a third given voltage and a fourth given voltage;
the first coordinate system given current comprises a first coordinate system d-axis given current id *And a first coordinate system q-axis given current iq *Wherein the first coordinate system d axis gives a current id *Given by a flux weakening controller, the feedback voltage v of said flux weakening controllermThe first given voltage and the second given voltage are obtained;
subjecting the first given voltage and the second given voltage to vector control TparkInversely transforming to obtain a fifth given voltage and a sixth given voltage, wherein the third given voltage and the fourth given voltage are controlled by a vector RdqzPerforming inverse transformation to obtain a seventh given voltage and an eighth given voltage;
carrying out spatial decoupling inverse transformation on the fifth given voltage, the sixth given voltage, the seventh given voltage and the eighth given voltage to obtain given value parameters of each phase voltage of the double three-phase permanent magnet synchronous motor;
and driving a switching device of an inverter by using PWM (pulse-width modulation) according to given value parameters of each phase voltage of the double three-phase permanent magnet synchronous motor.
2. The method of claim 1, wherein said transforming the six-phase stator currents to three sub-planes through spatial decoupling comprises: transforming the six-phase stator currents to three sub-planes using a spatial decoupling transformation matrix,
wherein the spatial decoupling transformation matrix is defined as formula (1);
wherein [ T6]Is the spatial decoupling transformation matrix.
3. The method of claim 1, wherein the three sub-planes are a fundamental sub-plane, a harmonic sub-plane, and a second harmonic sub-plane, respectively;
the sub-plane current comprises a fundamental sub-plane current iα、iβHarmonic sub-plane current iz1、iz2And a second harmonic sub-plane current io1、io2。
4. The method of claim 1, wherein the first coordinate system q-axis gives a current iq *From torque current command iqcmd *Obtained by means of a current limiter.
5. Method according to claim 1, characterized in that the feedback voltage v of the field weakening controllermObtained according to the following equation (2):
wherein v ismIs the feedback voltage, v, of the field weakening controllerd *Is said first given voltage, vq *Is the second given voltage.
6. Method according to claim 1, characterized in that said second coordinate system gives a given current value idz *And iqz *Are all 0.
7. Method according to claim 3, characterized in that the fundamental sub-plane current i is injectedα、iβAnd the harmonic sub-plane current iz1、iz2And performing rotation transformation to obtain the coordinate system feedback current, wherein:
the fundamental wave sub-plane current iα、iβT defined by formula (3)parkIs converted into the first coordinate system feedback current id、iq；
The harmonic sub-plane current iz1、iz2T defined by equation (4)dqzIs converted into the second coordinate system feedback current idz、iqz；
Wherein, thetaeIs the rotor electrical angle, Fd、FqIs a component in a fundamental sub-plane in a first coordinate system of the double three-phase permanent magnet synchronous motor, Fdz、FqzIs a harmonic sub-plane under a second coordinate system of the double three-phase permanent magnet synchronous motorComponent (b) of (1), Fα、FβIs a component in the fundamental sub-plane of the double three-phase PMSM, Fz1And Fz2Are components in the harmonic sub-plane.
8. The method of claim 1, wherein calculating a given first voltage and a given second voltage by a first regulator that differs a first coordinate system given current from the first coordinate system feedback current, and calculating a third given voltage and a fourth given voltage by a second regulator that differs a second coordinate system given current from the second coordinate system feedback current comprises:
adjusting the difference between the given current of the second coordinate system and the feedback current of the second coordinate system through a second regulator defined by formula (5) to obtain a third given voltage and a fourth given voltage;
wherein G isPR6(s) is the second regulator, KpAnd KIProportional and harmonic coefficients, ω, of the second regulator, respectivelycTo cut-off frequency, ω6Is the resonance frequency, which is six times the electrical frequency of the fundamental wave of the motor.
9. The method of claim 1, wherein the fifth given voltage, the sixth given voltage, the seventh given voltage and the eighth given voltage are subjected to spatial decoupling inverse transformation to obtain voltage set-point parameters of each phase of the double three-phase permanent magnet synchronous motor, and further comprising:
the given voltage provided by the second harmonic sub-plane is a zero-sequence voltage, and the reference value of the given voltage is 0.
10. An apparatus for flux weakening control of a double three-phase permanent magnet synchronous motor, characterized by being configured to perform the method for flux weakening control of a double three-phase permanent magnet synchronous motor according to any one of claims 1 to 9.
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CN202011412252.7A CN112398398A (en) | 2020-12-03 | 2020-12-03 | Method and device for field weakening control of dual-phase permanent magnet synchronous motor |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CN202011412252.7A CN112398398A (en) | 2020-12-03 | 2020-12-03 | Method and device for field weakening control of dual-phase permanent magnet synchronous motor |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CN112398398A true CN112398398A (en) | 2021-02-23 |
Family
ID=74604401
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CN202011412252.7A Pending CN112398398A (en) | 2020-12-03 | 2020-12-03 | Method and device for field weakening control of dual-phase permanent magnet synchronous motor |
Country Status (1)
| Country | Link |
|---|---|
| CN (1) | CN112398398A (en) |
Cited By (7)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN113839595A (en) * | 2021-08-11 | 2021-12-24 | 湖南大学 | Harmonic and Unbalanced Current Suppression Methods for Dual-phase Permanent Magnet Synchronous Motors |
| CN113938061A (en) * | 2021-09-15 | 2022-01-14 | 安徽理工大学 | Method for reducing speed fluctuation of linear feeding system |
| CN114050748A (en) * | 2021-10-20 | 2022-02-15 | 西安理工大学 | Five-phase double-excitation synchronous motor partition coordination control method |
| CN115528959A (en) * | 2021-06-24 | 2022-12-27 | 中车株洲电力机车研究所有限公司 | Pulse Width Modulation Method and Related Devices for Six-Phase Motor with Dual Y Windings Shifted by 30 Degrees |
| CN115765559A (en) * | 2022-12-05 | 2023-03-07 | 哈尔滨工业大学 | Nonlinear self-learning method for inverter of double-three-phase permanent magnet synchronous motor driving system |
| CN118631117A (en) * | 2024-06-24 | 2024-09-10 | 长沙理工大学 | Optimization method of flux weakening control of dual three-phase permanent magnet motor based on torque adaptive adjustment |
| CN118826565A (en) * | 2024-06-19 | 2024-10-22 | 哈尔滨工业大学 | Instantaneous power observation method for dual three-phase permanent magnet synchronous motor system |
Citations (7)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| EP2169821A1 (en) * | 2008-09-26 | 2010-03-31 | Vestas Wind Systems A/S | Method and apparatus for dynamic load sharing |
| CN102195552A (en) * | 2010-03-09 | 2011-09-21 | 通用汽车环球科技运作有限责任公司 | Methods, systems and apparatus for approximation of peak summed fundamental and third harmonic voltages in a multi-phase machine |
| TW201236356A (en) * | 2011-02-21 | 2012-09-01 | Jonq-Chin Hwang | Driving controller of synchronous motor and the driving control method thereof |
| WO2017077567A1 (en) * | 2015-11-02 | 2017-05-11 | 三菱電機株式会社 | Electric power steering device and method for controlling same |
| WO2018110242A1 (en) * | 2016-12-12 | 2018-06-21 | 株式会社デンソー | Control device for rotating electrical machine, and rotating electrical machine system |
| CN111277186A (en) * | 2018-12-05 | 2020-06-12 | 哈尔滨工业大学 | Induction motor field weakening control method based on optimized six-beat operation |
| CN111293943A (en) * | 2020-02-19 | 2020-06-16 | 湖南大学 | Control method for open-phase operation of double three-phase motor |
-
2020
- 2020-12-03 CN CN202011412252.7A patent/CN112398398A/en active Pending
Patent Citations (7)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| EP2169821A1 (en) * | 2008-09-26 | 2010-03-31 | Vestas Wind Systems A/S | Method and apparatus for dynamic load sharing |
| CN102195552A (en) * | 2010-03-09 | 2011-09-21 | 通用汽车环球科技运作有限责任公司 | Methods, systems and apparatus for approximation of peak summed fundamental and third harmonic voltages in a multi-phase machine |
| TW201236356A (en) * | 2011-02-21 | 2012-09-01 | Jonq-Chin Hwang | Driving controller of synchronous motor and the driving control method thereof |
| WO2017077567A1 (en) * | 2015-11-02 | 2017-05-11 | 三菱電機株式会社 | Electric power steering device and method for controlling same |
| WO2018110242A1 (en) * | 2016-12-12 | 2018-06-21 | 株式会社デンソー | Control device for rotating electrical machine, and rotating electrical machine system |
| CN111277186A (en) * | 2018-12-05 | 2020-06-12 | 哈尔滨工业大学 | Induction motor field weakening control method based on optimized six-beat operation |
| CN111293943A (en) * | 2020-02-19 | 2020-06-16 | 湖南大学 | Control method for open-phase operation of double three-phase motor |
Non-Patent Citations (2)
| Title |
|---|
| YASHAN HU: "《Control of Dual Three一Phase Permanent Magnet Synchronous Machine B ased on Five一Leg Inverter》", 《IEEE TRANSACTIONS ON POWER ELECTRONTCS》 * |
| 韩天亮: "《基于分数阶滑模的矿用牵引电机调速控制研究》", 《中国优秀硕士学位论文全文数据库（电子期刊）》 * |
Cited By (12)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN115528959A (en) * | 2021-06-24 | 2022-12-27 | 中车株洲电力机车研究所有限公司 | Pulse Width Modulation Method and Related Devices for Six-Phase Motor with Dual Y Windings Shifted by 30 Degrees |
| CN115528959B (en) * | 2021-06-24 | 2025-09-12 | 中车株洲电力机车研究所有限公司 | Pulse width modulation method and related device for six-phase motor with double Y windings shifted 30 degrees |
| CN113839595A (en) * | 2021-08-11 | 2021-12-24 | 湖南大学 | Harmonic and Unbalanced Current Suppression Methods for Dual-phase Permanent Magnet Synchronous Motors |
| CN113839595B (en) * | 2021-08-11 | 2024-05-10 | 湖南大学 | Harmonic wave and unbalanced current suppression method for double three-phase permanent magnet synchronous motor |
| CN113938061A (en) * | 2021-09-15 | 2022-01-14 | 安徽理工大学 | Method for reducing speed fluctuation of linear feeding system |
| CN113938061B (en) * | 2021-09-15 | 2023-10-27 | 安徽理工大学 | A method to reduce the speed fluctuation of linear feed system |
| CN114050748A (en) * | 2021-10-20 | 2022-02-15 | 西安理工大学 | Five-phase double-excitation synchronous motor partition coordination control method |
| CN115765559A (en) * | 2022-12-05 | 2023-03-07 | 哈尔滨工业大学 | Nonlinear self-learning method for inverter of double-three-phase permanent magnet synchronous motor driving system |
| CN115765559B (en) * | 2022-12-05 | 2023-07-14 | 哈尔滨工业大学 | Inverter nonlinear self-learning method for dual-three-phase permanent magnet synchronous motor drive system |
| CN118826565A (en) * | 2024-06-19 | 2024-10-22 | 哈尔滨工业大学 | Instantaneous power observation method for dual three-phase permanent magnet synchronous motor system |
| CN118631117A (en) * | 2024-06-24 | 2024-09-10 | 长沙理工大学 | Optimization method of flux weakening control of dual three-phase permanent magnet motor based on torque adaptive adjustment |
| CN118631117B (en) * | 2024-06-24 | 2025-03-07 | 长沙理工大学 | Double three-phase permanent magnet motor flux weakening control optimization method based on torque self-adaptive adjustment |
Similar Documents
| Publication | Publication Date | Title |
|---|---|---|
| CN112398398A (en) | Method and device for field weakening control of dual-phase permanent magnet synchronous motor | |
| JP5120670B2 (en) | Control device for motor drive device | |
| JP5120669B2 (en) | Control device for motor drive device | |
| CN103401503B (en) | A kind of method and device in harmonic wave plane on-line identification double three-phase machine parameter | |
| EP3494635B1 (en) | Control arrangement for a generator | |
| JP5370769B2 (en) | Control device for motor drive device | |
| CN103997267B (en) | A kind of series compensation Direct Torque Control of driving winding permanent magnet synchronous motor | |
| Singh et al. | Investigation of performance parameters of PMSM drives using DTC-SVPWM technique | |
| US10340831B2 (en) | Wind power generator vibration inhibition method and device | |
| Ammar et al. | Sensorless stator field oriented-direct torque control with SVM for induction motor based on MRAS and fuzzy logic regulation | |
| Kakodia et al. | A comparative study of DFOC and IFOC for IM drive | |
| CN111669091A (en) | Direct torque control method for motor | |
| Lee et al. | Torque control for IPMSM in the high speed range based on voltage angle | |
| Rezaei et al. | Modelling and analysis of indirect field-oriented control of SVPWM-driven induction motor drive based on a voltage source inverter | |
| CN107947669A (en) | A kind of hybrid exciting synchronous motor Nonlinear backstepping tracking and controlling method | |
| Fan et al. | Sensorless control of dual-three phase PMSM based aircraft electric starter/generator system using model reference adaptive system method | |
| JP2018057131A (en) | Flywheel power storage system | |
| Xu et al. | Backstepping direct torque control of permanent magnet synchronous motor with RLS parameter identification | |
| EP2747273B1 (en) | Method and arrangement for torque estimation of a synchronous machine | |
| Can | Analysis and performance with vertical divided multilevel voltage on phase of induction engine | |
| JP5370748B2 (en) | Control device for motor drive device | |
| Liu et al. | Analysis of indirect rotor field oriented control-based induction machine performance under inaccurate field-oriented condition | |
| Kumar et al. | Modified direct torque control of three-phase induction motor drives with low ripple in flux and torque | |
| Cho et al. | Time-optimal transition method for six-step operation of PMSM via geometric prediction of flux trajectory | |
| Zhang et al. | A new scheme to direct torque control of interior permanent magnet synchronous machine drives for constant inverter switching frequency and low torque ripple |
Legal Events
| Date | Code | Title | Description |
|---|---|---|---|
| PB01 | Publication | ||
| PB01 | Publication | ||
| SE01 | Entry into force of request for substantive examination | ||
| SE01 | Entry into force of request for substantive examination | ||
| RJ01 | Rejection of invention patent application after publication | ||
| RJ01 | Rejection of invention patent application after publication |
Application publication date: 20210223 |