CN1855613A - Bandpass wave filter and RF communication device using same - Google Patents

Abstract

A bandpass filter comprising: first to sixth resonators (1 to 6) with one end grounded and a length of substantially 1/4 wavelength; an input coupled to the non-ground end of the first resonator part (IN); and an output part (OUT) coupled to the non-ground end of the sixth resonator, the second to fifth resonators (2 to 5) are electromagnetically coupled to each other, and the second resonator The third resonator and the third resonator are respectively capacitively coupled to the first resonator through the first capacitor (C1) and the second capacitor (C2), and the third resonator and the fourth resonator are capacitively coupled to the first resonator through the third capacitor (C3), the fourth Capacitors (C4) are capacitively coupled to the sixth resonators, respectively, and the input unit and output unit are respectively coupled to the first resonator and the sixth resonator via input and output capacitors (C5, C6). Therefore, it can be applied to ultra-wideband UWB, and a small and low-loss bandpass filter can be realized.

Description

Band-pass filter and wireless communication device using same

technical field

The present invention relates to a UWB (Ultra Wide Band) band pass filter (Band Pass Filter) with wide-band pass characteristics and steep attenuation characteristics applicable to the field of wireless communication, and a wireless communication device using the band pass filter .

UWB is expected to be used as a data transmission medium for PC peripherals such as external storage devices, printers, and scanners, or as a data communication medium for digital televisions, projectors, digital cameras, and digital video cameras.

Background technique

In recent years, UWB has been attracting attention as a new communication method. This UWB is different from a wireless local area network (hereinafter referred to as W-LAN) used as one of data communication methods in terms of communication distance and data transmission speed.

In IEEE802.11.b, which is one of W-LAN standards, the communication distance is 30 to 100 m, the transmission power is 500 mW, and the communication speed is about 11 Mbps. On the other hand, in UWB, when the frequency band is 3.1 to 4.9 GHz, the communication distance is as short as 10m, but the transmission power is 100mW, which is low power consumption. 480Mbps, high-speed data communication is possible compared with W-LAN.

In this way, as one of the characteristics of UWB, a high transmission rate is realized by using a wide frequency band. This bandwidth ratio (bandwidth/center frequency) is 40% or more, and may be 110% or more depending on the case.

In addition, as another characteristic of UWB, the average transmission power density of UWB is specified as a low value of less than -41.25 dBm/MHz. Here, -41.25dBm/MHz is equivalent to generating a radiation power with an electric field strength of 54dBμV=500μV/m within a distance of m from the wave source.

If you give an example of spectrum masking (spectre mask) in an outdoor environment, from 3.16GHz to 4.75GHz, the bandpass of wireless equipment is used as a reference (0dB), and it is stipulated that it is less than -20dB at 3.1GHz and less than -20dB at 1.61GHz. 30dB. In addition, under actual use conditions, it is necessary to prevent interference with W-LAN (IEEE802.11.a/b/g), and attenuation characteristics are required at 2.48GHz and 5.15GHz.

As described above, it can be seen that in UWB wireless communication equipment, a bandpass filter inserted into a transmission path of a transmission/reception signal is required to have a wide frequency band (a bandwidth ratio of 40% or more), low loss, and high attenuation.

Conventionally, in narrow frequency bands, SAW filters or BAW filters based on crystal or piezoelectric ceramics capable of obtaining a high Q value have been used as low-loss and high-attenuation bandpass filters. These bandwidth ratios are 3 to 4% or less when the center frequency is 2 GHz, and if the passband is 0.06 to 0.08 GHz, it is two orders of magnitude narrower than the UWB frequency bandwidth. The frequency bandwidth of these materials is determined by the electromechanical coupling coefficient of the crystal or the piezoelectric substrate, and it is difficult to expand the frequency bandwidth to form a wide-band bandpass filter from the material point of view.

Therefore, generally in the frequency band of 2 to 5 GHz, as a method of obtaining a bandpass filter having a steep attenuation characteristic, a dielectric filter using a combination of a plurality of dielectric resonators with excellent Q values is known. However, in the case of a dielectric filter having an attenuation characteristic of less than -30dB under the conditions of a center frequency of 3.98GHz, a passband of 1.6GHz, and a W-LAN of 2.48GHz and 5.15GHz, the size is about 10×3×1.5 mm, so there is a very big problem. In this way, in the dielectric filter, it is impossible to achieve both broadband and miniaturization.

Contents of the invention

An object of the present invention is to provide a small-sized bandpass filter which has a wide passband in UWB and can obtain a steep attenuation characteristic with a narrow frequency width, and a wireless communication device using the same.

The bandpass filter of the present invention is provided with a plurality of resonators formed on a dielectric layer, where the transmission wavelength of the approximate center frequency of the passband is λ, and the plurality of resonators are substantially equal to each other by the length of each signal transmission direction. It is composed of a conductor pattern of λ/4, and one end of the plurality of resonators is respectively grounded as a ground terminal, and on the dielectric layer, the ground terminals are arranged on the same side and arranged side by side in sequence, and are located at the first position The non-ground end of the resonator is coupled to the input terminal electrode, the non-ground end of the resonator at the last position is coupled to the output terminal electrode, and the resonator at the middle position is electromagnetically coupled between adjacent resonators, so The non-ground end of the resonator at the first position is coupled to the non-ground end of the resonator at the middle position respectively via a capacitor, and the non-ground end of the resonator at the last position is connected to the resonator at the middle position The non-ground terminals of the resonators are respectively coupled via capacitors.

The bandpass filter with this configuration can realize electromagnetic coupling between the resonators located in the middle. Due to this coupling, a wide band of the bandpass filter is possible by appropriately selecting the coupling amount of each resonator. In addition, by preparing a plurality of resonators, the attenuation characteristic can be sharpened.

In addition, in the coupling of the non-ground terminal of the resonator located at the first position and the input terminal electrode, and the coupling of the non-ground terminal of the resonator located at the last position and the output terminal electrode, capacitors may be used. or inductance. At this time, by setting the constant of the element to a predetermined value, strong coupling can be obtained at the time of input and output of signals in the input unit and the output unit, so that the pass loss of the band-pass filter can be reduced.

The shape of the conductor pattern constituting the resonator is substantially rectangular (rectangular).

The resonator can be formed, for example, from a strip line, a microstrip line or a coplanar line.

In addition, it is desirable to ground the non-ground terminals of any or all of the resonators at the middle position via capacitors (such as C7 - C10 shown in FIG. 2 ).

Accordingly, since the length of the resonator is made smaller than 1/4 wavelength, the longitudinal dimension of the bandpass filter can be reduced, and the bandpass filter can be mounted at a higher density.

Generally, in the energy distribution of a resonator with one end grounded, the electric field energy is highest at the non-grounded end with respect to the length direction of the line, and the electric field energy becomes weaker as it goes to the other end that is grounded. On the other hand, the magnetic field energy is highest at the grounded end and becomes weaker as you go toward the non-grounded end. Electric field energy is defined as CV ² /2 (C is capacitance, V is voltage), and magnetic field energy is LI ² /2 (L is inductance, I is current). To shorten the resonator length, just try to get the same amount of energy out of the resonator. Therefore, increasing the capacitance of the non-ground terminal or increasing the inductance of the ground terminal can be cited. Even in the bandpass filter of the present invention, the length of the resonator can be shortened by providing a capacitance at the non-ground end of the resonator. Therefore, miniaturization of the bandpass filter becomes possible.

In addition, if the bandpass filter of the present invention is formed on a dielectric multilayer substrate on which a plurality of dielectric layers are laminated, by using a dielectric with a high dielectric constant, the bandpass filter can be miniaturized and reduced in height. .

Furthermore, if the resonators are sandwiched between the upper and lower sides by the ground electrodes formed on the dielectric layer, grounding can be easily obtained from the ground electrodes to the ground terminals of the plurality of resonators. Furthermore, an electromagnetic shielding (shield) effect can also be obtained by the ground electrodes sandwiched between the upper and lower sides.

It is desirable that the interval between the upper and lower ground electrodes is 1.0 mm or less. Thus, the thickness of the dielectric multilayer substrate can be reduced.

The number of the plurality of resonators may be set to six, for example.

Generally, since losses exist in resonators, increasing the number of resonators increases the loss in the passband. For example, in the case of a bandpass filter using a Chebyshev function with 3.16GHz to 4.75GHz as the passband, when the fluctuation is 0.2dB and the resonator Q=180, the loss in the five-stage resonator configuration is About -1.0dB, but the attenuation is -18dB, which is insufficient. In the 7-stage resonator configuration, the attenuation is about -32dB, and the loss is as large as -1.9dB. It can be confirmed theoretically that in a bandpass filter composed of 6-stage resonators, the loss is -1.6dB and the attenuation is -25dB, and a result that satisfies both aspects is obtained.

Therefore, using 6-stage resonators, the parallel resonance phenomenon formed by the strong capacitive coupling (also called electrical coupling) and weak inductive coupling (also called magnetic coupling) between the first resonator and the second resonator, and the second The parallel resonance phenomenon of strong capacitive coupling and weak inductive coupling between the six resonators and the fifth resonator can form a steep attenuation pole on the low frequency side of the passband.

In addition, by the capacitor (first capacitor) provided between the first resonator and the second resonator, the second resonator, and the inductive coupling between the second resonator and the third resonator, in the high pass band The steep attenuation characteristic can be realized on the frequency side, and the capacitor (second capacitor) arranged between the first resonator and the third resonator, the third resonator, and the inductance between the third resonator and the fourth resonator coupling, another attenuation characteristic that is steep can be achieved on the high frequency side of the passband.

Moreover, the structure of the band-pass filter utilizing the 6-stage resonator of the present invention can become a symmetrical system centered on the third resonator and the fourth resonator, so it is different from the band-pass filter utilizing 5 resonators. Or, compared with a bandpass filter using 7 resonators, there is also an advantage that it is easy to place the circuit into a pattern.

Furthermore, it is preferable that the distances between the non-ground terminals of all the resonators of the first to sixth resonators and the capacitors connected to the ground terminals are substantially equal when viewed from the stacking direction.

According to this configuration, the lengths of the six resonators from the first resonator to the sixth resonator include the lengths of the conductor lines connecting the first capacitor to the fourth capacitor to the first resonator to the sixth resonator, respectively. They are substantially equal, and patterning can be performed without changing the resonant frequencies of the first to sixth resonators. Therefore, the passband generated by the inductive coupling between adjacent resonators from the second resonator to the fifth resonator can be concentrated at 3.16 GHz to 4.75 GHz. In addition, the combination of the second to fifth resonators and the first to fourth capacitors, and the capacitors formed between the input electrode terminal and the output electrode terminal can make the high frequency side The attenuation is extremely concentrated around 5.3GHz. Furthermore, on the low frequency side, since an attenuation pole can be formed near 2.3 GHz, the pass characteristic and attenuation characteristic required for UWB can be realized with high performance. This action can reduce communication quality degradation due to interference between the 2.48 GHz W-LAN and the 5.15 GHz W-LAN.

Here, in the bandpass filter according to the present invention, it is preferable that the ground terminals of the first resonator and the sixth resonator be arranged in the second resonator to the fifth resonator that are arranged in parallel. The position of the ground end of the first resonator is also deviated from the non-ground end side by a predetermined distance, and the portion adjacent to the non-ground end of the first resonator is bent toward the second resonator, and is in line with the sixth resonator A portion adjacent to the non-ground end is bent toward the fifth resonator.

According to this configuration, the distances between the non-ground terminals of all the resonators of the first to sixth resonators and the capacitors (first to fourth capacitors) connected to the non-ground terminals can be minimized, and the frequency band can be minimized. Frequency adjustment and frequency adjustment of the attenuation pole become easy. Moreover, since the inductive coupling between the first resonator and the second resonator, and the inductive coupling between the fifth resonator and the sixth resonator are weak, the lengths of the first resonator and the sixth resonator are not changed, even to Even if the non-ground terminal side is shifted from the ground terminal, it does not have a great influence on the characteristics of the bandpass filter.

In addition, in the bandpass filter of the present invention, it is preferable that the portion adjacent to the non-ground end of the second resonator bends toward the first resonator, and the portion adjacent to the non-ground end of the fifth resonator bends toward the first resonator. The sixth resonator is curved. Furthermore, it is preferable that a portion adjacent to the non-ground end of the third resonator bends toward the first resonator, and a portion adjacent to the non-ground end of the fourth resonator bends toward the sixth resonator. .

Thereby, the arrangement of each capacitor (the first capacitor to the fourth capacitor) can be adjusted arbitrarily, and the characteristic control of the bandpass filter becomes easy. In addition, a first capacitor may be formed between the first resonator and the second resonator, a second capacitor may be formed between the first resonator and the third resonator, and a capacitor may be formed between the fourth resonator and the sixth resonator. The third capacitor and the fourth capacitor formed between the fifth resonator and the sixth resonator can realize miniaturization of the bandpass filter.

In addition, in the bandpass filter of the present invention, it is preferable that the capacitors form capacitances in the lamination direction by conductive patterns arranged to face each other on different dielectric layers, and are different from the first resonator. A first capacitor is formed between the conductor pattern connected to the ground terminal and the conductor pattern connected to the non-ground terminal of the second resonator, and the conductor pattern connected to the non-ground terminal of the first resonator and the conductor pattern connected to the second resonator A second capacitor is formed between the conductor patterns connected to the non-ground terminals of the three resonators, the conductor pattern connected to the non-ground terminal of the sixth resonator and the conductor pattern connected to the non-ground terminal of the fourth resonator A third capacitor is formed therebetween, and a fourth capacitor is formed between the conductor pattern connected to the non-ground terminal of the sixth resonator and the conductor pattern connected to the non-ground terminal of the fifth resonator.

By forming the first capacitor to the fourth capacitor on a layer different from the layers constituting the first to sixth resonators, the occurrence of inductive coupling between the capacitor and the resonator can be suppressed, and good characteristics can be obtained. In addition, since the first to fourth capacitors can be formed in multiple layers by interposing via-hole conductors, arbitrary electrostatic capacitances can be formed, and the control of the pass-band gate and attenuation poles of the band-pass filter becomes variable. easy.

In particular, in the bandpass filter of the present invention, it is preferable that the conductor pattern connected to the non-ground terminal of the first resonator be arranged above and below the conductor pattern connected to the non-ground terminal of the first resonator, so that forming the first capacitor, and disposing a conductor pattern connected to the non-ground terminal of the third resonator above and below a conductor pattern connected to the non-ground terminal of the first resonator to form the second capacitor; The conductor pattern connected to the non-ground terminal of the sixth resonator is arranged above and below the conductor pattern connected to the non-ground terminal of the fourth resonator to form the third capacitor, and the conductor pattern connected to the non-ground terminal of the sixth resonator The conductor pattern connected to the non-ground terminal of the fifth resonator is arranged above and below the conductor pattern connected to the non-ground terminal of the fifth resonator to form the fourth capacitor.

Thereby, the coupling from the second resonator to the fifth resonator can be enhanced, and realization of a wide band becomes easy.

Furthermore, the bandpass filter of the present invention preferably has an upper ground electrode and a lower ground electrode so as to sandwich the first to sixth resonators and the first to sixth capacitors up and down from the stacking direction. Fourth capacitor.

By being sandwiched between the upper and lower sides by the ground electrode, inductive coupling with external noise can be prevented, and a band-pass filter with a strong structure can be realized in which the band-pass filter does not become a source of interference to the outside.

Furthermore, in the bandpass filter of the present invention, it is preferable to perform capacitive coupling by connecting the input terminal electrode and the output terminal electrode via a capacitor. By capacitively coupling the input terminal electrode and the output terminal electrode through the capacitor, the signal passing through the capacitor and the signal passing through the input capacitor to the first to sixth resonators, the first to fourth capacitors, and the input/output capacitor In the frequencies where the phases of the signals of the constituted circuit are 180 degrees, the respective signals cancel each other to form an attenuation pole. By this action, the attenuation pole on the low-frequency side can be shifted to the band side, and a part of the attenuation pole on the high-frequency side can be shifted to the band side, whereby steeper attenuation characteristics can be obtained.

Specifically, it is preferable that the capacitor forms a capacitance in the lamination direction by being arranged as conductor patterns facing each other on different dielectric layers, and the layer with the conductor pattern connected to the input terminal electrode and the layer with the conductor pattern connected to the input terminal electrode are preferably formed with Independent conductor patterns are provided on different layers of the conductor patterns to which the output terminal electrodes are connected.

In this way, by making the independent conductor pattern and the conductor pattern connected to the input terminal electrode and the conductor pattern connected to the output terminal electrode face each other, the series connection of the capacitance generated between them is realized. Conductor pattern, so it can be made into a simple bandpass filter with a structure that is resistant to lamination deviation.

Furthermore, the present invention is a wireless communication device including the bandpass filter. Accordingly, it is possible to achieve improvement of receiving sensitivity, broadband communication, low power consumption, and prevention of mutual interference with other wireless communication devices such as wireless LAN.

The above-mentioned or other advantages, features, and effects of the present invention will become more apparent from the description of the embodiments described below with reference to the accompanying drawings.

Description of drawings

FIG. 1 is a diagram showing an equivalent circuit of a bandpass filter according to an embodiment of the present invention;

2 is a diagram showing an equivalent circuit of a bandpass filter according to another embodiment of the present invention;

3 is a perspective view of the bandpass filter shown in FIG. 2 as viewed from the stacking direction, wherein the conductor patterns formed on different layers of the stack composed of multiple dielectric layers are superimposed;

4A to FIG. 4H are explanatory diagrams of the state in which the band-pass filter shown in FIG. 3 is developed and viewed from above in each layer of the dielectric layer, showing from the surface layer to the eighth layer;

5A to 5E are explanatory diagrams of the state in which the band-pass filter shown in FIG. 3 is developed on each layer of the dielectric layer and viewed from above, showing layers from the ninth to the twelfth and the back;

6 is a diagram showing an equivalent circuit of a bandpass filter according to another embodiment of the present invention;

7 is a perspective view of the bandpass filter shown in FIG. 6 as viewed from the stacking direction, wherein the conductor patterns formed on different layers of the stack composed of multiple dielectric layers are superimposed;

8A to 8E are explanatory diagrams of the state in which the bandpass filter shown in FIG. 7 is developed and viewed from above on each layer of the dielectric layer, showing the ninth layer to the twelfth layer and the back surface;

9 is a block diagram showing a configuration example of a wireless communication device equipped with a bandpass filter of the present invention;

Fig. 10 is a graph showing the pass characteristic and reflection characteristic of the embodiment of the bandpass filter shown in Fig. 2;

Fig. 11 is a graph showing the pass characteristic and reflection characteristic of the embodiment of the bandpass filter shown in Fig. 6;

Fig. 12 is a graph showing the relationship between the dielectric thickness of the bandpass filter and the maximum value of the loss in the passband;

13 is a graph showing the relationship between the high-frequency conductivity (in terms of Q) and the loss in the passband of electrodes constituting the bandpass filter;

FIG. 14 is a diagram showing an equivalent circuit of a sample in which the relationship between the distance between resonators and the coupling coefficient was measured;

Fig. 15 is a graph showing the results of measuring the relationship between the distance between resonators and the coupling coefficient;

16 is a diagram showing an equivalent circuit of a sample simulating the relationship between the capacitance of a resonator and the coupling coefficient;

FIG. 17 is a graph showing the result of simulating the relationship between the capacitance of a resonator and the coupling coefficient;

Fig. 18 is an equivalent circuit diagram around the resonant frequency of a resonator grounded at one end;

FIG. 19 is a diagram showing reactance around a resonant frequency of a resonator with one end grounded.

Detailed ways

Embodiments of the present invention will be described below with reference to the drawings.

FIG. 1 is a diagram showing a circuit configuration of a bandpass filter according to an embodiment of the present invention.

The bandpass filter includes six resonators 1 to 6 (respectively, a first resonator, a second resonator, a resonator 3, a fourth resonator, a fifth resonator, and a sixth resonator) stacked up and down. These resonators 1 to 6 are in the shape of rectangular conductor plates and are formed of strip lines, microstrip lines or coplanar lines.

The non-ground terminals of resonator 1 and resonator 2 are connected via capacitor C1 (corresponding to the first capacitor), and the non-ground terminals of resonator 1 and resonator 3 are connected via capacitor C2 (corresponding to the second capacitor), The resonator 6 is connected to the non-ground terminal of the resonator 4 via a capacitor C3 (corresponding to a third capacitor), and the resonator 6 is connected to the non-ground terminal of the resonator 5 via a capacitor C4 (corresponding to a fourth capacitor).

Assuming that λ is the propagation wavelength inside the dielectric layer at the approximate center frequency of the passband, basically all the lengths of the six resonators 1 to 6 described above are λ/4.

Among the six resonators, at least four resonators 2 to 5 are arranged in parallel on the same dielectric plane.

However, instead of being on the same dielectric surface, they may be arranged to overlap each other when viewed from the stacking direction.

With this arrangement, the four resonators 2 to 5 are mutually electromagnetically coupled, and in particular, the inductive coupling is enhanced (indicated by M in FIG. 1 ).

All of the ends located on one side (lower side in FIG. 1 ) of the six resonators 1 to 6 are grounded (referred to as ground ends).

The non-ground ends of the resonators 1 and 6 are capacitively coupled to the input electrode IN and the output electrode OUT via input and output capacitors C5 and C6, respectively. These capacitively coupled parts are called "input part" and "output part".

The input and output capacitors C5 and C6 constituting the input unit and the output unit may be lumped constant circuits or distributed constant circuits.

According to this structure, the inductive coupling M of the resonators 2 to 5 is enhanced, the coupling coefficient is increased, and the passband can be widened.

In addition, by arranging the four resonators 2 to 5 facing each other, it is also possible to reduce the size of the bandpass filter.

It is desirable that the electrostatic capacitance of the input/output capacitors C5 and C6 is not less than 0.5 pF and less than 1.5 pF.

In the conventional bandpass filter, since the passband is relatively narrow, it is desired that the circuit Q and Qe indicating the steepness of the circuit have high values. Therefore, when the input/output load and the filter circuit are coupled by capacitance, since Qe is a function of the reciprocal of capacitance, it becomes a small capacitance of 0.1 pF or less.

On the other hand, in the bandpass filter of the present invention, since a bandwidth of about 1.5 GHz or more is required, Qe is desirably small. Therefore, as the above-mentioned capacitance, a large capacitance of 0.5 pF or more is required.

In addition, when the capacitance of the above capacitor is made too large, the frequency band becomes wider, but the steepness of attenuation is lost. Since the bandpass filter used in UWB requires steep attenuation characteristics in a narrow frequency band of 0.4 GHz to 0.6 GHz, even if the capacitance of the capacitor is too large, it is not suitable from the viewpoint of attenuation characteristics. Therefore, less than 1.5pF is desired.

The dielectric constant of the above-mentioned dielectric is preferably set to 10 or less at 3.1 GHz to 10.6 GHz of UWB. Generally, a resonator near the resonance frequency can be equivalently represented by a parallel circuit of an equivalent inductance Lp and an equivalent capacitor Cp as shown in FIG. 18 . The Q of the resonator at this time is proportional to the frequency ω and the capacitance of the equivalent capacitor Cp. When a dielectric with a high dielectric constant is used, the equivalent capacitor Cp becomes large, and the Q of the resonator increases. The high Q of a resonator means that the passband of the resonator is narrowed, and thus the passband of a bandpass filter using a resonator with a high Q is narrowed. When this fact is shown in FIG. 19 , it can be seen that when the resonance frequency is constant, the passband becomes wider as Cp becomes smaller. Therefore, the dielectric constant is desirably 10 or less.

Fig. 2 shows the structure: on the basis of the composition of Fig. 1, the non-ground ends of the resonators 2 to 5 are capacitively coupled to the input electrode IN and the output electrode OUT through the capacitors C1 to C4 respectively, and the capacitors C7-C10 ground the non-ground ends of resonators 2-5.

That is, the non-ground terminal of resonator 2 is grounded via capacitor C7, the non-ground terminal of resonator 3 is grounded via capacitor C8, the non-ground terminal of resonator 4 is grounded via capacitor C9, and the non-ground terminal of resonator 5 is grounded via capacitor C10.

Furthermore, the capacitors C7 to C10 may be lumped constants or distributed constants.

Comparing this configuration with that in FIG. 1 , the non-ground terminals of the resonators 2 to 5 are grounded via the capacitors C7 to C10 . Accordingly, part of the effective length of the resonators 2 to 5 is replaced by the capacitors C7 to C10, and the length of the resonators 2 to 5 can be made smaller than 1/4 wavelength.

Therefore, in this bandpass filter, the lengths of the resonators 2 to 5 can be shortened, which is more advantageous in terms of miniaturization.

FIG. 3 is a diagram showing a configuration example of the bandpass filter shown in FIG. 2 . This figure is a perspective view of a plurality of dielectric layers viewed from the stacking direction, and conductor patterns formed on different dielectric layers are superimposed and illustrated.

In addition, FIGS. 4A to 4H and FIGS. 5A to 5E are explanatory diagrams for developing the bandpass filter shown in FIG. 3 in each layer of the dielectric layer. ~ Figure 5E shows from the ninth floor to the twelfth floor and the back.

This bandpass filter is constituted by the following structure, for example, in a stack of multilayer dielectric layers 17 with a dielectric constant of about 5.0 to 60 and a thickness of 0.03 to 0.1 mm, via conductors penetrating through each dielectric layer are included or formed in conductor pattern on each dielectric layer 17 .

In this example, as shown in FIGS. 4A to 4H and FIGS. 5A to 5E , there are 12 dielectric layers.

On the surface of the laminate (the surface dielectric layer), input terminal electrodes 13 and output terminal electrodes 15 are provided, and ground patterns 14 as upper ground electrodes are provided ( FIG. 4A ). On the other hand, a ground pattern 16 as a lower ground electrode is provided on the back surface of the laminate ( FIG. 5E ).

In addition, one end is grounded respectively, and if the propagation wavelength inside the dielectric layer at the approximate center frequency of the passband is λ, the length in the signal transmission direction is basically λ/4 length of the conductor pattern. The resonators (resonator 1, resonator 2, resonator 3, resonator 4, resonator 5, resonator 6) are formed on the same dielectric layer inside the stack (on the dielectric layer of layer 7) (Fig. 4G ).

The six resonators have their respective ground terminals arranged in the same direction when viewed from the lamination direction, and resonators 1 to 6 are arranged side by side in sequence. That is, resonator 1, resonator 2, resonator 3, resonator 4, resonator 5, and resonator 6 are arranged in order.

Furthermore, the length in the signal transmission direction is "basically λ/4", which means that the length is shorter than λ/4 by changing the capacitance of the ground plane of the non-ground terminal.

One end (ground end) of the resonator 1 is connected to the ground pattern 14 formed on the surface of the laminate and the ground pattern 16 formed on the back surface of the laminate through via-hole conductors 28 . In addition, the non-ground end of the resonator 1 is connected to the input terminal electrode 13 via the input capacitor C5.

Specifically, the non-ground end of the resonator 1 is connected to the conductive pattern 11 formed on the seventh dielectric layer via the conductor line 26, and the conductive pattern 11 on the seventh dielectric layer is connected to the ninth via conductor. The conductor pattern 11 on the layer dielectric layer is connected.

On the other hand, the input terminal electrodes 13 are connected to the conductor patterns 11 on the dielectric layers of the sixth, eighth, and tenth layers via via conductors. Since the conductive pattern 11 formed on each layer overlaps when viewed from the stacking direction, it functions as an input capacitor C5 that forms capacitance in the stacking direction.

One end (ground end) of the resonator 6 is connected to the ground pattern 14 formed on the surface of the laminated body and the ground pattern 16 formed on the back surface of the laminated body via the via-hole conductor 30 .

In addition, the non-ground end of the resonator 6 is connected to the output terminal electrode 15 via the output capacitor C6. Specifically, the non-ground end of the resonator 6 is connected to the conductive pattern 12 formed on the seventh layer through the conductive line 27, and the conductive pattern 12 on the seventh layer dielectric layer is connected to the ninth layer dielectric layer through the via conductor. The conductor patterns 12 on the layers are connected.

In addition, the output terminal electrodes 15 are connected to the conductor patterns 12 on the dielectric layers of the sixth layer, the eighth layer, and the tenth layer through via conductors. Since the conductive patterns 12 formed on the respective layers overlap each other when viewed from the stacking direction, they function as output capacitors C6 that form capacitance in the stacking direction.

In the formation of the input capacitor C5 and the output capacitor C6, various structures that form capacitance in the stacking direction can be adopted, but it is preferable to arrange the conductor pattern connected to the input terminal electrode 13 and the output terminal electrode 15 at the top as in this embodiment. Next, to form the structure of the capacitor. In addition, it is not limited to a capacitor, but may be connected via an inductor.

One ends (ground ends) of the resonators 2 to 5 are connected to each other, and are connected to a ground pattern formed on the back surface of the laminate through a via-hole conductor 29 .

In addition, the interval between resonators 2 to 5 is arranged so that adjacent resonators are inductively coupled as the main coupling interval, while between resonator 1 and resonator 2 and between resonator 5 and resonator The intervals between 6 are wider than the intervals between adjacent resonators from resonator 2 to resonator 5, and the coupling between them is weak inductive coupling.

Furthermore, the non-ground terminals of the resonator 1 and the resonator 2 are connected through the capacitor C1 to be capacitively coupled.

Specifically, the non-ground end of the resonator 1 is connected to the conductive line 18 on the dielectric layer of the fifth layer and the third layer through a via conductor, and the conductive line 18 is connected to the conductive pattern 7 constituting the capacitor C1.

On the other hand, the non-ground end of the resonator 2 is connected to the conductor lines 19 on the dielectric layers of the second layer, the fourth layer, and the sixth layer through via conductors, and the conductor lines 19 are connected to the conductor pattern constituting the capacitor C1. 7 connections.

In this way, the capacitor C1 is preferably provided with a capacitance in the stacking direction by being provided as conductive patterns facing each other on different dielectric layers. In this example, it becomes the following structure: the third The conductor patterns 7 on the dielectric layer are arranged up and down, the conductor patterns 7 on the second layer and the fourth dielectric layer connected to the non-ground end of the resonator 2 are arranged, and the conductor patterns 7 on the fifth layer connected to the non-ground end of the resonator 1 are arranged. Conductive patterns 7 on the fourth and sixth dielectric layers connected to the non-ground terminals of the resonator 2 are placed above and below the conductive patterns 7 on the dielectric layer.

Similarly, the non-ground terminals of the resonator 1 and the resonator 3 are capacitively coupled by being connected via the capacitor C2.

Specifically, the non-ground end of the resonator 1 is connected to the conductor lines 20 on the dielectric layers of the ninth layer and the eleventh layer through via conductors, and the conductor lines 20 are further connected to the conductor pattern 8 constituting the capacitor C2.

On the other hand, the non-ground end of the resonator 3 is connected to the conductor line 21 on the dielectric layer of the 8th layer, the 10th layer, and the 12th layer through a via conductor, and the conductor line 21 is connected to the conductor pattern constituting the capacitor C2. 8 connections.

In this way, the capacitor C2 is preferably formed with capacitance in the stacking direction by being arranged as conductive patterns facing each other on different dielectric layers. The conductor patterns 8 on the dielectric layer are arranged up and down, the conductor patterns 8 on the 8th layer connected to the non-ground end of the resonator 3 and the conductor patterns 8 on the 10th layer dielectric layer are arranged, and the conductor patterns 8 on the 11th layer connected to the non-ground end of the resonator 1 Conductor patterns 8 on the first and second dielectric layers are placed above and below the conductive patterns 8 on the tenth and twelfth dielectric layers connected to the non-ground terminals of the resonator 3 .

The non-ground terminals of the resonator 6 and the resonator 4 are connected through a capacitor C3 to be capacitively coupled.

Specifically, the non-ground end of the resonator 6 is connected to the conductor lines 22 on the dielectric layers of the ninth and eleventh layers through via conductors, and the conductor lines 22 are further connected to the conductor pattern 9 constituting the capacitor C3.

On the other hand, the non-ground end of the resonator 4 is connected to the conductor line 23 on the dielectric layer of the 8th layer, the 10th layer, and the 12th layer through a via conductor, and the conductor line 23 is connected to the conductor pattern constituting the capacitor C3. 9 connections.

In this way, the capacitor C3 is preferably formed with capacitance in the lamination direction by being arranged as opposite conductor patterns on different dielectric layers, and in this example, it becomes a structure: on the ninth layer connected to the non-ground terminal of the resonator 6 The conductive pattern 9 on the dielectric layer is placed above and below the conductor pattern 9 on the 8th layer connected to the non-ground end of the resonator 4 and the conductive pattern 9 on the 10th layer of the dielectric layer, and on the 11th layer connected to the non-ground end of the resonator 6 The conductive patterns 9 on the dielectric layer are placed above and below the conductive patterns 9 on the tenth and twelfth dielectric layers connected to the non-ground terminal of the resonator 4 .

The non-ground terminals of the resonator 6 and the resonator 5 are connected through a capacitor C4 to be capacitively coupled.

Specifically, the non-ground end of the resonator 6 is connected to the conductive line 24 on the dielectric layer of the fifth layer and the third layer through a via conductor, and the conductive line 24 is connected to the conductive pattern 10 constituting the capacitor C4.

On the other hand, the non-ground end of the resonator 5 is connected to the conductor line 25 on the dielectric layer of the second layer, the fourth layer, and the sixth layer through a via conductor, and the conductor line 25 is connected to the conductor pattern constituting the capacitor C4. 10 connections.

In this way, the capacitor C4 is preferably formed with capacitance in the stacking direction by being arranged as conductive patterns facing each other on different dielectric layers. In this example, it becomes the following structure: the third The conductor patterns 10 on the dielectric layer are arranged up and down, the conductor patterns 10 on the second layer and the fourth dielectric layer connected to the non-ground end of the resonator 5 are arranged, and the conductor patterns 10 on the fifth layer connected to the non-ground end of the resonator 6 are arranged. Above and below the conductive patterns 10 on the dielectric layer, the conductive patterns 10 on the fourth and sixth dielectric layers connected to the non-ground terminal of the resonator 5 are arranged.

In this way, capacitors C1 to C4 have a structure in which two conductors connected to the non-ground ends of resonator 1 or resonator 6 are sandwiched from top to bottom by three conductor patterns connected to the non-ground ends of resonators 2 to 5. Patterns, that is, the conductor patterns connected to the non-ground terminals of the resonators 2 to 5 are arranged on the uppermost and the lowermost, and overlapped when viewed from the lamination direction, so that more capacitance can be formed between these conductor patterns, and The effect of forming the capacitors of the resonators 2 to 5 in the relationship between the ground pattern 14 and the ground pattern 16 can be obtained. In addition, the number of laminations is not particularly limited, and can be appropriately set.

Furthermore, the ground terminals of the resonators 2 to 5 are arranged in substantially the same row (on a line perpendicular to the long axis) with respect to the longitudinal direction, and the ground terminals of the resonator 1 and the resonator 6 are arranged on a row that is smaller than those arranged side by side. The positions of the ground terminals of the resonators 2 to 5 are also shifted by a predetermined distance to the non-ground terminals.

In addition, a portion of the resonator 1 adjacent to the non-ground terminal is bent toward the resonator 2 , and a portion of the resonator 6 adjacent to the non-ground end is bent toward the resonator 5 .

Furthermore, in this embodiment, the part of the resonator 2 that is adjacent to the non-ground end is bent toward the resonator 1, and the part of the resonator 5 that is adjacent to the non-ground end is bent toward the resonator 6, and the part of the resonator 3 that is adjacent to the non-ground end is bent toward the resonator 6. The portion adjacent to the non-ground terminal is bent toward the resonator 1 , and the portion of the resonator 4 adjacent to the non-ground end is curved toward the resonator 6 .

By adopting such a structure, the distances between the non-ground ends of all resonators of resonators 1 to 6 and the capacitors connected to the non-ground ends are substantially equal when viewed from the stacking direction. That is, the conductor lines 18 , 19 , 20 , 21 , 22 , 23 , 24 , and 25 are substantially equal in length.

Here, since these distances are approximately equal when viewed from the stacking direction, the lengths of the conductor lines connecting the capacitors C1 to C4 from the resonator 1 to the resonator 6 are approximately equal, and there is no need to change the resonator 1 to the resonator 6. Resonant frequency, that is, the effect of patterning.

In addition, the term "non-ground end" refers to the pattern region from the tip that becomes the non-ground side of the resonator to the ground side of 200 μm. In addition, the term that the lengths are substantially equal means that the difference between the maximum length and the minimum length of the lengths of the respective conductor lines is 100 μm or less.

In addition, if the lengths of all the conductor lines are approximately equal, the resonators 2 to 5 may not be bent, but by bending them in this way, the arrangement of each of the capacitors C1 to C4 can be adjusted arbitrarily, and the characteristic control of the bandpass filter can be changed. easy.

In addition, capacitor C1 may be formed between resonator 1 and resonator 2, capacitor C2 may be formed between resonator 1 and resonator 3, capacitor C3 may be formed between resonator 4 and resonator 6, and capacitor C3 may be formed between resonator 5 and resonator 6. A capacitor C4 is formed between the resonators 6, and the bandpass filter can be miniaturized.

In addition, in this embodiment, resonators 1 to 6 and capacitors C1 to C4 are formed in a region sandwiched between ground pattern 14 serving as an upper ground electrode and ground pattern 16 serving as a lower ground electrode. middle.

In this way, by sandwiching between the upper and lower ground electrodes, inductive coupling with noise from the outside can be prevented, and furthermore, there is an effect that the bandpass filter does not become a source of interference to the outside.

In addition, the capacitors C1 to C4 in this embodiment are formed so that the conductor pattern on the side connected to the non-ground end of the resonators 2 to 5 faces the ground pattern 14 and the ground pattern 16, and can also be used as The electrodes of the capacitor are connected in parallel between the ground, thus achieving the simplification of the conductor pattern.

According to this structure, the coupling from the resonator 2 to the resonator 5 can be enhanced, and realization of a wide band becomes easy. The reasons are stated below.

The passband of the bandpass filter is determined by the magnitude of the coupling coefficient between resonators. According to theoretical calculations using the Chebyshev function (Chebyshev function), in the case of fabricating a bandpass filter with a passband of 3.1 to 4.9 GHz, the coupling coefficient needs to be 0.4. The coupling coefficient can be controlled according to the interval between resonators arranged in the same layer, and the coupling coefficient can be improved by narrowing the interval.

An equivalent circuit is shown in FIG. 14 . On a ceramic substrate with a dielectric constant of 9.4 and a thickness of 0.9 mm, two λ/4 strip transmission line resonators 31 and 32 with a width of 0.1 mm and a length of 3.2 mm are fabricated on the same layer, and the distance between the resonators is d The change in the coupling coefficient when changing in the range of 0.075 mm to 0.125 mm was measured. Furthermore, two stripline resonators may be weakly inductively coupled with respect to the input and output electrodes.

As a result, as shown in FIG. 15 , it can be seen that when the interval between the resonators is 0.075 mm, the coupling coefficient can be obtained only about 0.04 at most. In order to enhance coupling, it has been proposed to set the interval d between resonators to be less than 0.075 mm. However, when the interval d between resonators is narrowed, there is a problem that the accuracy of the interval becomes stricter in terms of manufacturing.

On the other hand, as another method of enhancing the coupling, it is proposed to increase the capacitance with respect to the ground plane of the non-ground end of the resonator. When the capacitance is increased, the electric field component of the single resonator is concentrated on the ground plane via the capacitor, and the coupling by the magnetic field between the resonators is strengthened, thereby increasing the coupling coefficient.

Using Ansoft's electromagnetic field simulator HFSS, the change in coupling coefficient when the capacitor corresponding to the ground is changed for the λ/4 strip transmission line resonator arranged in the same layer is simulated by eigenvalue analysis. An equivalent circuit is shown in FIG. 16 .

A capacitor C13 and a capacitor C14 are respectively connected to the non-ground terminals of the resonator 3 and the resonator 4 . As simulation conditions, the interval between resonators having a dielectric constant of 9.4, a thickness of 0.9 mm, a resonator width of 0.1 mm, and a resonator length of 3.2 mm was set to 0.1 mm. Here, the capacitors C13 and C14 are calculated by the capacitance calculation formula of a parallel plate obtained by using the electrode area and the distance between the electrode surface and the GND surface.

As shown in FIG. 17 , as a result, the coupling coefficient can be increased by increasing the capacitors C13 and C14 , and it can be seen that a coupling coefficient of 0.4 can be obtained with a capacitor of about 0.2 pF.

In the present invention, capacitors C7 to C10 are connected to resonators 2 to 5 as shown in FIG. The capacitor C8 of the resonator 3 is formed between the conductor pattern constituting the capacitor C2 and the ground, the capacitor C9 of the resonator 4 is formed between the conductor pattern constituting the capacitor C3 and the ground, and the capacitor C9 of the resonator 4 is formed between the conductor pattern constituting the capacitor C4 and the ground. Capacitor C10 of resonator 5 .

Therefore, it is not necessarily necessary to specially add the capacitors C7 to C10 as chip components, which facilitates fabrication of the filter.

Furthermore, this configuration prevents the capacitors C1 and C2 connected to the resonator 1 and the capacitors C3 and C4 connected to the resonator 6 from being inductively coupled to other electrode patterns.

Next, the equivalent circuit shown in FIG. 6 performs capacitive coupling by connecting the input terminal IN and the output terminal OUT via the input-output capacitor C11 in addition to the configuration of FIG. 2 . This circuit is connected between the input terminal IN and the output terminal OUT of the equivalent circuit shown in FIG. 2 via the input/output capacitor C11.

As a function of the structure of FIG. 6, the signal passing through the circuit formed from the input capacitor C5 to resonator 1 to resonator 6, interstage capacitor C1 to interstage capacitor C4, and output capacitor C6, and the signal passing through the input and output capacitor At frequencies where the phases of the signals of C11 are 180 degrees out of phase, the signals cancel each other out and an attenuation pole can be formed.

The attenuation pole on the low frequency side generated by the inductive coupling M of the resonator 1 and the resonator 2 and the parallel resonance phenomenon of the capacitor C1 can be made to operate near the passband, and the inductance of the capacitor C1 and the resonator 2 and the resonator 3 can be made The attenuation pole on the high frequency side formed by coupling the resonance phenomenon of M and resonator 2 and the inductive coupling of capacitor C2 and resonator 3 and resonator 4 to the resonance phenomenon of M and resonator 3 operates near the passband. Therefore, a steeper decay characteristic can be obtained.

The input/output capacitor C11 forms capacitance in the stacking direction by being provided as conductor patterns facing each other on different dielectric layers. Specifically, by providing an independent conductor pattern in a different layer from the layer provided with the conductor pattern connected to the input terminal electrode and the layer provided with the conductor pattern connected to the output terminal electrode, the independent conductor pattern and the input The conductor pattern connected to the terminal electrode and the conductor pattern connected to the output terminal electrode face each other, and a series connection of capacitance generated therebetween is realized.

An example of this structure is shown in FIG. 7 . Fig. 7 is a perspective view viewed from the stacking direction, illustrating conductor patterns formed superimposed on different layers of a stack composed of a plurality of dielectric layers.

In addition, FIGS. 8A to 8E are explanatory diagrams in which the bandpass filter shown in FIG. 7 is developed for each layer of the dielectric layer, showing the ninth to twelfth layers and the back surface. Furthermore, since the first to eighth layers of the bandpass filter shown in FIG. 7 are the same as those shown in FIGS. 4A to 4H , description thereof will be omitted.

As shown in FIGS. 8A to 8E , conductive pattern 99 is arranged on the eleventh layer so as to be capacitively coupled to conductive pattern 11 on the tenth layer and conductive pattern 12 on the tenth layer. In addition, the aforementioned independent conductor pattern means that it is not electrically connected to other conductor patterns like this conductor pattern 99 .

The conductor pattern 11 of the 10th layer is connected to the input terminal electrode 13 via the via-hole conductor, and the conductor pattern 12 of the 10th layer is connected to the output terminal electrode 15 via the via-hole conductor, so it is equivalent to passing through the conductor pattern 99 of the 11th layer, A capacitor is used to couple between the input terminal electrode 13 and the output terminal electrode 15 . The capacitance at this time is a series connection of the capacitance formed by the conductive pattern 11 and the conductive pattern 99 and the capacitance formed by the conductive pattern 12 and the conductive pattern 99 .

Next, a method of manufacturing the bandpass filter described in FIGS. 1 to 8 will be described.

The bandpass filter has a structure in which the above-mentioned resonator is formed on each dielectric layer of a dielectric multilayer substrate in which a plurality of dielectric layers are laminated.

A dielectric multilayer substrate is formed by stacking a plurality of dielectric layers of the same size and shape, and a conductor layer composed of a predetermined conductor pattern is formed on each dielectric layer.

Each dielectric layer, for example, the dielectric layer is formed of low temperature sintered ceramics (LTCC: Low Temperature Co-fired Ceramics), and the conductor layer formed on each dielectric layer is formed of a low-resistance conductor such as copper or silver.

Such a multilayer substrate is formed by a known multilayer ceramic technology. For example, a conductive paste is applied to the surface of a ceramic green sheet, and conductor patterns constituting each resonator and each capacitor are formed and then laminated. It is formed by thermocompression bonding and sintering at low temperature.

In addition, on each dielectric layer, via conductors necessary for connecting upper and lower conductor layers are appropriately formed over multiple layers.

In addition, the wireless communication device of the present invention is composed of, for example, a baseband IC for processing baseband signals, an RFIC for processing high-frequency signals, a balun for converting balanced signals and unbalanced signals, the above-mentioned band pass A filter, a high-frequency switch for switching transmission and reception, and an antenna are configured. The band-pass filter passes the transmission and reception signals in the UWB frequency band, and attenuates the signals outside the frequency band sharply.

Examples of such wireless communication devices include mobile phones, peripheral storage devices compatible with wireless communication, PC peripherals such as printers and scanners, digital televisions, projectors, digital cameras, digital video cameras, and the like.

Next, a configuration example of a wireless communication device including the above-described bandpass filter is shown in FIG. 9 .

According to Fig. 9, the wireless communication device is composed of a baseband IC45 for processing baseband signals, an RFIC44 for processing high frequency signals, a high frequency switch 41 for switching transceivers, a balanced-unbalanced converter 43 for converting balanced signals and unbalanced signals, a bandpass Filter 42 and antenna constitute.

The RFIC 44 performs frequency conversion and high-frequency amplification of the transmission signal acquired by the baseband IC 45 , and also performs low-noise amplification of the reception signal. The above-mentioned high-frequency switch 41 is a switch for temporally switching the transmission and reception paths.

The band-pass filter 42 is a band-pass filter of the present invention that passes the frequency band of the UWB transmission/reception signal and sharply attenuates signals outside the frequency band. According to the function of this bandpass filter 42, not only can the transmission and reception signals not be attenuated, but also the mutual interference with other systems can be prevented.

(Example)

The pass characteristic S21 and reflection characteristic S11 of the bandpass filter formed by the wiring pattern shown in FIGS.

At this time, ceramics with a dielectric constant of 9.0 were used, the thickness of one dielectric layer was 75 μm, and a 12-layer structure was adopted. At this time, the size of the dielectric is 4.5×3.2mm. The curves of the transmission characteristic S21 and the reflection characteristic S11 are shown in FIG. 10 .

In addition, under the same conditions, the pass characteristics S21 and reflection characteristics S11 of the band-pass filter having the structure of adding a conductive pattern 99 as shown in FIGS. To measure. The result is shown in FIG. 11 .

According to the results shown in FIG. 10, the pass loss in the frequency band of about 1.5 GHz from 3.16 GHz (indicated by ml) to 4.75 GHz (indicated by m2) is less than 1.5 dB. In addition, the attenuation at 2.48 GHz (expressed by m3) of IEEE802.11b/g to which W-LAN is applied can be obtained at 30 dB or more. On the other hand, at 5.15 GHz where IEEE802.11a to which W-LAN is applied, an attenuation characteristic of about 32 dB can be obtained.

In addition, according to the results shown in Figure 11, the pass loss from 3.16GHz (expressed by ml) to 4.75GHz (expressed by m2) is less than 1.5dB, and the attenuation at 2.48GHz (expressed by m3) can be obtained above 30dB, The result is the same as that shown in Fig. 10 . Furthermore, the amount of attenuation at 5.15 to 5.35 GHz is 30 dB or more, which can be improved by 8 dB or more compared with the example shown in FIG. 10 .

Next, using the circuit simulator ADS of Agilent Technologies, under the condition of a ceramic substrate with a dielectric constant of 9.4, the bandpass filter formed by the wiring pattern shown in FIGS. 4A to 4H and FIGS. 5A to 5E was simulated.

Fig. 12 is a graph showing the relationship between the thickness of the dielectric and the maximum value of insertion loss in the passband.

According to FIG. 12 , in a dielectric having a dielectric constant of 9.4 and a dielectric thickness of 0.9 mm, the loss in the passband is 1.44 dB. In addition, when the dielectric thickness is 0.86mm, the insertion loss is 1.5dB or more. When converted to a dielectric thickness of 0.9mm, when the dielectric constant is 9.83, the insertion loss is 1.5dB. Therefore, it can be seen that the dielectric constant of the dielectric used in the band-pass filter of the present invention is desirably 10 or less.

On the other hand, when the dielectric thickness is 1.0 mm, according to FIG. 12 , the loss in the pass band is 1.27 dB, and the pass characteristic is good. This is the same case where the dielectric constant is reduced to 8.46 at a dielectric thickness of 0.9mm. When the thickness of the dielectric is increased, the loss in the passband of the filter becomes smaller. However, in consideration of mounting in mobile phones in recent years, the height of parts is desired to be 1.0 mm or less. Therefore, it is not preferable that the dielectric thickness is larger than 1.0 mm.

As described above, in the bandpass filter of the present invention, it is desirable that the distance between the upper and lower ground planes be 1.0 mm or less.

For the verification described, a resonator with Q=163 was used.

Fig. 13 is a graph showing the relationship between the insertion loss of the bandpass filter of the present invention and the Q of the distributed constant line. It can be seen that by increasing the Q value of the distributed constant line, the loss of the band-pass filter can be reduced. The Q value of a distributed constant line is increased by increasing the conductivity of the line at high frequencies.

When the bandpass filter of the present invention is constructed on the ADS of the circuit simulator to investigate the pass characteristics, when the capacitance of the input capacitor C5 and the output capacitor C6 is 0.8pF, the maximum loss at the frequency from 3.16GHz to 4.75GHz is -1.32dB. On the other hand, when the capacitance of the input capacitor C5 and the output capacitor C6 is set to 0.4pF, there are ripples in the passband and the frequency band is narrowed, and the maximum loss is 1.68dB. On the other hand, when the capacitance of the input capacitor C5 and the output capacitor C6 is set to 1.5 pF, the steepness of attenuation is lost, and the attenuation amount becomes higher below 3.1 GHz. With this phenomenon, the pass characteristic at 3.16 GHz deteriorates to 1.66 dB.

As described above, in the bandpass filter of the present invention, it is desirable that the electrostatic capacitances of the input capacitor C5 and the output capacitor C6 are 0.5 pF or more and less than 1.5 pF.

In addition, the MB-OFDM method, which is one method of UWB, is cited as an example here as the passband. It can also be discussed in the same way. By modulating the length, width, and interval from resonator 1 to resonator 6, and the capacitances of capacitors C1 to C4, the bandpass filter of the present invention can be used even in DS-CDMA-based UWB.

Claims (17)

1. band pass filter, it possesses a plurality of resonators that are formed at dielectric layer,

The transmission wavelength of the approximate centre frequency of passband is made as λ, described a plurality of resonators by each side signal transmission to length be that the conductive pattern of λ/4 constitutes substantially,

One end of described a plurality of resonators is ground connection as earth terminal and respectively, and on described dielectric layer, described earth terminal is disposed at the same side, and in order and establish,

Be positioned at the non-earth terminal and the coupling of input terminal electrode of the described resonator of first position,

Be positioned at the non-earth terminal and the coupling of lead-out terminal electrode of the described resonator of last position,

Be positioned at electromagnetic coupled between the adjacent resonators of resonator in centre position,

The non-earth terminal of the non-earth terminal of the described resonator that is positioned at first position and the described resonator that is positioned at the centre position is respectively via capacitor-coupled, and the non-earth terminal of the non-earth terminal of the described resonator that is positioned at last position and the described resonator that is positioned at the centre position is respectively via capacitor-coupled.

2. band pass filter according to claim 1, wherein,

Constitute described resonator conductive pattern be shaped as rectangle.

3. band pass filter according to claim 1, wherein,

The non-earth terminal of the described resonator that is positioned at the centre position is ground connection via capacitor.

4. band pass filter according to claim 3, wherein,

The length of this resonator is less than 1/4 wavelength.

5. band pass filter according to claim 1, wherein,

On described a plurality of resonator has been formed on the dielectric multilayer substrate of dielectric layer stacked.

6. band pass filter according to claim 5, wherein,

Upside grounding electrode and downside grounding electrode are set, with the described a plurality of resonators of clamping about stacked direction.

7. band pass filter according to claim 5, wherein,

Described input terminal electrode and lead-out terminal electrode are formed on the surface and the back side of the duplexer that is made of a plurality of dielectric layers, and described a plurality of resonators are formed at the inside of described duplexer,

Described a plurality of resonator has 6 from first resonator to sixth resonator,

The non-earth terminal of described first resonator is via capacitor or inductance and be connected with described input terminal electrode,

The non-earth terminal of described sixth resonator is via capacitor or inductance and be connected with described lead-out terminal electrode,

Electromagnetic coupled the adjacent resonators till from described second resonator to described the 5th resonator,

Between the non-earth terminal of described first resonator and described second resonator, between the non-earth terminal of described first resonator and described the 3rd resonator, between the non-earth terminal of described sixth resonator and described the 4th resonator, connect via capacitor respectively between the non-earth terminal of described sixth resonator and described the 5th resonator, and be coupled.

8. band pass filter according to claim 7, wherein,

The distance of the non-earth terminal of whole resonators of described first resonator～described sixth resonator and the capacitor that is connected with this non-earth terminal about equally.

9. band pass filter according to claim 7, wherein,

The earth terminal of described first resonator and described sixth resonator is configured in, than the position of the earth terminal in these and described second resonator～described the 5th resonator established also to non-earth terminal lateral deviation on the position of predetermined distance, and

With the contiguous position of the non-earth terminal of described first resonator to the described second resonator bending, and with the contiguous position of the non-earth terminal of described sixth resonator to described the 5th resonator bending.

10. band pass filter according to claim 7, wherein,

With the contiguous position of the non-earth terminal of described second resonator to the described first resonator bending, and with the contiguous position of the non-earth terminal of described the 5th resonator to described sixth resonator bending.

11. band pass filter according to claim 7, wherein,

With the contiguous position of the non-earth terminal of described the 3rd resonator to the described first resonator bending, and with the contiguous position of the non-earth terminal of described the 4th resonator to described sixth resonator bending.

12. band pass filter according to claim 7, wherein,

Described capacitor forms electric capacity by the conductive pattern that is set to subtend on different dielectric layers on stacked direction,

The conductive pattern that is connected with the non-earth terminal of described first resonator and with conductive pattern that the non-earth terminal of described second resonator is connected between form first capacitor,

The conductive pattern that is connected with the non-earth terminal of described first resonator and with conductive pattern that the non-earth terminal of described the 3rd resonator is connected between form second capacitor,

The conductive pattern that is connected with the non-earth terminal of described sixth resonator and with conductive pattern that the non-earth terminal of described the 4th resonator is connected between form the 3rd capacitor,

The conductive pattern that is connected with the non-earth terminal of described sixth resonator and with conductive pattern that the non-earth terminal of described the 5th resonator is connected between form the 4th capacitor.

13. band pass filter according to claim 12, wherein,

Dispose the conductive pattern that is connected with the non-earth terminal of described second resonator up and down at the conductive pattern that is connected with the non-earth terminal of described first resonator, to form described first capacitor, and dispose the conductive pattern that is connected with the non-earth terminal of described the 3rd resonator up and down at the conductive pattern that is connected with the non-earth terminal of described first resonator, to form described second capacitor;

Dispose the conductive pattern that is connected with the non-earth terminal of described the 4th resonator up and down at the conductive pattern that is connected with the non-earth terminal of described sixth resonator, to form described the 3rd capacitor, dispose the conductive pattern that is connected with the non-earth terminal of described the 5th resonator up and down at the conductive pattern that is connected with the non-earth terminal of described sixth resonator, to form described the 4th capacitor.

14. band pass filter according to claim 12, wherein,

Upside grounding electrode and downside grounding electrode are set, with the described a plurality of resonators of clamping about stacked direction,

Described first capacitor～described the 4th capacitor is formed on the zone by described upside grounding electrode and described downside grounding electrode clamping.

15. band pass filter according to claim 1, wherein,

By being connected via capacitor with described lead-out terminal electrode, described input terminal electrode is coupled.

16. band pass filter according to claim 15, wherein,

Described capacitor forms electric capacity by the conductive pattern that is set to subtend on different dielectric layers on stacked direction,

With the layer that is provided with the conductive pattern that is connected with described input terminal electrode and be provided with the layer of the conductive pattern that is connected with described lead-out terminal electrode different layer on, the independent conductors pattern is set.

17. a Wireless Telecom Equipment,

It possesses antenna, make the described band pass filter of claim 1, RFIC that handles described receiving and transmitting signal that the receiving and transmitting signal by this antenna receiving-sending passes through and the baseband I C that handles baseband signal.

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