HK1071816B - Array antenna communication device - Google Patents

Description

Array antenna communication device

Technical Field

The present invention relates to a radio apparatus structure capable of changing antenna directivity in real time, and more particularly to a radio apparatus structure for an adaptive array radio base station.

Background

In recent years, in a mobile communication system, various transmission channel allocation methods for effectively utilizing frequencies have been proposed, some of which have been applied.

Fig. 3 is a channel configuration diagram in various communication systems such as Frequency Division Multiple Access (FDMA), Time Division Multiple Access (TDMA), and PDMA.

First, FDMA, TDMA, and PDMA will be briefly explained with reference to fig. 3. Fig. 3(a) shows FDMA, in which analog signals of users 1 to 4 are transmitted after being frequency-divided by radio waves having different frequencies f1 to f4, and the signals of the users 1 to 4 are separated by frequency filters.

In the TDMA shown in fig. 3(b), the digitized signals of the users are time-divided and transmitted at predetermined time intervals (time slots) by radio waves having different frequencies f1 to f4, and the signals of the users are separated from the time synchronization between the base station and the mobile terminal devices of the users by a frequency filter.

On the other hand, with the recent spread of portable telephones, the PDMA system has been proposed in order to improve the frequency use efficiency of radio waves. As shown in fig. 3(c), the PDMA scheme spatially divides 1 slot at the same frequency and transmits data of a plurality of users. In this PDMA, the signals of each user are separated by using a frequency filter, time synchronization between the base station and each user mobile terminal apparatus, and a device for removing mutual interference such as an adaptive array (adaptive array).

Fig. 4 is a schematic diagram illustrating the operation principle of the adaptive array radio base station. In fig. 4, 1 adaptive array radio base station 1 includes an array antenna 2 including n antennas #1, #2, #3, # and # n, and a range to which the radio wave reaches is indicated by a1 st shaded area 3. On the other hand, the range where the radio wave of the adjacent other radio base station 6 reaches is indicated by a2 nd shaded area 7.

In the area 3, radio wave signals are transmitted and received between the mobile phone 4 as the terminal of the user a and the adaptive array radio base station 1 (arrow 5). On the other hand, in the area 7, radio wave signals are transmitted and received between the mobile phone 8 as the terminal of the other user B and the radio base station 6 (arrow 9).

Here, when the frequency of the radio wave signal of the mobile phone 4 of the user a and the frequency of the radio wave signal of the mobile phone 8 of the user B are occasionally equal to each other, the radio wave signal from the mobile phone 8 of the user B becomes an unnecessary interference signal in the area 3 depending on the position of the user B, and is mixed into the radio wave signal between the mobile phone 4 of the user a and the adaptive array base station 1.

In this way, in the adaptive array radio base station 1 that receives the mixed radio wave signals from both the users a and B, if no processing is performed, the signals obtained by mixing the signals from both the users a and B are output, which hinders the call of the user a who should originally make a call.

Therefore, in the adaptive array radio base station 1, the following processing is performed to remove the signal from the user B from the output signal. Fig. 5 is a schematic block diagram showing the configuration of the adaptive array radio base station 1.

First, when a signal from a user a is denoted by a (t) and a signal from a user B is denoted by B (t), a received signal x1(t) of the 1 st antenna #1 constituting the array antenna 2 of fig. 4 is expressed by the following equation.

Equation 1

x1(t)＝a1×A(t)+b1×B(t)

Here, a1 and b1 are coefficients that change in real time, as described later.

The reception signal x2(t) of the 2 nd antenna #2 is expressed by the following equation.

Equation 2

x2(t)＝a2×A(t)+b2×B(t)

Here, a2 and b2 are also coefficients that change in real time.

The reception signal x3(t) of the 3 rd antenna #3 is expressed by the following equation.

Equation 3

x3(t)＝a3×A(t)+b3×B(t)

Here, a3 and b3 are also coefficients that change in real time.

Similarly, the reception signal xn (t) of the nth antenna # n is expressed by the following equation.

Equation 4

xn(t)＝an×A(t)+bn×B(t)

Wherein an and bn are also coefficients that change in real time.

The coefficients a1, a2, a3,. and.. an indicate that the antennas #1, #2, #3,. and # n constituting the array antenna 2 are different in phase position with respect to the radio wave signal from the user a (for example, the antennas are arranged at intervals of about 1 meter, which is 5 times the wavelength of the radio wave signal), and that the reception intensity of each antenna is different.

The coefficients B1, B2, B3,. and bn indicate that the reception intensities of the antennas #1, #2, #3,. and # n are different from each other with respect to the radio signal from the user B, as well. As users move, their relationships change in real time.

Signals x1(t), x2(t), x3(t), xn (t) received by each antenna enter a receiving part 1R forming the adaptive array wireless base station 1 through corresponding switches 10-1, 10-2, 10-3, 10.

Weights w1, w2, w3,. and wn for the received signals of the respective antennas are applied from the weight vector control section 11 to the other input of the multipliers. These weights are calculated in real time by the weight vector control unit 11 as will be described later.

Therefore, the received signal x1(t) of antenna #1 becomes w1 × (a1A (t) + b1B (t)) by the multiplier 12-1, the received signal x2(t) of antenna #2 becomes w2 × (a2A (t) + b2B (t)) by the multiplier 12-2, the received signal x3(t) of antenna #3 becomes w3 × (a3A (t) + b3B (t)) by the multiplier 12-3, and the received signal xn (t) of antenna # n becomes wn × (an t) + bnb (t)) by the multiplier 12-n.

The adder 13 adds the outputs of the multipliers 12-1, 12-2, 12-3,. and 12-n, the outputs of which are described below.

Equation 5

w1(a1A(t)+b1B(t))

+w2(a2A(t)+b2B(t))

+w3(a3A(t)+b3B(t))

+…+wn(anA(t)+bnB(t))

If it is divided into a term for signal A (t) and a term for signal B (t), the output is

Equation 6

(w1a1+w2a2+w3a3+…，+wnan)A(t)

+(w1b1+w2b2+w3b3+…，+wnbn)B(t)

Here, the adaptive array wireless base station 1 identifies the user A, B, and calculates the above-described weights w1, w2, w3,. and wn so that only the signal from the desired user can be extracted. For example, in the example of fig. 5, in order to extract only the signal a (t) from the user a who should originally talk, the weight vector control unit 11 calculates the weights w1, w2, w3, a, b1, b2, b3, a, and bn such that the coefficients of the signal a (t) are all 1 and the coefficients of the signal b (t) are all 0, taking the coefficients a1, a2, a3, a.

That is, the weight vector control unit 11 calculates weights w1, w2, w3,. and wn in which the coefficient of the signal a (t) is 1 and the coefficient of the signal b (t) is 0 in real time by solving the following simultaneous linear equation.

Equation 7

w1a1+w2a2+w3a3+…，+wnan＝1

w1b1+w2b2+w3b3+…，+wnbn＝0

Although the solution of the simultaneous linear equation is not described, it is well known from the literature listed above and is currently used in an adaptive array radio base station.

Thus, by setting the weights w1, w2, w3,. and wn, the output signal of the adder 13 is

Calculator 8 (output signal)

(output signal) ═ 1 × a (t) +0 × b (t) ═ a (t)

[1, user recognition and training signals ]

The identification of the user A, B proceeds as follows. Fig. 6 is a schematic diagram showing a frame structure of a radio wave signal of a mobile phone. Radio signals of a mobile phone are mainly composed of preambles (preambles) composed of signal sequences known to the radio base station and data (voice and the like) composed of signal sequences unknown to the radio base station.

The preamble signal sequence includes information for distinguishing whether the user is a desired user to talk via the radio base station. The weight vector control unit 11 (fig. 5) of the adaptive array radio base station 1 compares the training signal corresponding to the user a retrieved from the memory 14 with the received signal series, and performs weight vector control (determination weighting) so as to extract a signal which is considered to include the signal series corresponding to the user a. The signal of the user A thus extracted is taken as the output signal S_RX(t) is outputted from the adaptive array radio base station 1 to the outside.

On the other hand, in fig. 5, an input signal S from the outside_TX(T) enters a transmission unit 1T constituting the adaptive array radio base station 1, and is supplied to one input of multipliers 15-1, 15-2, 15-3. Weights w1, w2, w3,. and wn calculated in advance from the received signals by the weight vector control unit 11 are copied and applied to the other inputs of the multipliers, respectively.

The input signals weighted by these multipliers are transmitted to the corresponding antennas #1, #2, #3, # n via the corresponding switches 10-1, 10-2, 10-3, #.

Here, since the signal transmitted by using the same array antenna 2 as in the reception is given a weight targeting the user a as in the case of the reception signal, the transmitted radio wave signal is received by the mobile phone 4 of the user a as if it has directivity to the user a. Fig. 7 is a diagram illustrating radio wave signal transmission between the user a and the adaptive array radio base station 1. As compared with the area 3 of fig. 4 showing the actual radio wave arrival range, the state of radio wave signals radiated from the adaptive array radio base station 1 with directivity is imaged with the mobile phone 4 of the user a as a target as shown by the assumed area 3a of fig. 7.

As described above, the PDMA scheme requires a technique for removing the same channel interference. In this regard, since the adaptive array adaptively subtends zero to the interference wave, the interference wave can be effectively suppressed even in the case where the level of the interference wave is higher than that of the desired wave, and therefore, is an effective technique.

However, when the base station uses an adaptive array, it is possible to reduce unnecessary radiation during transmission as well as to remove interference during reception. In this case, a method of regenerating the array pattern at the time of transmission using the array pattern at the time of reception or from the result of arrival direction estimation or the like is considered. The latter is applicable to both fdd (frequency Division duplex) and tdd (time Division duplex), but requires complicated processing. On the other hand, when the former is used for FDD, compensation such as array configuration and weight is necessary because the array pattern of the transmission and reception is different. Therefore, generally, good characteristics are obtained when the external time slots are continuous, on the premise of TDD application.

As described above, in the TDD/PDMA scheme in which the base station uses the adaptive array, when the downlink uses an array pattern (weight vector pattern) obtained in the uplink, if a dynamic rayleigh transmission rate with a wide angle is assumed, the error rate in the downlink may be deteriorated due to a time difference between the upper and lower lines. That is, there is a time interval from transmission of a radio wave from a user terminal to a base station via an uplink (uplink) line to radiation of a radio wave from a base station to a user terminal via a downlink (downlink) line, and therefore, when the moving speed of the user terminal cannot be ignored, the error rate deteriorates due to an error between the direction in which a radio wave from the base station is radiated and the direction in which the actual user terminal exists.

As a method for estimating the weight for the downlink in consideration of such a fluctuation of the transmission path, a method of performing first-order extrapolation using a weight vector value obtained in the uplink is proposed in non-patent document 1 or non-patent document 2.

However, if the temporal change of the weight is actually observed, the change is not a straight line, and therefore, the conventional method of first-order extrapolation of the weight vector has a problem of large error.

Therefore, in order to solve such a problem, attention has been paid to a technique of expressing weights of adaptive arrays uniquely by response vectors in respective antenna elements, and a technique of estimating temporal variations of response vectors to indirectly estimate weights, thereby suppressing deterioration of an error rate in a downlink due to a time difference between an upper line and a lower line even in the TDD/PDMA system, assuming a dynamic rayleigh transmission path such as a wide angle, has been proposed.

[2, Wireless device for suppressing deterioration of error rate in downlink due to time difference between upper and lower lines ]

Fig. 8 is a schematic block diagram showing a configuration of a radio apparatus (radio apparatus) 1000 of a PDMA base station according to conventional mode 1. In the configuration of fig. 8, 4 antennas #1 to #4 are provided in order to identify the users PS1 and PS 2. It is needless to say that the number of antennas is more generally N (N: natural number).

In the transmission/reception system 1000 of fig. 8, a reception unit SR1 for receiving signals from the antennas #1 to #4 and separating signals from corresponding users, for example, user PS1, and a transmission unit ST1 for transmitting signals to the user PS1 are provided. The connection of the antennas #1 to #4, the receiver SR1, and the transmitter ST1 is selectively switched by the antennas 10-1 to 10-4. I.e. the received signal RX received by each antenna₁(t)、RX₂(t)、RX₃(t)、RX₄(t) enters a receiving section SR1 via corresponding antennas 10-1, 10-2, 10-3, and 10-4, and is supplied to a reception weight vector computer 20 and a reception coefficient vector computer 22, and is supplied to corresponding multipliers 12-1, 12-2, 12-3, and 12-4, respectivelyOne input.

To the other inputs of these multipliers are applied weighting coefficients wrx11, wrx21, wrx31, wrx41 of the received signals for the respective antennas from the reception weight vector computer 20. These weighting coefficients are calculated in real time by the reception weight vector computer 20, as in the conventional example.

The transmission unit ST1 includes a transmission coefficient vector estimator 32 for receiving the reception coefficient vector calculated by the reception coefficient vector calculator 22, estimating a transmission path at the time of transmission, that is, a virtual reception coefficient vector at the time of transmission, as described later, and obtaining a transmission coefficient vector; a memory 34 for storing and holding data by transferring data to and from the transmission coefficient vector estimator 32; a transmission weight vector calculator 30 for calculating a transmission weight vector based on the estimation result of the transmission coefficient vector estimator 32; and multipliers 15-1, 15-2, 15-3, 15-4, respectively, receiving the transmission signals at one input and applying weighting coefficients wrx11, wrx21, wrx31, wrx41 from the transmission weight vector computer 30 to the other input. The outputs from multipliers 15-1, 15-2, 15-3, 15-4 are provided to antennas #1- #4 via antennas 10-1-10-4. Although not shown in fig. 8, each user may be provided with the same configuration as the receiver SR1 and the transmitter ST 1.

[3, operation principle of adaptive array ]

The operation of the receiver SR1 will be briefly described below.

Reception signal RX received by an antenna₁(t)、RX₂(t)、RX₃(t)、RX₄(t) is represented by the following formula.

Equation 9

RX₁(t)＝h₁₁Srx₁(t)+h₁₂Srx₂(t)+n₁(t)…(1)

RX₂(t)＝h₂₁Srx₁(t)+h₂₂Srx₂(t)+n₂(t)…(2)

RX₃(t)＝h₃₁Srx₁(t)+h₃₂Srx₂(t)+n₃(t)…(3)

RX₄(t)＝h₄₁Srx₁(t)+h₄₂Srx₂(t)+n₄(t)…(4)

Wherein the signal RX_j(t) represents a reception signal of the jth antenna (j is 1, 2, 3, 4), and the signal Srx_iAnd (t) represents a signal transmitted by the ith (i-1, 2) user. In addition, the coefficient h_jiRepresenting the complex coefficient, n, of the signal from the ith user received at the jth antenna_j(t) represents noise contained in the jth received signal.

The expressions (1) to (4) are expressed in vector form as follows.

Equation 10

X(t)＝H₁Srx₁(t)+H₂Srx₂(t)+N(t)…(5)

X(t)＝[RX₁(t)，RX₂(t)，...，RX₄(t)]^T …(6)

H_i＝[h_1i，h_2i，...，h_4i]^T，(i＝1，2)…(7)

N(t)＝[n₁(t)，n₂(t)，...，n₄(t)]^T …(8)

Where x (t) represents an input signal vector, Hi represents a reception coefficient vector of the i-th user, and n (t) represents a noise vector. In addition, [. ] T denotes a transposition of [. ] T.

As shown in fig. 8, the adaptive array antenna multiplies the input signals from the antennas by weighting coefficients wrx11 to wrx41 to synthesize signals, which are received signals SR_XAnd (t) outputting.

Further, under the above preparation, for example, the signal Srx transmitted by the 1 st user is extracted₁The operation of the adaptive array at (t) is as follows.

The output signal y1(t) of adaptive array 100 is expressed by the following expression by multiplying the vector of input signals x (t) by the vector of weight vector W1.

Equation 11

y1(t)＝X(t)W₁ ^T …(9)

W₁＝[wrx₁₁，wrx₂₁，wrx，₃₁，wrx₄₁]^T …(10)

That is, the weight vector W1 is multiplied by the jth input signal RX_jThe weighting coefficient wrxj1(j 1, 2, 3, 4) of (t) is a vector of elements.

Note that, when the input signal vector x (t) expressed by the equation (5) is substituted into y1(t) expressed by the equation (9), the following is expressed.

Equation 12

y1(t)＝H₁W₁ ^TSrx₁(t)+H₂W₁ ^TSrx₂(t)+N(t)W₁ ^T…(11)

However, when the adaptive array 100 operates ideally, the weight vector control unit 11 sequentially controls the weight vector W1 so as to satisfy the following simultaneous equations by a known method.

Equation 13

H₁W₁ ^T＝1 …(12)

H₂W₁ ^T＝0 …(13)

If the weight vector W1 is completely controlled to satisfy equations (12) and (13), the output signal y1(t) from the adaptive array 100 is finally as shown below.

Equation 14

y1(t)＝Srx₁(t)+N₁(t) …(14)

N₁(t)＝n₁(t)w₁₁+n₂(t)w₂₁+n₃(t)w₃₁+n₄(t)w₄₁

…(15)

That is, the output signal y1(t) is the signal Srx sent by the 1 st of the two users₁(t)。

[4, overview of operation of Wireless device 1000 ]

Fig. 9 is a flowchart illustrating an overview of the actions of a conventional wireless device 1000. The wireless device 1000 is focused on a weight vector (weighting coefficient vector) uniquely representing the adaptive array by the reception coefficient vector of each antenna element, and estimates the temporal variation of the reception coefficient vector to indirectly estimate the weight.

First, the reception unit SR1 estimates a propagation path of the reception signal from the reception signal (step S100). Estimation of the transmission path corresponds to obtaining the impulse response of the signal transmitted from the user in equations (1) to (4). In other words, in expressions (1) to (4), if the reception coefficient vector H1 can be estimated, for example, the transmission path when receiving a signal from the user PS1 is possible.

Then, the transmission coefficient vector estimator 32 predicts the transmission path at the time of transmission, that is, predicts the reception coefficient vector at the time of transmission from the reception coefficient vector at the time of reception (step S102). The predicted reception coefficient vector corresponds to a transmission coefficient vector at the time of transmission.

Then, the transmission weight vector calculator 30 calculates a transmission weight vector from the predicted transmission coefficient vector, and outputs the transmission weight vector to the multipliers 15-1 to 15-4 (step S104).

[5, operation of reception coefficient vector computer 22 ]

Next, the operation of the reception coefficient vector computer 22 of the conventional mode 1 shown in fig. 8 will be described. First, when the number of antenna elements is 4 and the number of users performing simultaneous communication is 2, signals output from the receiving circuit via each antenna are expressed by the above equations (1) to (4).

In this case, if expressions for expressing the received signals of the antennas represented by expressions (1) to (4) by vectors are described again, the expressions (5) to (8) will be described below.

Equation 15

X(t)＝H₁Srx₁(t)+H₂Srx₂(t)+N(t) …(5)

X(t)＝[RX₁(t)，RX₂(t)，...，RX_a(t)]^T …(6)

H_i＝[h_1i，h_2i，...，h_ni]^T，(i＝1，2) …(7)

N(t)＝[n₁(t)，n₂(t)，...，n_n(t)]^T …(8)

However, when the adaptive array operates well, since signals from the respective users are separated and extracted, all of the signals srxi (t) (i 1 and 2) are known values.

In this case, using srxi (t) as a known signal, the reception coefficient vector H can be derived as described below₁＝[h₁₁、h₂₁、h₃₁、h₄₁]And H₂＝[h₁₂、h₂₂、h₃₂、h₄₂]. That is, the received signal is compared with a known user signal, for example, a signal Srx from the 1 st user₁(t) the results are multiplied to calculate the overall average (time average) as follows.

Equation 16

E[X(t)·Srx₁ ^*(t)]＝H₁·E[Srx₁(t)·Srx₁ ^*(t)]

+H₂·E[Srx₂(t)·Srx₁ ^*(t)]+E[N(t)·Srx₁ ^*(t)]

…(16)

In formula (16), E [. ] represents a time average, and S [ ((t) ] represents a complex conjugate of S (t)).

When the time for obtaining the average is sufficiently long, the average value is as follows.

Equation 17

E[Srx₁(t)·Srx₁ ^*(t)(t)]＝1 …(17)

E[Srx₂(t)·Srx₁ ^*(t)]＝0 …(18)

E[N(t)·Srx₁ ^*(t)]＝0 …(19)

Wherein the value of the formula (18) is 0 because of the signal Srx₁(t) and Srx₂(t) are not related to each other. The value of expression (19) is 0 because of the signal Srx₃(t) and noise N (t) are uncorrelated with each other.

Therefore, the ensemble averaging result of equation (16) is equal to the reception coefficient vector H as shown below₁。

Equation 18

E[X(t)·Srx₁ ^*(t)]＝H₁…(20)

Through the above steps, the reception coefficient vector H of the signal transmitted from the 1 st user PS1 can be estimated₁。

Similarly, by inputting the vector X (t) of the signal and the signal Srx₂(t) ensemble averaging, the reception coefficient vector H of the signal transmitted from the 2 nd user PS2 is estimated₂。

The ensemble averaging is performed on, for example, the first predetermined number of data symbol sequences and the last predetermined number of data symbol sequences in 1 slot received.

[6 estimation of Transmission coefficient vector ]

Fig. 10 is a schematic diagram illustrating the operation of the transmission coefficient vector estimator 32. Consider an 8-slot structure in which 4 users are allocated to each of the upper and lower lines as PDMA radio bursts (bursts). For example, the slot structure sets the first 31 symbols as the 1 st training symbol sequence, the next 68 symbols as the data symbol sequence, and the last 31 symbols as the 2 nd training symbol sequence.

As described above, training symbol columns are set at the beginning and the end of the uplink slot, and the reception coefficient vectors of both are calculated using the algorithm of the reception coefficient vector computer 22.

Next, the reception coefficient vector for the downlink is estimated by straight line extrapolation. That is, when the value of 1 arbitrary time t of the element of the reception coefficient vector is f (t), the value f (t) of the downlink slot at time t can be predicted from the value f (t0) of the start training symbol sequence of the uplink slot at time t0 and the value f (t1) of the last training symbol sequence of the uplink slot at time t1 as follows.

Equation 19

f(t)＝[f(t1)-f(t0)]/(t1-t0)×(t-t0)

+f(t0)

In the above description, the training symbol sequence is set at the beginning and the end of the uplink slot, and the first-order extrapolation is performed, but the training symbol sequence may be set in the central part of the uplink slot, and the value f (t) at the time t may be estimated by 2-order extrapolation from the 3-point value in the uplink slot of the reception coefficient vector. Alternatively, the positions of the training symbol sequences set in the uplink timeslot may be increased to perform higher-order extrapolation.

[7, determination of Transmission weight vector ]

When the estimated value of the reception coefficient vector at the transmission time is obtained as described above, the transmission weight vector can be obtained by one of the following 3 methods.

i) The method based on orthogonalization comprises the following steps:

consider a weight vector W for a user PS1 at time T-iT (i: natural number, T: unit time interval)⁽¹⁾(i)＝[wtx₁₁、wtx₁₂、wtx₁₃、wtx₁₄]. In order to give invalidity to the user PS2, the following condition is preferably satisfied.

The transmission path (reception coefficient vector) predicted for the user PS2 is set to V⁽²⁾(i)＝[h1’⁽²⁾(i)、h2’⁽²⁾(i)、h3’⁽²⁾(i)、h4’⁽²⁾(i)]. Wherein, hp'^(q)(i) Is the predicted value of the reception coefficient vector of the qth user to the pth antenna with respect to time t. Similarly, the transmission path V is measured for the user PS1⁽¹⁾(i)。

At this time, W is determined⁽¹⁾(i) Let W be⁽¹⁾(i)TV⁽²⁾(i) 0. As the limiting conditions, the following conditions c1), c2) are set.

c1) W (1) (i) TV (1) (i) ═ g (constant value)

c2) Let | | | W (1) (i) | be the minimum.

Condition c2) corresponds to minimizing the transmit power.

ii) method of using analog correlation matrix

Here, as described above, the adaptive array is configured by several antenna elements and a portion for controlling the weight value of each element. In general, when an input vector of an antenna is represented by x (t) and a weight vector is represented by W, the weight vector is controlled so that an output y (t) becomes W^TWhen the mean square error between x (t) and the reference signal d (t) is minimum (MMSE criterion: least square error criterion), the optimum weight Wopt is given by the following equation (Wiener). That is to say that the first and second electrodes,

equation 20

Wherein it is necessary to satisfy

Equation (21).

R_xx＝E[x^*(t)x^T(t)]…(22)

r_xd＝E[x^*(t)d(t)]…(23)

Wherein, Y^TDenotes the transpose of Y, Y denotes the complex area of Y, E [ Y]Indicating the ensemble average. The array pattern is generated by the weight value, and unnecessary interference waves are suppressed by the adaptive array.

However, in the method using the analog correlation matrix, equation (21) is calculated from the analog correlation matrix described below. That is, the estimated complex received signal coefficient h 'is used'^(k)n (i) to calculate a weight vector W for user k^(k)(i) In that respect When the array call vector of the kth user is set as V^(k)(i) The time is determined as follows.

Calculator (22)

In this case, the autocorrelation array rxx (i) of the dummy received signal at t ═ iT uses V^(k)(i) And is represented by the following formula.

Calculator (23)

Where N is a virtual noise term added for the purpose of taking an integer rxx (i), and in this calculation, for example, N is 1.0 × 10^-3。

The correlation vector τ xd (i) of the received signal and the reference signal is expressed by the following equation.

Calculator (24)

Therefore, the weight for the downlink at time t ═ iT can be obtained by equations (21), (25), and (26). The inverse matrix operation of equation (25) can perform optimum calculation for user k by the aid of the inverse matrix theorem. In particular, in the case of 2 users, the weight can be calculated by the following simple expression.

Calculator (25)

When the autocorrelation array is provided in this manner, a method of calculating the weight vector is described in non-patent document 3, for example.

iii) method of directing an electron beam (beam) towards a user PS 1:

when focusing on so-called directing of the electron beam to the user PS1, it is preferable that the following equation is satisfied

Equation (26).

W(1)⁽ⁱ⁾＝V⁽¹⁾(i)^*

In any of the methods described above, when a dynamic rayleigh transmission path such as a wide angle is assumed after a weight vector is determined and transmitted at the time of transmission, it is possible to suppress deterioration of an error rate in a downlink due to a time difference between an upper line and a lower line even in the TDD/PDMA system.

[8, modification of conventional mode 1 ]

In conventional mode 1, the propagation path is estimated by using the ensemble average of equation (20). Fig. 11 is another configuration diagram of the reception coefficient vector computer 22 according to the modification of the conventional mode 1. As shown in fig. 11, when the signal from the i-th antenna is multiplied by the complex conjugate signal Srx1(t) of the signal Srx1(t) from the 1 st user PS1 output from the adaptive array antenna by the multiplier 40 and then passed through the narrow band filter 42, the output from the narrow band filter 42 is changed to h_i1(t) of (d). By performing the above operation for all the antennas, the reception coefficient vector with respect to the user PS1 can be obtained.

Similarly, after multiplying the signal from the i-th antenna by the complex conjugate signal Srx2(t) of the signal Srx2(t) from the 2 nd user PS2 output from the adaptive array antenna, the signal is multiplied by the complex conjugate signal Srx2(t)When the output of the narrow-band filter is changed to h by a narrow-band filter (not shown)_i2(t) of (d). By performing the above operation for all the antennas, the reception coefficient vector with respect to the user PS2 can be obtained.

The subsequent steps of prediction of the transmission path and determination of the transmission weight vector can be performed as in conventional mode 1. Therefore, this structure can achieve the same effect as in the conventional mode 1.

[9, conventional mode 2]

In conventional mode 1, the propagation path is estimated by using the ensemble average of equation (20). In contrast, in conventional mode 2, the propagation path is estimated using the correlation vector in the adaptive array.

That is, as shown in the above equations (21) to (23), when the adaptive array operates according to the MMSE criterion, the optimal weight vector Wopt is expressed as described above using the reference signal d (t), the autocorrelation matrix Rxx, and the correlation vector rxd.

Equation 27

R_xx＝E[x^*(t)x^T(t)] …(22)

r_xd＝E[x^*(t)d(t)] …(23)

When the weight vector for the 1 st user PS1 is obtained, each component of the correlation vector rxd is as follows.

Equation 28

r_xd＝[E[R_x1(t)d(t)^*]，…，E[R_X4(t)d(t)^*

]]^T

～[h₁₁、h₂₁、h₃₁、h₄₁]

That is, in the process of obtaining the weight vector for the 1 st user PS1 by the reception weight vector calculator 22, the reception coefficient vector of the user PS1 can be obtained by using the value of the derived correlation vector rxd.

Therefore, for example, if the start and the end of the uplink slot include training symbol sequences, the propagation path of the user PS1 can be estimated at times t0 and t1, and the propagation path at time t at the time of transmission can be predicted, as in fig. 10. The same is true for other users. The following steps of prediction of transmission paths and determination of transmission weights may be performed as in conventional mode 1. Therefore, even such a step can achieve the same effect as in the conventional form 1.

[10, conventional mode 3]

In conventional mode 2, the propagation path is estimated by using the correlation vector. Here, as conventional mode 3, another calculation method of the reception coefficient vector computer 22 will be described.

The signal Srx1(t) from the 1 st user PS1 output from the adaptive array antenna and the virtual reception coefficient vector h 'are subtracted from the value of the signal Rxi (t) of the 1 st antenna'_i1The value obtained as a result of the multiplication (t) is referred to as RXi' (t). That is to say that the first and second electrodes,

equation 29

RXi′(t)＝RXi(t)-h′_i1(t)·Srx1(t)

In the reception coefficient vector computer 22 of the conventional form 3, calculation of the number of cells to be realized by E [ | RXi' (t) & gt) is performed by the following sequential method²]Minimum h'_i1(t) of (d). Here, data from k-0 to k-M (e.g., 119) is included in 1 uplink timeslot.

When the receiving coefficient vector at true is hⁱ¹(t), E [ | RXI' (t) & gt²]The minimum value is when the following conditions are satisfied.

Equation 30

h′_i1(t)＝h_i1(t)

If the fastest descent method is used, then relative h 'is obtained'_i1(k) (value at time t: kT, k: natural number).

Equation 31

h′_i1(k+1)＝h′_i1(k)

+μ{RXi(k)-h′_i1(k)·Srx1(k)}·Srx1^*(k

)

Where the constant μ is the step size. Further, although not particularly limited, h 'is preferably used'_i1(k) Is set to h'_i1(0)＝0。

Fig. 12 is a schematic diagram illustrating a principle of estimating a propagation path when sequentially estimating. Fig. 12 is a graph in comparison with fig. 10. Corresponding to the determination of h 'by asymptotic equation'_i1(k) In the uplink slot, time t0 is the end time of the preamble, and time t1 is the end time of the uplink slot. Therefore, it is preferable that the training symbol sequence exists only in the beginning of the uplink slot.

When this operation is performed for all the antennas, a reception coefficient vector for the user PS1 is obtained, and a transmission path is measured. When the same processing is performed for the user PS2, a reception coefficient vector for the user PS2 is obtained, and a transmission path is predicted. The subsequent determination step of the transmission weight vector can be performed as in conventional mode 1. Therefore, this structure can achieve the same effect as in the conventional mode 1.

In addition, even in the method based on another asymptotic expression described below, the estimation of the propagation path can be performed in the same manner. In fig. 12, the time t0 is the end time of the preamble, but the time t0 is not necessarily limited to this position. The time t0 may be present in the training symbol sequence, or may be present in the data symbol sequence. Note that the time t1 is the end time of the uplink timeslot, but the time t1 is not necessarily limited to this position.

[11, conventional mode 4]

In conventional mode 3, a reception coefficient vector is sequentially obtained for each user. As conventional mode 4, a further calculation method of the reception coefficient vector computer 22 will be described below.

The signal Srx1(t) from the 1 st user PS1 output by the adaptive array antenna and the virtual reception coefficient vector h 'are subtracted from the value of the signal Rxi (t) of the ith antenna'_i1(t) the result of the multiplication and the vector h 'of virtual reception coefficients from the signal Srx2(t) from the 2 nd user PS 2'_i2The value obtained as a result of the multiplication (t) is referred to as RXi' (t). That is to say that the first and second electrodes,

equation 32

RXi′(t)＝Rxi(t)-h′_i1(t)·Srx1(t)

-h′_i2(t)·Srx2(t)

In the reception coefficient vector computer 22 of the conventional form 4, calculation of E [ | Rxi' (t) & gt survival rate is carried out in the following manner²]Minimum h'_i1(t) and h'_i2(t) of (d). That is to say that the first and second electrodes,

equation 33

H′_i(t)＝[h′_i1(t)，h′_i2(t)]^TSRX(t)

＝[Srx1(t)，Srx2(t)]^T

At this time, since the signal is based on the so-called E [ | RXI' (t) |2]About vector H'_i(t) gradient of 0, when the true reception coefficient vector is set to H_iOPT(t) the following equation is derived.

Equation 34

H_iOPT(t)＝R_ss ^-ir_sxR_ss ^-1

＝E[SRX^*(t)SRX^T(t)]r_sx

＝E[SRX^*(t)RX_i(t)]

The principle in the case of estimating the transmission path is performed in the same manner as the principle diagram shown in fig. 10, for example. By performing the above operation for all antennas, reception coefficient vectors for the user PS1 and the user PS2 can be obtained, and the transmission path can be measured. The subsequent determination of the transmission weight vector is performed in the same manner as in conventional mode 1. Therefore, the same effect as that of the conventional mode 1 can be achieved by this configuration.

[12, conventional mode 5]

As conventional mode 5, another calculation method of the reception coefficient vector computer 22 will be described. The following description is equivalent to the so-called recursive minimum 2 multiplication (RLS algorithm: recursiveLeast-Squares algorithm).

Subtracting an output signal vector SRx (t) output from the adaptive array antenna and a virtual reception coefficient vector H 'from a signal Rxi (t) of the ith antenna'_i ^TThe value obtained as a result of the multiplication (t) is referred to as RXi' (t). That is to say that the first and second electrodes,

equation 35

Rxi′(t)＝RXi(t)-H′_i ^T(t)SRX(t)

According to the RLS algorithm, the following holds.

Equation 36

H′_i(k+1)＝H′_i(k)+R_ss ^-1(k)SRX^*(k)RXi′(k) …(29)

RXi′(k)＝RXi(k)-H′₁ ^T(k)SRX()k …(30)

R_ss ^-1(k)＝1/λ·R_ss ^-1(k-1)

-1/λ·[R_ss ^-1(k-1)SRX^*(k)SRX(k)^T R_ss ^-1(k-1)]/[λ+

SRX(k)^TR_ss ^-1(k-1)SRX^*(k)] …(31)

Here, data from k-0 to k-M (e.g., 119) is included in one uplink slot (slot). Wherein the constant lambda (lambda is more than 0 and less than or equal to 1) is the forgetting coefficient. H'_iThe initial value of each element of (t) is also not particularly limited, but is preferably 0.

In this way, the estimation of the transmission path is performed in the same manner as the schematic diagram shown in fig. 12. When this operation is performed for all the antennas, a reception coefficient vector for the user PS1 is obtained, and a transmission path is predicted. When the same processing is performed for the user PS2, a reception coefficient vector for the user PS2 is obtained, and a transmission path is predicted. The subsequent determination step of the transmission weight vector can be performed as in conventional mode 1. Therefore, this structure can achieve the same effect as in the conventional mode 1.

[13, modification of conventional mode 5]

In conventional mode 5, the transmission path is predicted from two points of data at time t0 and time t1 according to the principle shown in fig. 12. In the modification of conventional mode 5, a regression curve is calculated from the number of data symbols +1 impulse responses sequentially obtained in the uplink slot interval, and first-order extrapolation is performed.

Fig. 13 is a schematic diagram illustrating a principle of calculating a regression curve from impulse responses sequentially obtained in an uplink slot section and estimating a transmission path (impulse response). The estimation error can be suppressed to be small by a large increase in the number of data as compared with extrapolation of only two points.

As the extrapolation method based on the regression curve, not only the above-described first-order extrapolation, but also an extrapolation method using a higher-order extrapolation curve or an extrapolation method using a regression based on a periodic function such as a sine-cosine function may be used.

[14, conventional mode 6]

As conventional mode 6, another calculation method of the reception coefficient vector computer 22 will be described. The following description is equivalent to the so-called fastest descent method (LMS algorithm).

In the same manner as in conventional mode 5, the output signal vector srx (t) output from the adaptive array antenna and the virtual reception coefficient vector H 'are subtracted from the signal rxi (t) of the ith antenna'_i ^TThe value obtained as a result of the multiplication (t) is referred to as RXi' (t). That is to say that the first and second electrodes,

equation 37

RXi′(t)＝RXi(t)-H′_iT(t)SRX(t)

According to the LMS algorithm, the following holds.

Equation 38

H′_i(k+1)＝H′_i(k)+μSRX^*(k)RXi′(k)

Here, data from k-0 to k-M (e.g., 119) is included in one uplink timeslot.

Where the constant μ is a step length, the following relationship must be satisfied depending on the convergence condition.

Equation 39

0＜μ＜1/λmax

Where λ max is the maximum eigenvalue of the correlation matrix Rxx. And, H'_iThe initial value of each element of (t) is also not particularly limited, but is preferably 0.

In this way, the estimation of the transmission path is performed in the same manner as the schematic diagram shown in fig. 12. When this operation is performed for all the antennas, a reception coefficient vector for the user PS1 is obtained, and a transmission path is predicted. When the same processing is performed for the user PS2, a reception coefficient vector for the user PS2 is obtained, and a transmission path is predicted. The subsequent determination step of the transmission weight vector can be performed as in conventional mode 1. Therefore, this structure can achieve the same effect as in the conventional mode 1.

In addition, in conventional form 6, as in the modification of conventional form 5, a regression curve may be calculated from the number of data symbols sequentially obtained in the uplink slot interval plus 1 impulse response, and first-order extrapolation may be performed. The method of estimating the transmission path is not limited to the methods of conventional form 1 to conventional form 6 described above, and may be, for example, a direct Solver (SMI) or the like. The case of the SMI mode can predict a transmission path according to the principle shown in fig. 10.

[15, conventional mode 7]

As conventional mode 7, another calculation method of the reception coefficient vector computer 22 will be described. The following description is equivalent to a so-called AR model (autoregersive model).

Hereinafter, one of the elements of the reception coefficient vector is representatively denoted by f (t). That is, fig. 14 is a schematic diagram of the AR model of conventional form 7 shown in fig. 1. As shown in fig. 14, the time variation of element f (t) is regarded as an AR model. Where v (t) is the prediction error (white gaussian noise).

Fig. 15 is a schematic diagram of the 2 nd principle of the AR model of the conventional mode 7. As shown in fig. 15, the AR model may be formed by a filter having the inverse characteristic of the filter a (z). When the above-mentioned v (t) is input to the AR model, the element f (t) can be reproduced, and when unknown white noise is input, the future of the element f (t) can be predicted.

Fig. 16 is a schematic block diagram showing the structure of the filter a (z) shown in fig. 14. In FIG. 16, the multiplication coefficient a0-aM is determined so that E [ | v (k) & gt²]And minimum. If { f (k) } is an AR model of order M, then { v (k) } is a white Gaussian process. Fig. 17 is a schematic block diagram showing the structure of the inverse filter w (z) of the filter a (z) in the AR model. When k is within the observation interval, the error filter output v (k) in fig. 16 is set as the input in fig. 17. When the observation interval is exceeded, white Gaussian noise is provided as input. This calculation method can achieve the same effect as in the conventional mode 1, as in the other methods.

[16, conventional mode 8]

Fig. 18 is a schematic block diagram showing a configuration of a radio apparatus (radio base station) 2000 of a PDMA base station according to conventional embodiment 8. The difference from the configuration of the radio apparatus (radio base station) 1000 of conventional mode 1 shown in fig. 8 is that it further includes a moving speed determiner 52 for receiving an output from the reception coefficient vector computer 22 and determining a moving speed of the user terminal; and a changeover switch 54 for receiving the output of the reception weight vector calculator 20 and the output of the transmission weight vector calculator 30, and selectively supplying the outputs to the multipliers 15-1 to 15-4 in accordance with the determination result of the moving speed determiner 52. The configuration is the same as that of any of the conventional configurations 1 to 7 (radio base station).

That is, as described above, in the area where the moving speed of the user terminal is small, rather than performing such prediction in order to predict an error in the process of estimating a transmission path, predicting a transmission path, or the like, the reception weight vector is used as it is as the transmission weight vector as in the conventional configuration of fig. 5.

Therefore, in the wireless device 2000 of the conventional mode 8, when the moving speed determiner 52 determines that the terminal is moving at a speed lower than the predetermined moving speed, the reception weight vector is supplied as it is to the multipliers 15-1 to 15-4 by the changeover switch 54. When the moving speed determiner 52 determines that the terminal is moving at a speed higher than the predetermined moving speed, the output of the transmission weight vector computer 30 is supplied to the multipliers 15-1 to 15-4 by the changeover switch 54. With the above configuration, data transmission with a low error rate can be performed in a wide moving speed range of the terminal.

[17, conventional mode 9]

Fig. 19 is a schematic block diagram showing a configuration of a radio apparatus (radio base station) 3000 of a PDMA base station according to conventional embodiment 9. The configuration of the radio apparatus (radio base station) 1000 of the conventional mode 1 shown in fig. 8 is different in that it further includes a reception level calculator 56 for receiving signals from the array antennas #1 to #4 and calculating the levels of the received signals; a reception level determiner 58 for receiving the output from the reception level computer 56 and determining the reception level of the user terminal; and a changeover switch 54 for receiving the output of the reception weight vector calculator 20 and the output of the transmission weight vector calculator 30 and selectively supplying the outputs to the multipliers 15-1 to 15-4 in accordance with the determination result of the reception level determiner 56. The other structure is the same as that of the wireless device of any of the existing forms 1 to 7.

That is, in the region of the level of the received signal from the user terminal, because of the prediction error in the process of estimating the transmission path, predicting the transmission path, or the like, the reception weight vector is used as it is as the transmission weight vector as in the conventional configuration of fig. 5 without performing such prediction.

Therefore, in the wireless device 3000 of the conventional mode 9, when the reception level determiner 58 determines that the level of the reception signal from the terminal is lower than the predetermined reception level, the reception weight vector is supplied as it is to the multipliers 15-1 to 15-4 by the changeover switch 54. When the reception level determiner 58 determines that the level of the reception signal from the terminal is higher than a predetermined reception level, the output of the weight vector computer 30 is transmitted to the multipliers 15-1 to 15-4 as it is by the changeover switch 54.

With the above configuration, data transmission with a low error rate can be performed over a wide range of received signal levels. The received signal level of the signal from the user PS1, for example, is obtained from the reception coefficient vector by the following equation.

Equation 40

P₁＝H₁ ²/N＝(h₁₁ ²+h₂₁ ²+h₃₁ ²+h₄₁ ²)/N …(32)

The same is true for received signal levels from other users.

[18, conventional mode 10]

Fig. 20 is a schematic block diagram showing a configuration of a radio apparatus (radio base station) 4000 of a PDMA base station according to conventional embodiment 10. The difference from the configuration of the radio apparatus (radio base station) 3000 of the conventional mode 9 shown in fig. 19 is that the reception level determiner 58 is a terminal moving speed determiner/reception level determiner 60 and has a moving speed determining function similar to the moving speed determiner 52 of the conventional mode 8 in addition to the function of determining the reception level. The configuration is the same as that of the conventional configuration 9, i.e., the configuration of the radio apparatus (radio base station) 3000.

With the above configuration, data transmission with a low error rate can be performed in a wide moving speed range and a wide received signal level range of the mobile terminal.

According to the above-described conventional technique, by estimating the temporal variation of the reception coefficient vector of the adaptive array and indirectly estimating the variation of the weight, it is possible to suppress the error rate deterioration in the downlink due to the time difference between the upper and lower lines even in a dynamic rayleigh transmission path having a wide angle.

Further, according to the above conventional example, data with a low error rate can be transmitted in a wide moving speed range or/and a wide received signal level range of the mobile terminal.

Non-patent document 1: jia Teng, Dajia, Xiaochuan, Yiteng, Xin theory (B-II), 1998 1 month, vol.J81-B-II, No.1, p.1-9

Non-patent document 2: soil dwelling, big Mega, Tang Ge, letter technical newspaper, 1 month 1997, vol. RCS97-68, p.27-32

Non-patent document 3: tianzhong, Dajia, Xiaochuan, Yiteng, 10 months of 1998, technical news, vol. RCS98-117, p.103-108

In the case of the above-described conventional example, the estimation of the transmission path is a factor for determining the performance of the transmit adaptive array antenna. Here, extrapolation from the first training symbol and the last training symbol is used in the estimation of the transmission path. Here, noise can be averaged if there are samples having a sufficiently long training symbol length, and the noise can be estimated from a high-order regression curve having a first-order function or more. However, this training symbol length is not generally considered to be sufficiently long, and particularly, if a transmission path is estimated from the first training symbol and the last training symbol, the transmission path is generally estimated based on a regression curve, and there is a problem that an operation of obtaining the regression curve from a signal including noise is complicated.

Disclosure of Invention

The present invention has been made to solve the above-mentioned problems, and an object of the present invention is to provide a radio apparatus which can suppress the error rate deterioration in a downlink due to the time difference between an upper line and a lower line even in the TDD/PDMA system when assuming a dynamic rayleigh transmission path having a wide angle by focusing on the weight which uniquely indicates an adaptive array by a response vector of each antenna element, estimating the time variation of the response vector by averaging the estimated time difference signal, and indirectly estimating the weight (weight).

According to the array antenna communication device of the present invention, an antenna directivity is changed in real time, and signals are transmitted and received to and from a plurality of terminals in a time division manner, the array antenna communication device including:

a plurality of antennas arranged in a dispersed manner; and

a transmitting circuit and a receiving circuit sharing the plurality of antennas in performing transmission and reception of signals,

the receiving circuit includes:

a reception signal separation circuit configured to separate a signal from a specific terminal among the plurality of terminals based on signals from the plurality of antennas when receiving a reception signal; and

a reception transmission path averaging circuit that averages a transmission path from the specific terminal based on a time difference signal, i.e., a time difference of signals obtained by sampling signals from the plurality of antennas, when receiving the reception signal,

the transmission circuit includes:

a transmission path setting circuit that sets a transmission path for transmitting a transmission signal according to an average result of the reception transmission path averaging circuit; and

and a transmission directivity control circuit that updates the antenna directivity when the transmission signal is transmitted, based on a setting result of the transmission channel setting circuit.

In the array antenna communication device according to the present invention, it is preferable that an uplink slot of the signal transmitted and received from the specific terminal includes a training data area of a predetermined size of the uplink slot, the reception channel averaging circuit derives an average value of the transmission channel from the specific terminal based on a time difference signal in the training data area, and the transmission channel setting circuit predicts the transmission channel when the transmission signal is transmitted based on the average value.

In the array antenna communication device according to the present invention, it is preferable that the reception transmission path averaging circuit derives a reception coefficient vector corresponding to an impulse response from the specific terminal in the transmission path from the specific terminal based on the time difference signal in the training data region.

In the array antenna communication device according to the present invention, it is preferable that the reception channel averaging circuit derives the reception coefficient vector by a collective (ensemble) average of the time-difference signals of the reception signals from the plurality of antennas and the time-difference signal from the specific terminal separated by the reception signal separating circuit.

In the array antenna communication device according to the present invention, it is preferable that the uplink slot of the transmission/reception signal from the specific terminal includes a training data region having a predetermined number of training data provided in the uplink slot and a data region having a plurality of data representing information from the specific terminal, the reception transmission path averaging circuit derives a plurality of averages arranged in time series of the transmission path from the specific terminal based on a time difference signal between the training data region and the data region, and the transmission path setting circuit sets the transmission path for transmitting the transmission signal based on the plurality of averages.

In the array antenna communication device according to the present invention, it is preferable that the reception transmission path averaging circuit sequentially derives a plurality of reception coefficient vectors corresponding to an impulse response from the specific terminal on the transmission path from the specific terminal, based on the plurality of time-difference signals in the training data region and the data region.

In the array antenna communication device according to the present invention, it is preferable that the sequential derivation of the plurality of reception coefficient vectors is based on a fastest descent method.

In the array antenna communication device according to the present invention, it is preferable that the sequential derivation of the plurality of reception coefficient vectors is based on recursive minimum 2 multiplication.

In the array antenna communication device according to the present invention, it is preferable that the uplink slot of the transmission/reception signal from the specific terminal includes a training data region having a predetermined number of training data provided in the uplink slot and a data region having a plurality of data representing information from the specific terminal, the reception transmission path averaging circuit derives a plurality of estimated values of the transmission path from the specific terminal based on a time difference signal between the training data region and the data region, and the transmission path setting circuit predicts the transmission path when the transmission signal is transmitted by regression of the plurality of average values and extrapolation based on a regression result.

In the array antenna communication device according to the present invention, it is preferable that the reception transmission path averaging circuit sequentially derives a plurality of reception coefficient vectors corresponding to an impulse response from the specific terminal from the transmission path from the specific terminal based on a plurality of time difference signals in the training data region and the data region.

In the array antenna communication device according to the present invention, it is preferable that the sequential derivation of the plurality of reception coefficient vectors is based on a fastest descent method.

In the array antenna communication device according to the present invention, it is preferable that the sequential derivation of the plurality of reception coefficient vectors is based on recursive minimum 2 multiplication.

In the array antenna communication device according to the present invention, it is preferable that the received signal separating circuit includes: a reception weight vector calculation unit configured to receive reception signals from the plurality of antennas and derive a reception weight vector for separating the time-difference signal from the specific terminal in real time; a plurality of 1 st multipliers, each of which receives the reception time difference signals from the plurality of antennas at one input and receives an element corresponding to the reception weight vector at the other input; and an adder that adds signals from the plurality of multipliers, the transmission directivity control circuit including: a transmission weight vector calculation unit that derives a transmission weight vector from the estimation result from the transmission channel averaging circuit; and a plurality of 2 nd multiplication units which receive a transmission signal at one input, receive the transmission weight vectors at the other input, and supply the transmission weight vectors to the plurality of antennas.

Effects of the invention

As described above, according to the present invention, it is possible to set a transmission path in a setting circuit by an extremely simple method such as averaging, and a significant effect is obtained in that it is not a method of extrapolating an operation of an adaptive array antenna accompanied by a doppler frequency shift or a frequency difference of a reference clock between a base station and a mobile station in mobile communication based on a regression curve. Further, there are advantages that an error is small and an adaptive array operation can be performed before the frequency difference is compensated by the AFC.

Drawings

Fig. 1 is a schematic block diagram of a main part of an array antenna communication apparatus according to an embodiment of the present invention.

Fig. 2 is a diagram showing an example of signal point transition of the array antenna communication device according to the embodiment of the present invention.

Fig. 3 is a channel configuration diagram of various communication systems of frequency division multiple Access, time division multiple Access, and Path Division Multiple Access (PDMA).

Fig. 4 is a schematic diagram schematically showing a basic operation of the adaptive array radio base station.

Fig. 5 is a schematic block diagram showing the configuration of an adaptive array radio base station.

Fig. 6 is a schematic diagram showing a frame structure of a radio wave signal of a mobile phone.

Fig. 7 is a schematic diagram illustrating radio wave signal transmission between the adaptive array radio base station and the user.

Fig. 8 is a schematic block diagram showing a configuration of a radio apparatus (radio base station) 1000 of a PDMA base station according to conventional mode 1.

Fig. 9 is a flowchart illustrating an overview of the operation of the wireless device (wireless base station) 1000.

Fig. 10 is a schematic diagram illustrating the operation of the transmission coefficient vector estimator 32.

Fig. 11 is another configuration diagram of the reception coefficient vector computer 22 according to the modification of the conventional mode 1.

Fig. 12 is a schematic diagram showing a principle of estimating a propagation path when estimation is performed sequentially.

Fig. 13 is a schematic diagram showing the principle of estimating a transmission path by calculating a regression curve from impulse responses sequentially obtained in an uplink slot section.

Fig. 14 is a schematic diagram of the AR model 1 according to the conventional embodiment 7.

Fig. 15 is a schematic diagram of the 2 nd principle of the AR model of the conventional mode 7.

Fig. 16 is a schematic block diagram showing the structure of the filter a (z) shown in fig. 14.

Fig. 17 is a schematic block diagram showing the structure of the inverse filter w (z) of the filter a (z) in the AR model.

Fig. 18 is a schematic block diagram showing a configuration of a radio apparatus (radio base station) 2000 of a PDMA base station according to conventional embodiment 8.

Fig. 19 is a schematic block diagram showing a configuration of a radio apparatus (radio base station) 3000 of a PDMA base station according to conventional embodiment 9.

Fig. 20 is a schematic block diagram showing a configuration of a radio apparatus (radio base station) 4000 of a PDMA base station according to conventional embodiment 10.

Fig. 21 is a schematic block diagram showing a main part of an array antenna communication device which is not differentiated.

Detailed Description

In the description of the prior art, the actions of the receive coefficient vector computer 22 are as follows [ 5]]Shown is a signal Srx in the formula (5) to the formula (8)_i(t) (i is 1 or 2) as a known value, Srx_i(t) multiplying the reception signal X (t) by the reception coefficient vector H obtained by calculating the ensemble average formula (16)₁. The received signal x (t) is represented by formula (5). In investigating the received coefficient vector H to User1₁In the case of the details of (1), when focusing on fundamental waves constituting a multichannel in addition to vectors due to transmission paths, each fundamental wave is caused by the movement of User1A phase change due to a doppler frequency shift, and a phase change due to a difference in transmission/reception frequency between the User1 and a reference clock of the base station.

For quantitative survey, as an example, let us say that the transmission/reception time interval is 2.5msec, the transmission/reception frequency is 2GHz, the maximum interval between the 1 st training symbol and the 2 nd training symbol is 0.625msec, and User1 moves at 100km per hour. At this point, User1 is at a transmit-receive interval: the distance moved in 2.5msec does not exceed 6.9 cm. Even if the User1 moves to a location 10m from the base station, for example, the angle change thereof does not exceed 0.4 ° at the maximum, and therefore the pattern of the adaptive array antenna changes to a value substantially close to zero at the transceiving time interval.

On the other hand, the doppler frequency shift amount of the fundamental wave of the multichannel is maximum when moving toward the base station, and the doppler frequency shift amount at this time is 185Hz, and the phase changes by 41.6 degrees between the maximum interval of 0.625msec between the 1 st training symbol and the 2 nd training symbol. In the case of PHS, the frequency difference between the transmission and reception based on the reference clock difference between User1 and the base station is assumed to be ± 3ppm at the base station and the mobile station, respectively, and therefore, when the base station demodulates the signal from the mobile station, the maximum frequency difference is ± 12kHz, and the phase change amount between the maximum time difference of 0.625msec between the 1 st training symbol and the 2 nd training symbol is ± 2700 degrees. In consideration of the above, it is considered that the reception coefficient vector H is obtained₁The difficulty in this case is that the directivity changes due to the change in the transmission path, and there is a problem in the received data for estimating the transmission path.

Generally, AFC (automatic frequency control) is known as a method for canceling a frequency offset observed by these base stations, and a method for estimating a carrier frequency from a received signal and adjusting the frequency of a local frequency transmitter is generally used. However, since the AFC estimates the carrier frequency from the received signal, time is required for the convergence, and the correct reception coefficient vector H1 cannot be estimated within the convergence time.

Here, in order to further clarify the present invention, synchronous detection and delay detection of known technologies related to the present invention will be described.

As detection methods for demodulating a signal modulated with data, there are synchronous detection and delay detection. The synchronous detection is a detection method in which a modulated signal is multiplied by a carrier wave synchronized with the center frequency of the modulated wave to obtain a demodulated signal. Since a carrier wave of a center frequency of a modulation wave originally used in modulation is used for demodulation, the data error rate characteristic after detection has a good value. However, in order to multiply the carrier wave synchronized with the center frequency of the modulated wave, it is necessary to estimate the center frequency of the modulated wave on the receiving side, and the estimation accuracy determines the demodulation characteristics of the synchronous detection.

On the other hand, delay detection is a method of performing detection by performing complex multiplication of a received signal and a received signal delayed by 1 symbol, and measuring a difference (including a phase and, in some cases, an amplitude change amount) between 1 symbol of a modulated wave. In the case of delay detection, since carrier reproduction is not required, a simple structure can be realized. On the other hand, since the reference signal for demodulation is the signal itself 1 symbol ago, there is a problem that the reception error rate characteristic is deteriorated as compared with the synchronous detection of carrier reproduction.

In mobile communication, the variation of the transmission path is large, and it is sometimes impossible to secure a time enough to reproduce only a carrier wave in synchronous detection. On the other hand, in the case of delay detection, since the signal itself before 1 symbol is used as a reference signal, only the change of the transmission path between the symbols becomes a problem, and therefore, in many cases, a good characteristic is obtained in the delay detection method. That is, the nature of the carrier frequency used for detection is reflected more in the detection characteristic than the characteristic of the detection method itself.

Here, the estimation of the transmission path of the array antenna communication device according to the present invention will be described while considering the relationship between synchronous detection and delay detection.

Fig. 1 is a block diagram of an array antenna communication device 10 according to an embodiment of the present invention. B-1 is an antenna, B-2 is a transmit/receive switch, B-3 is a receive signal differential, and B-3-1 in the receive signal differential B-3 is a delay element. B-4 is a reception weighting unit, B-5 is an adder, B-6 is a reference signal differentiator, B-7 is a reception adaptive processing unit, B-8 is a transmission adaptive processing unit, and B-9 is a transmission weighting unit.

The reception signal inputted from the antenna B-1 is inputted to the reception signal differential B-3 through the transceiving switch B-2. Here, the difference from the signal providing the unit time delay is obtained by the delay element B-3-1 and output. The output signals are weighted by a reception weighting unit B-4 and added by an adder B-5 to be output as reception signals. Here, the output of the adder B-5 is compared with the signal obtained by taking the signal of the previous stage of each antenna reception weighting device B-4 into the reception adaptive processing section and the reception output signal is differentiated from the reference signal by the reference signal differentiator B-6, and the differential signal of the desired signal included in the reception signal is compared with the differential signal of the reference signal, so that the signals from the plurality of antennas are appropriately weighted by B-4, whereby the differential signals of the desired signal are additively combined, and the interference signal is combined so that the sum of the differential signals is zero, whereby the differential signals of the desired signal included in the reception signal are additively mixed, and adaptive processing for removing the interference signal or noise is performed.

Specifically, when the time difference for calculating the differential signals is Δ t, the desired signal (complex representation) input from the antenna i at time t is xi (t), the interference signal (complex representation) input from the antenna i at time t is yi (t), and the weighted value of the antenna i is Wi, it is sufficient to obtain Wi (i.e., | sd (t) where the amplitude of the sum sd (t) of the differential signals of the desired wave signals received by the plurality of antennas is the largest and Wi (i.e., | sa (t)) where the amplitude of the sum sa (t) of the differential signals of the interference signals received by the antennas is the smallest (i.e., | sa (t)) | is the smallest Wi). Wherein the content of the first and second substances,

equation 41

SD(t)＝∑{WiXi(t)-WiXi(t-Δt)}

SA(t)＝∑{WiYi(t)-WiYi(t-Δt)}

Here, when the transmission environment is always constant and the change rate of the transmission environment is sufficiently delayed from the differential time difference Δ t, the amplitude of sd (t) is maximum and the amplitude of sa (t) is minimum when the amplitude of Σ wixi (t) is maximum and the amplitude of Σ wiyi (t) is minimum. On the other hand, the maximum amplitude of sd (t) and the minimum amplitude of sa (t) are limited to the maximum amplitude of Σ wixi (t) and the minimum amplitude of Σ wiyi (t) in any of xi (t) and yi (t). Therefore, the optimal weight value Wi based on the differential signal coincides with the optimal weight value Wi based on the non-differential signal.

In view of the above properties, the optimal weighting values in the embodiment of fig. 1 are consistent with the weighting values in the non-differentiated structure of fig. 21. A transmitting side weight value is determined based on the optimized weight value Wi, and a transmission signal is weighted by a transmission weighter B-9 and transmitted from each antenna, thereby realizing a transmitting/receiving adaptive array antenna. The above relationship also provides the same effect in the LMS algorithm that minimizes the average square error or the RLS algorithm that uses the conventional value for regression.

With such a configuration, the following effects are obtained as compared with the conventional example. Fig. 2(a) shows signal points in the IQ plane in the case of receiving a signal by QPSK modulation with white circles, and shows a shift from one signal point on the I axis to the next signal point with an arrow. The displacement rate in the IQ plane is determined by modulation of minimum phase transition, a rate of fall of a signal, or the like. Here, when there is no carrier frequency difference between the received signal and the reference signal, the signal point of the desired signal included in the received signal coincides with the signal point of the reference signal (when the amplitudes are equal), and the correlation value between these two signals is the largest. On the other hand, when there is a doppler frequency shift of the transmission path or a frequency difference of the reference clock between the base station and the mobile station, the signal point of the white circle shifts to a signal point of a shaded circle as shown in fig. 2 (b). Here, when the symbol rate is fs and the frequency difference is Δ f, the amount θ of phase point rotation between 1 symbols is expressed by 2 π Δ f/fs radians. Here, in the case where the 1 st training data interval is N1 radians and N1 · θ is a numerical value that cannot be ignored compared to 2 pi radians, in the conventional method, the phase error of the transmission path estimation value and the reference signal is N1 · θ at the maximum in the 1 st training data interval, the cross-correlation value does not match the autocorrelation value, and compared to the result that the transmission path estimation value includes an error, in the aspect of the present invention, even if the differentiation time Δ t is equal to the symbol time width, the phase error is an integral multiple of θ because only the change of θ is not generated and only the phase difference of a certain value θ is generated in the 1 st training interval, and the phase error is an integer multiple of θ with the passage of the symbol. Therefore, it is not necessary to find the estimation of the transmission path by the conventional method by extrapolation based on the regression line, and the transmission path is constant regardless of the frequency difference and can be easily calculated by averaging the differential results. Here, although the differential time difference is set to the symbol time, by setting the value to be equal to or less than this (for example, to 1/M), the phase error of each differential signal becomes θ/M, and the transmission path can be set more accurately.

Here, as a known method of removing the frequency error, there is a method using Automatic Frequency Control (AFC). That is, the method of detecting the rotation of the signal point based on the frequency difference, changing the frequency of the local transmitter of the base station so that the rotation of the signal point becomes 0, and deleting the rotation of the signal point. However, in the AFC, in order to accurately obtain the frequency difference, there is a problem that the convergence time of the frequency difference compensation is long, and the high-precision adaptive antenna operation cannot be performed during the convergence time. On the contrary, according to the method of the present invention, there is an advantage that it is possible to perform adaptive processing without operating AFC in particular, and it is possible to perform data demodulation prior to data demodulation without using a signal subsequent to an adaptive array antenna circuit for estimating a signal point transition accompanying data demodulation, and to perform data demodulation after improving a signal-to-interference ratio.

In particular, since the method of the present invention is similar to the delay detection method, compared to the synchronous detection method in which the reference signal is synchronized using AFC and then processed, there is some deterioration in the transmission path in a stable state, that is, in a fixed transmission path, compared to the synchronous method. However, in the case of mobile communication, the method of the present invention is effective in the case where the transmission path changes with time and carrier synchronization is not easily achieved.

Further, although the delay element B-3-1, the reception weighting unit B-4, the adder B-5, the reference signal differential unit B-6, the reception adaptive processing unit B-7, the transmission adaptive processing unit B-8, the transmission weighting unit B-9, and the like in the reception signal differential B-3 and the reception signal differential B-3 may be constituted by independent elements, respectively, the same effect can be obtained even when the DSP or the like performs software processing or when the DSP or the like is constituted by general-purpose logic elements such as an FPGA or the like, needless to say, the same effect can be obtained even when the DSP or the like performs the same function. In fig. 1, the reference signal is differentiated by the reference signal differentiator B-6 as an input signal, but it goes without saying that a signal obtained by differentiating the reference signal may be stored in a memory or the like and used.

In the configuration of fig. 1, it is needless to say that functional blocks for performing frequency conversion, which is an original purpose, such as a transmission unit and a reception unit are necessary, but illustration and description thereof are omitted because they are not directly related to the purpose of describing the present invention.

Claims (13)

1. An array antenna communication device that changes antenna directivity in real time and transmits and receives signals to and from a plurality of terminals in a time-division manner, the array antenna communication device comprising:

a plurality of antennas arranged in a dispersed manner; and

a transmitting circuit and a receiving circuit sharing the plurality of antennas in performing transmission and reception of signals,

the receiving circuit includes:

a reception signal separation circuit configured to separate a signal from a specific terminal among the plurality of terminals based on signals from the plurality of antennas when receiving a reception signal; and

a reception channel averaging circuit that averages a channel from the specific terminal based on a time difference signal obtained by time-differentiating a signal obtained by sampling signals from the plurality of antennas when the reception signal is received,

the transmission circuit includes:

a transmission path setting circuit that sets a transmission path for transmitting a transmission signal according to an average result of the reception transmission path averaging circuit; and

and a transmission directivity control circuit that updates the antenna directivity when the transmission signal is transmitted, based on a setting result of the transmission channel setting circuit.

2. The array antenna communication device according to claim 1, wherein:

an uplink slot of the transceived signal from the specific terminal contains a training data region of a prescribed size of the uplink slot,

the reception transmission path averaging circuit derives an average value of transmission paths from the specific terminal based on the time-differential signal in the training data region,

the transmission channel setting circuit predicts a transmission channel when the transmission signal is transmitted, from the average value.

3. The array antenna communication device according to claim 2, wherein:

the reception transmission path averaging circuit derives a reception coefficient vector corresponding to the impulse response from the specific terminal from the transmission path from the specific terminal based on the time-difference signal in the training data region.

4. The array antenna communication device according to claim 3, wherein:

the reception transmission path averaging circuit derives the reception coefficient vector by averaging the time-difference signals of the reception signals from the plurality of antennas and the time-difference signal from the specific terminal separated by the reception signal separating circuit.

5. The array antenna communication device according to claim 1, wherein:

an uplink slot of the transmission/reception signal from the specific terminal includes a training data region having a predetermined number of training data and a data region having a plurality of data indicating information from the specific terminal, respectively, which are provided in the uplink slot,

the reception transmission path averaging circuit derives a plurality of averages arranged in time series of the transmission path from the specific terminal based on the time differential signals in the training data region and the data region,

the transmission channel setting circuit sets a transmission channel for transmitting the transmission signal by using the plurality of average values.

6. The array antenna communication device according to claim 5, wherein:

the reception transmission path averaging circuit sequentially derives a plurality of reception coefficient vectors corresponding to the impulse response from the specific terminal from the transmission path from the specific terminal based on the plurality of time-difference signals in the training data region and the data region.

7. The array antenna communication device according to claim 6, wherein:

the sequential derivation of the plurality of receive coefficient vectors is based on a fastest descent method.

8. The array antenna communication device according to claim 6, wherein:

the sequential derivation of the plurality of receive coefficient vectors is based on recursive minimum 2 multiplication.

9. The array antenna communication device according to claim 1, wherein:

an uplink slot of the transmission/reception signal from the specific terminal includes a training data region having a predetermined number of training data and a data region having a plurality of data indicating information from the specific terminal, respectively, which are provided in the uplink slot,

the reception transmission path averaging circuit derives a plurality of predicted values of the transmission path from the specific terminal based on the time-difference signals in the training data area and the data area,

the transmission path setting circuit predicts a transmission path when the transmission signal is transmitted by regressing the plurality of predicted values and extrapolating according to a regression result.

10. The array antenna communication device according to claim 9, wherein:

the reception transmission path averaging circuit sequentially derives a plurality of reception coefficient vectors corresponding to the impulse response from the specific terminal from the transmission path from the specific terminal, based on the plurality of time-difference signals in the training data region and the data region.

11. The array antenna communication device according to claim 10, wherein:

the sequential derivation of the plurality of receive coefficient vectors is based on a fastest descent method.

12. The array antenna communication device according to claim 10, wherein:

the sequential derivation of the plurality of receive coefficient vectors is based on recursive minimum 2 multiplication.

13. The array antenna communication device according to claim 1, wherein:

the received signal separating circuit includes: a reception weight vector calculation unit that receives reception signals from the plurality of antennas and derives a reception weight vector for separating the time-difference signal from the specific terminal in real time; a plurality of 1 st multipliers, each of which receives the reception time difference signals from the plurality of antennas at one input and receives an element corresponding to the reception weight vector at the other input; and an adder adding signals from the plurality of 1 st multipliers,

the transmission directivity control circuit includes: a transmission weight vector calculation unit that derives a transmission weight vector based on a prediction result from the transmission channel averaging circuit; and a plurality of 2 nd multiplying units which receive a transmission signal at one input, receive the transmission weight vector at the other input, and supply the transmission weight vector to the plurality of antennas.