JP2001136741A - Switching power supply circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To improve a power conversion efficiency and simplify a circuit, while the efficiency improvement which is achieved by providing a large capacity and low resistance current-limited resistor in an AC input line and providing a circuit which short-circuits the current limited resistor and composed of an electromagnetic relays, thyristor-TRIACs, etc., is not necessary. SOLUTION: In addition to a power factor improvement circuit, which feeds a switching output back to a rectified current path by electrostatic coupling or magnetic coupling and supplies a current intermittently, to improve the power factor, a soft switching means using a transistor is provided, in order to limit a rush current at power supply closing.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply circuit provided as a power supply for various electronic devices, and more particularly to a switching power supply circuit having a configuration for improving a power factor.

[0002]

2. Description of the Related Art The applicant has previously proposed various power supply circuits having a resonance type converter on the primary side. In addition, various types of power supply circuits including a power factor improvement circuit for improving a power factor of a current resonance type converter have been proposed.

FIG. 10 is a circuit diagram showing an example of a switching power supply circuit constructed based on the invention previously filed by the present applicant. This power supply circuit has a configuration in which a power factor improvement circuit for improving a power factor is provided for a self-excited current resonance type switching converter.

The power supply circuit shown in FIG. 1 includes a bridge rectifier circuit Di for full-wave rectification of a commercial AC power supply AC. In this case, the rectified output rectified by the bridge rectification circuit Di is charged to the smoothing capacitor Ci via the power factor correction circuit 20, and both ends of the smoothing capacitor Ci correspond to the level of one time of the AC input voltage VAC. The rectified smoothed voltage Ei is obtained. Also, in the rectifying / smoothing circuit (Di, Ci), an inrush current limiting resistor Ri is inserted in the rectification current path so as to suppress an inrush current flowing into the smoothing capacitor when the power is turned on, for example. ing. The switch PS is a power-on switch.

In the power factor correction circuit 20 shown in FIG. 1, a filter choke coil LN, a high-speed recovery type diode D1 and a choke coil LS are provided between a positive output terminal of a bridge rectifier circuit Di and a positive terminal of a smoothing capacitor Ci. They are connected in series and inserted. The filter capacitor CN is inserted between the anode side of the high-speed recovery diode D1 and the positive terminal of the smoothing capacitor Ci to form a normal mode low-pass filter together with the filter choke coil LN.

In the power factor correction circuit 20, an end of a primary side series resonance circuit, which will be described later, is connected to a connection point between the cathode of the high-speed recovery type diode D1 and the choke coil LS. The switching output obtained in the series resonance circuit is fed back. The power factor improving operation by the power factor improving circuit 20 will be described later.

This power supply circuit includes a self-excited current resonance type converter using a rectified smoothed voltage Ei, which is a voltage across a smoothing capacitor Ci, as an operating power supply. In this current resonance type converter, as shown in the figure, switching elements Q1 and Q2 of two bipolar transistors are half-bridge-coupled and then inserted between the connection point on the positive side of the smoothing capacitor Ci and ground. It is connected. Starting resistors RS1 and RS2 are inserted between the collectors and the bases of the switching elements Q1 and Q2, respectively. The resistors RB1 and RB2 connected to the bases of the switching elements Q1 and Q2 set the base current (drive current) of the switching elements Q1 and Q2. Further, clamp diodes DD1 and Q2 are connected between the base and the emitter of the switching elements Q1 and Q2, respectively.
DD2 is inserted. The clamp diodes DD1 and DD2 are
A current path for a clamp current flowing through the base-emitter is formed during a period in which the switching elements Q1 and Q2 are turned off. And the resonance capacitors CB1, C
B2 is a drive winding N of a drive transformer PRT described below.
A series resonance circuit (self-excited oscillation drive circuit) for self-excited oscillation is formed together with B1 and NB2, and the switching elements Q1, Q2
Determine the switching frequency of 2.

[0008] Drive transformer PRT (Power Regulati
ng Transformer) is provided to drive the switching elements Q1 and Q2 and to perform constant voltage control by variably controlling the switching frequency. In this case, the drive windings NB1 and NB2 and the resonance current detection are performed. Winding N
D is wound, and a control winding N is provided for each of these windings.
C is an orthogonal saturable reactor wound in the orthogonal direction. One end of the drive winding NB1 of the drive transformer PRT is connected to the base of the switching element Q1 via a series connection of the resistor RB1 and the resonance capacitor CB1, and the other end is connected to the emitter of the switching element Q1. One end of the drive winding NB2 is grounded, and the other end is connected to the base of the switching element Q2 via a series connection of a resistor RB2 and a resonance capacitor CB2. The drive winding NB1 and the drive winding NB2 are wound so that voltages of opposite polarities are generated.

Insulated converter transformer PIT (Power Is
olation Transformer) is the switching element Q1, Q2
Is transmitted to the secondary side. One end of the primary winding N1 of the insulation converter transformer PIT is connected to the contact point (switching output point) between the emitter of the switching element Q1 and the collector of the switching element Q2 via the resonance current detection winding ND, thereby providing a switching output. Is obtained.

The other end of the primary winding N1 is connected to a connection point between the cathode of the high-speed recovery type diode D1 and the choke coil LS in the power factor correction circuit 20 via the series resonance capacitor C1. ing.

In this case, the series resonance capacitor C1 and the primary winding N1 are connected in series, but the capacitance of the series resonance capacitor C1 and the primary winding N1 are connected.
Converter transformer PIT including (series resonance winding)
The primary side series resonance circuit for making the operation of the switching converter a current resonance type is formed by the leakage inductance (leakage inductance) component.

On the secondary side of the insulated converter transformer PIT shown in FIG. 1, a center tap is provided for the secondary winding N2, and the rectifier diodes DO1, DO2, D
By connecting O3, DO4 and smoothing capacitors CO1, CO2 as shown in the figure, a set of [rectifier diodes DO1, DO2, smoothing capacitor CO1] and a set of [rectifier diodes DO3, DO4, smoothing capacitor CO2] are obtained. Two sets of full wave rectifier circuits are provided. The full-wave rectifier circuit composed of [rectifier diodes DO1, DO2, smoothing capacitor CO1] has a DC output voltage EO1.
And a full-wave rectifier circuit composed of [rectifier diodes DO3, DO4, smoothing capacitor CO2] generates a DC output voltage EO2. In this case, the DC output voltage EO1 and the DC output voltage EO2 are also branched and input to the control circuit 1. In the control circuit 1, the DC output voltage EO1 is used as a detection voltage, and the DC output voltage EO2 is used as an operation power supply of the control circuit 1.

The control circuit 1 supplies a DC current whose level varies in accordance with the level of the DC voltage output EO1 on the secondary side to the control winding NC of the drive transformer PRT as a control current, as will be described later. The constant voltage control is performed as described above.

In the switching operation of the power supply circuit having the above configuration, when a commercial AC power supply is first turned on by the switch PS, a starting current is supplied to the bases of the switching elements Q1 and Q2 via the starting resistors RS1 and RS2, for example. That is, for example, if the switching element Q1 is turned on first, the switching element Q2 is controlled to be turned off. As an output of the switching element Q1, a resonance current flows through the resonance current detection winding ND → the primary winding N1 → the series resonance capacitor C1, and when the resonance current becomes zero, the switching element Q2 is turned on and the switching element Q1 is turned on. It is controlled to be off. Then, a resonance current flows in the opposite direction through the switching element Q2. Thereafter, a self-excited switching operation in which the switching elements Q1 and Q2 are turned on alternately is started. As described above, the switching elements Q1 and Q2 are alternately opened and closed by using the terminal voltage of the smoothing capacitor Ci as the operating power supply.
A drive current close to the resonance current waveform is supplied to the primary winding N1, and an alternating output is obtained to the secondary winding N2.

The constant voltage control by the drive transformer PRT is performed as follows. For example, assuming that the secondary output voltage EO1 fluctuates so as to increase with the AC input voltage level or load fluctuation, the level of the control current flowing through the control winding NC as described above is also changed to the secondary output voltage EO1.
Is controlled so as to increase in accordance with the rise of. Under the influence of the magnetic flux generated in the drive transformer PRT by this control current, the drive transformer PRT tends to approach a saturation state, and acts to reduce the inductance of the drive windings NB1 and NB2. The switching frequency is controlled so as to increase by changing the condition of the oscillation circuit. In this power supply circuit, the switching frequency is set in a frequency region higher than the resonance frequency of the series resonance circuit of the series resonance capacitor C1 and the primary winding N1 (upper side control). Then, the switching frequency is separated from the resonance frequency of the series resonance circuit.
Thereby, the resonance impedance of the series resonance circuit with respect to the switching output increases.

By increasing the resonance impedance in this way, the drive current supplied to the primary winding N1 of the primary series resonance circuit is suppressed, and as a result, the secondary output voltage is suppressed. , Constant voltage control is achieved. Hereinafter, the constant voltage control method using the above method will be referred to as “switching frequency control method”.

The power factor improving operation by the power factor improving circuit 20 is as follows. In the configuration of the power factor correction circuit 20 shown in this figure, the switching output supplied to the series resonance circuit (N1, C1) is converted into a rectified current by a so-called magnetic coupling type through an inductive reactance assumed to be included in the choke coil LS itself. Return to the path.

The switching output fed back as described above causes the alternating voltage of the switching cycle to be superimposed on the rectified current path. The superimposed voltage of the switching cycle causes the high speed recovery type diode D1 to be superimposed. In this case, an operation of interrupting the rectified current in the switching cycle is obtained, and the apparent inductance of the filter choke coil LN and choke coil LS also increases due to the intermittent action. This allows the charging current to flow to the smoothing capacitor Ci even during a period in which the rectified output voltage level is lower than the voltage across the smoothing capacitor Ci. As a result, the average waveform of the AC input current is made closer to the waveform of the AC input voltage, and the conduction angle of the AC input current is increased. As a result, the power factor is improved.

FIG. 11 is a circuit diagram showing another example of the configuration of a switching power supply circuit that can be configured based on the invention previously proposed by the present applicant. This power supply circuit is also provided with a current resonance type converter in which two switching elements are half-bridge-coupled, and the drive system is separately excited. Also in this case, a power factor improving circuit for improving the power factor is provided. Note that the same parts as those in FIG. 10 are denoted by the same reference numerals and description thereof will be omitted.

The primary-side current resonance type converter shown in FIG. 1 includes two switching elements Q11 and Q12, for example, MOS-FETs. here,
By connecting the drain of the switching element Q11 to the line of the rectified and smoothed voltage Ei, connecting the source of the switching element Q11 and the drain of the switching element Q12, and connecting the source of the switching element Q12 to the primary side ground, it is compatible with the separately excited type. A half-bridge connection is obtained. These switching elements Q11 and Q12 are switching-driven by the oscillation drive circuit 2 so that on / off operations are alternately repeated, and intermittently output the rectified smoothed voltage Ei as a switching output. In this case, a clamp diode DD connected between the drain and source of each of the switching elements Q11 and Q12 in the direction shown in FIG.
1, DD2 are provided.

In this case, the switching element Q
11. By connecting one end of the primary winding N1 of the isolated converter transformer PIT to the source-drain connection point (switching output point) of Q12, the switching output is supplied to the primary winding N1. Is done. The other end of the primary winding N1 is connected via a series resonance capacitor C1 to a filter choke coil LN of a power factor improving circuit 21 described below.
And the anode of the high speed recovery type diode D1.

Also in this case, a series resonance circuit for making the switching power supply circuit a current resonance type is formed by the capacitance of the series resonance capacitor C1 and the leakage inductance component of the insulating converter transformer PIT including the primary winding N1. .

In this case, the control circuit 1 outputs a control signal having a level corresponding to the fluctuation of the DC output voltage EO1 to the oscillation drive circuit 2, for example. The oscillation drive circuit 2 changes the frequency of the switching drive signal supplied from the oscillation drive circuit 2 to each gate of the switching elements Q11 and Q12 based on the control signal supplied from the control circuit 1,
The switching frequency is made variable. Also, in the power supply circuit shown in this figure, similarly to the power supply circuit shown in FIG. 10, the switching frequency is set as a region higher than the series resonance frequency. When the DC output voltage EO1 rises, for example, the control circuit 1 controls the oscillation drive circuit 2 to increase the switching frequency according to the level. This allows
Constant voltage control is performed in the same manner as described with reference to FIG. The starting circuit 3 is provided for detecting the voltage or current obtained on the rectifying / smoothing line immediately after the power is turned on, and for starting the oscillation drive circuit 2. Is input as an operation power supply.

In the power factor correction circuit 21 shown in this figure, the positive output terminal of the bridge rectification circuit Di and the smoothing capacitor Ci
Of the filter choke coil LN−
A fast recovery diode D1 is inserted in series. Here, the filter capacitor CN is provided in parallel with the series connection circuit of the filter choke coil LN and the high-speed recovery type diode D1. Also in this connection form, the filter capacitor CN forms a normal mode low-pass filter together with the filter choke coil LN. The resonance capacitor C3
Are provided in parallel with the high-speed recovery type diode D1. Although a detailed description is omitted here, for example, the resonance capacitor C3 forms a parallel resonance circuit together with, for example, the filter choke coil LN and the like, so that its resonance frequency is substantially equal to the resonance frequency of the series resonance circuit described later. Is set. This has the effect of suppressing an increase in the rectified smoothed voltage Ei when the load is reduced. As described above, the power factor improving circuit 21 is connected to the end of the series resonance circuit (N1, C1) with respect to the connection point between the filter choke coil LN and the anode of the fast recovery type diode D1. Is connected.

In such a connection form, the switching output obtained on the primary winding N1 is connected to the series resonance capacitor C1.
The switching output is fed back to the rectified current path through the capacitive coupling of the switch. In this case, the switching output is applied to the connection point between the filter choke coil LN and the anode of the fast recovery diode D1 so that the resonance current obtained in the primary winding N1 flows. become.

As the switching output is fed back as described above, the alternating voltage of the switching cycle is superimposed on the rectified current path. In the diode D1, an operation of interrupting the rectified current in the switching cycle is obtained, and the apparent inductance of the filter choke coil LN also increases due to the interrupting action. Further, a voltage is generated at both ends of the resonance capacitor C3 when a current of a switching period flows through the resonance capacitor C3. However, the level of the rectified smoothed voltage Ei is reduced by the voltage across the resonance capacitor C3. Accordingly, even during the period when the rectified output voltage level is lower than the voltage across the smoothing capacitor Ci, the smoothing capacitor C
The charging current to i flows. As a result, the average waveform of the AC input current is made closer to the waveform of the AC input voltage, the conduction angle of the AC input current is expanded, and the power factor is also improved.

As described above, in the power supply circuits shown in FIGS. 10 and 11, the power factor can be improved by providing the power factor improvement circuits (20, 21). Since the power factor improving circuit is formed with a small number of components, it has the advantage that the power factor can be improved with high efficiency, low noise, small size, light weight, and low cost.

[0028]

Here, FIG. 12 shows the relationship between the AC input current IAC and the power supply circuit shown in FIGS. 10 and 11 when the switch PS is turned on at the peak of the AC input voltage VAC. The time change is shown. When the switch PS is turned on at the peak of the AC input voltage VAC,
Bridge rectifier circuit Di, filter choke coil LN,
An excessive rush charging current of 100 A or more flows to the smoothing capacitor Ci through the fast recovery type diode D1 and the choke coil LS, and exceeds the maximum allowable current of the bridge rectifier circuit Di, the high speed recovery type diode D1, and the smoothing capacitor Ci. May lead to destruction. For this reason, as shown in FIGS. 10 and 11, a large-capacity and low-resistance inrush current limiting resistor Ri is inserted into the AC line, and FIG.
As shown in FIG. 2, the inrush charging current is limited to 50 A or less.

However, when the load power is a heavy load of 200 W or more, the power loss increases and the efficiency decreases. AC
In the 100V system, the inrush current limiting resistor Ri is set to 1Ω / 30
As shown in FIG. 12, the AC input current IAC = 50 A at time t = 0 and the AC input current IAC = 5 at time t = 60 (msec) due to the large cement resistance of W, etc.
Although the state becomes the rated state of A, the power conversion efficiency decreases due to the power loss.

As a countermeasure, there is also a method in which the rush current limiting resistor Ri is short-circuited by an electromagnetic power relay or a semiconductor switch such as a thyristor or a triac one second after the switch PS is turned on, thereby reducing power loss. . However, in this case, by providing an electromagnetic power relay or a semiconductor switch and its peripheral circuit elements,
There are difficulties in terms of the number of components and cost.

FIG. 13 shows the relationship between the AC input voltage VAC and the power factor PF. Here, the maximum load power Pomax = 120 W and the minimum load power Pomin = 40
The characteristics under each condition at the time of W are shown. As shown in this figure, it can be seen that the power factor PF decreases proportionally as the AC input voltage VAC increases. As the power factor PF under the condition of the minimum load power Pomin = 40 W, the maximum load power Pomax = 12
The power factor is lower than 0W. That is, the power factor PF decreases due to fluctuations in the AC input voltage VAC and the load power Po. This is not a problem in home electric appliances in which the AC input voltage VAC and load conditions are specified, such as a color television receiver, but in office equipment and information equipment in which the AC input voltage VAC and load conditions may fluctuate. Means that the power supply circuit cannot be used.

The characteristics shown in FIG. 13 are shown as operating waveforms as shown in FIG. Here, FIG.
FIGS. 14A and 14B show the AC input voltage VAC and the AC input current IAC when the AC input voltage VAC = 100 V and the maximum load power Pomax = 120 W. FIGS. Minimum load power Pomin = 4 at voltage VAC = 100V
The AC input voltage VAC and the AC input current IAC at 0 W are shown. Assuming that the half cycle of the AC input voltage VAC is 10 ms, when the maximum load power Pomax is 120 W, the conduction period τ of the AC input current IAC is actually about 5 ms, and the power factor is PF = 0.85. . On the other hand, when the minimum load power Pomin = 40 W, the conduction period τ of the AC input current IAC is reduced to about 2.5 ms,
The power factor drops to about PF = 0.65. The value of the power factor PF obtained when the minimum load power Pomin = 40 W may not satisfy the value as the power factor required for practical use.

[0033]

In view of the above-mentioned problems, the present invention is configured as a switching power supply circuit as follows. In other words, a rectifying diode for inputting and rectifying commercial AC power and a rectifying / smoothing means for generating a rectified / smoothed voltage by a smoothing capacitor, and a gap are formed so as to obtain a required coupling coefficient that is loosely coupled, and a primary-side output is formed. An insulating converter transformer provided for transmission to a secondary side, switching means for intermittently outputting the rectified smoothed voltage by a switching element and outputting the rectified smoothed voltage to a primary winding of the insulating converter transformer, and at least the insulating converter transformer A primary-side parallel resonance circuit formed by a leakage inductance component including the primary winding and the capacitance of the parallel resonance capacitor and making the operation of the switching means a voltage resonance type; and magnetically coupling a switching output obtained in the primary winding. Or, it returns to the rectification current path by capacitive coupling and this feedback Power factor improving means for improving the power factor by intermittently commutating a rectified current based on the switching output, and a rush current to the smoothing capacitor which is arranged at a stage preceding the power factor improving means and which is turned on when the power is turned on. Is formed on the secondary side by a soft switch means formed by using a transistor so as to limit the leakage inductance component of the secondary winding of the insulated converter transformer and the capacitance of the secondary side resonance capacitor. A secondary-side resonance circuit and an alternating voltage formed including the secondary-side resonance circuit and obtained in a secondary winding of the insulated converter transformer are input and rectified to generate a secondary-side DC output voltage. DC output voltage generating means configured as described above, and a switching circuit of the switching means according to the level of the secondary DC output voltage. So as to comprise a constant voltage control means arranged to by variably controlling the number performs constant voltage control for the secondary side DC output voltage.

The soft switch means includes an NPN transistor connected to the positive electrode of the rectifier diode, a resistor arranged between the collector and base of the NPN transistor, and a resistor connected between the base of the NPN transistor and ground. And a capacitor arranged. Alternatively, the soft switch means includes a PNP transistor connected to a negative electrode side of the rectifier diode,
It is formed to have a resistor disposed between the collector and the base of the PNP transistor and a capacitor disposed between the base and the ground of the PNP transistor.

According to the above configuration, the inrush current at the time of turning on the power can be limited by the soft switch means using the transistor, whereby a large-capacity low-resistance current limiting resistor can be provided on the AC input line. It is not necessary to improve the efficiency by forming a circuit for short-circuiting the current limiting resistor by using an electromagnetic relay, a thyristor, or a triac.

[0036]

FIG. 1 is a circuit diagram showing a configuration of a switching power supply circuit according to a first embodiment of the present invention.

In the power supply circuit shown in this figure, a bridge rectification circuit Di is used as a rectification smoothing circuit for receiving a commercial AC power supply (AC input voltage VAC) and obtaining a DC input voltage.
And a full-wave rectifier circuit composed of a smoothing capacitor Ci, and generates a rectified smoothed voltage Ei corresponding to a level that is one time the AC input voltage VAC. The switch PS is a power-on switch. In this rectifying and smoothing circuit, the inrush current limiting resistor (Ri) as shown in FIGS. 10 and 11 is not inserted into the AC input line.
To suppress the inrush current flowing into the smoothing capacitor Ci when the power is turned on, a soft switch circuit 13 is provided between the positive electrode of the bridge rectifier circuit Di and the power factor correction circuit 10. Details will be described later.

A voltage resonance type switching converter (voltage resonance type converter) is provided on the primary side of the power supply circuit shown in FIG. A power factor improving circuit 10 is provided for this voltage resonance type converter. The configuration of the power factor improving circuit 10 will be described later, and first, the configuration of the voltage resonance type converter will be described. Here, the voltage resonance type converter employs a self-excited configuration including one switching element Q1. In this case, the switching element Q1 includes a high withstand voltage bipolar transistor (BJ).
T: junction type transistor).

The base of the switching element Q1 is connected to a smoothing capacitor Ci (rectified smoothing voltage E) via a starting resistor RS.
It is connected to the positive electrode side of i) so that the base current at the time of starting can be obtained from the rectification smoothing line. Between the base of the switching element Q1 and the primary side ground, a resonance circuit (self-excited oscillation drive circuit) for driving self-oscillation composed of a series connection circuit of a drive winding NB, a resonance capacitor CB, and a base current limiting resistor RB. Connected. The clamp diode DD inserted between the base of the switching element Q1 and the negative electrode (primary ground) of the smoothing capacitor Ci forms a path for a clamp current flowing when the switching element Q1 is off. Switching element Q1
Is connected to the positive terminal of the smoothing capacitor Ci through a series connection of the primary winding N1 and the detection winding ND. The emitter is grounded to the primary side ground.

Further, between the collector and the emitter of the switching element Q1, capacitors Cr1 and Cr2 connected in series are connected as parallel resonance capacitors. This parallel resonance capacitor Cr (Cr1, Cr
2) forms a primary parallel resonance circuit of a voltage resonance type converter by its own capacitance and a leakage inductance L1 on a primary winding N1 side of an insulating converter transformer PIT described later. Although the detailed description is omitted here, when the switching element Q1 is turned off, the resonance capacitor Cr (C
The voltage between both ends of r1, Cr2) is actually a sinusoidal pulse waveform so that a voltage resonance type operation can be obtained.

The orthogonal control transformer PRT shown in FIG.
Is a saturable reactor on which a detection winding ND, a drive winding NB, and a control winding NC are wound. The orthogonal transformer PRT drives the switching element Q1 and is provided for constant voltage control. As the structure of the orthogonal control transformer PRT, although not shown, a three-dimensional core is formed by joining the ends of the two magnetic legs of two double U-shaped cores having four magnetic legs. I do. And
The detection winding ND and the drive winding NB are wound around the predetermined two magnetic legs of the three-dimensional core in the same winding direction, and the control winding NC is further connected to the detection winding ND and the drive winding ND. It is configured by being wound in a direction orthogonal to the line NB.

In this case, the detection winding ND of the orthogonal control transformer PRT (frequency variable means) is connected in series with the primary winding N1 of the insulation converter transformer PIT, which will be described later, so that the switching of the switching element Q1 is performed. The output is transmitted to the detection winding ND via the primary winding N1.
In the orthogonal control transformer PRT, the switching output obtained on the detection winding ND is excited by the drive winding NB via the transformer coupling, so that an alternating voltage as a drive voltage is generated on the drive winding NB. . This drive voltage is applied to a series resonance circuit (NB,
CB) is output to the base of the switching element Q1 as a drive current via the base current limiting resistor RB.
As a result, the switching element Q1 performs a switching operation at a switching frequency determined by the resonance frequency of the series resonance circuit (NB, CB).

Insulated converter transformer P of the present embodiment
As shown in FIG. 2, the IT is provided with an EE-type core in which E-type cores CR1 and CR2 made of, for example, a ferrite material are combined so that their magnetic legs face each other. , The primary winding N using the divided bobbin B
1 and the secondary winding N2 are wound separately. As shown in the figure, the gap G
Is formed. As a result, loose coupling with a required coupling coefficient can be obtained. The gap G can be formed by forming the central magnetic legs of the E-shaped cores CR1 and CR2 shorter than the two outer magnetic legs.
The coupling coefficient k is set to a loosely coupled state of, for example, k ≒ 0.85, so that a saturated state is hardly obtained.

One end of the primary winding N1 of the insulating converter transformer PIT is connected to the collector of the switching element Q1, and the other end is connected to the positive electrode (rectified smoothing voltage) of the smoothing capacitor Ci through a series connection of the detecting winding ND. Ei).

On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, the secondary parallel resonance capacitor C2 is connected in parallel to the secondary winding N2, so that the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary parallel resonance capacitor C2 are determined. A parallel resonance circuit is formed. With this parallel resonance circuit,
The alternating voltage excited by the secondary winding N2 becomes a resonance voltage.
That is, a voltage resonance operation is obtained on the secondary side.

That is, in this power supply circuit, the primary side is provided with a parallel resonance circuit for performing a voltage resonance type switching operation, and the secondary side is provided with a parallel resonance circuit for obtaining a voltage resonance operation. . In the present specification, such a switching converter configured to operate with a resonance circuit provided on the primary side and the secondary side is also referred to as a “composite resonance type switching converter”.

In this case, for the secondary side parallel resonance circuit formed as described above, a tap is provided for the secondary winding N2, and then the rectifier diodes DO1, DO2, DO3,
By connecting DO4 and the smoothing capacitors CO1 and CO2 as shown in the figure, two sets of a set of [rectifying diodes DO1, DO2, smoothing capacitor CO1] and a set of [rectifying diodes DO3, DO4, smoothing capacitor CO2] are provided. Is provided. A full-wave rectifier circuit composed of [rectifier diodes DO1, DO2, smoothing capacitor CO1] generates a DC output voltage EO1, and generates a [rectifier diode DO3, DO4, smoothing capacitor CO1].
O2] generates a DC output voltage EO2. In this case, the DC output voltage EO1 and the DC output voltage EO2 are also branched and input to the control circuit 1. In the control circuit 1, the DC output voltage EO1 is used as a detection voltage, and the DC output voltage EO2 is used as an operation power supply of the control circuit 1.

The control circuit 1 supplies, as a control current, a DC current whose level varies in accordance with the level of the DC voltage output EO1 on the secondary side to the control winding NC of the drive transformer PRT, as will be described later. The constant voltage control is performed as follows.

By the way, the insulation converter transformer PIT
, The polarity (winding direction) of the primary winding N1 and the secondary winding N2 and the rectifier diode DO (DO1, DO2, DO3, DO4)
, The mutual inductance M between the inductance L1 of the primary winding N1 and the inductance L2 of the secondary winding N2 may be + M or -M. For example, when the connection configuration shown in FIG. 3A is adopted, the mutual inductance is + M (additive polarity: forward method), and when the connection configuration shown in FIG. 3B is employed, the mutual inductance is −M (depolarization: Flyback method). If this is made to correspond to the operation of the secondary side of the power supply circuit shown in FIG. 1, for example, when the alternating voltage obtained in the secondary winding N2 has a positive polarity, the rectifier diode DO1 (DO3)
Can be regarded as an operation mode of + M (forward system). Conversely, when the alternating voltage obtained in the secondary winding N2 has a negative polarity, the rectifier diode DO2
The operation in which the rectified current flows through (DO4) can be regarded as the -M operation mode (flyback method). That is, in this power supply circuit, each time the alternating voltage obtained in the secondary winding becomes positive / negative, the mutual inductance becomes + M / -M
Mode.

The control circuit 1 varies the level of the control current (DC current) flowing through the control winding NC according to the change in the secondary-side DC output voltage level (EO1), thereby winding the control current to the orthogonal control transformer PRT. The inductance LB of the mounted drive winding NB is variably controlled. Thereby, the switching element Q1 formed including the inductance LB of the drive winding NB is formed.
The resonance condition of the series resonance circuit in the self-excited oscillation driving circuit for the above changes. This is an operation of varying the switching frequency of the switching element Q1, and this operation has an effect of stabilizing the secondary DC output voltage.

In the circuit shown in this figure, when the switching frequency is varied, the period during which the switching element Q1 is off is fixed, and the period during which the switching element Q1 is on is variably controlled. In other words, in this power supply circuit, as a constant voltage control operation, an operation is performed to variably control the switching frequency, thereby performing resonance impedance control on the switching output, and at the same time, controlling the conduction angle of the switching element in the switching cycle (PWM). Control). This complex control operation is realized by a set of control circuit systems. Here, as the switching frequency control, when the secondary output voltage increases due to, for example, a tendency toward light load, the control is performed so as to suppress the secondary output by increasing the switching frequency. It is supposed to be done.

Next, the configuration of the power factor improving circuit 10 will be described. In the power factor correction circuit 10 shown in FIG.
A high-speed recovery diode D1 and a choke coil LS are connected in series to a positive output terminal of the bridge rectifier circuit Di via a soft switch circuit 13. The end of the choke coil LS is connected to the positive terminal of the smoothing capacitor Ci. The filter capacitor CN is inserted between the anode side of the high-speed recovery diode D1 and the positive terminal of the smoothing capacitor Ci to form a normal mode low-pass filter.

In the power factor correction circuit 10, the connection of the series connected capacitors Cr1 and Cr2 constituting the parallel resonance capacitor to the connection point between the cathode of the high speed recovery type diode D1 and the choke coil LS is performed. The connection point is connected so that the switching output (voltage resonance pulse voltage) obtained in the primary side parallel resonance circuit is fed back.

The power factor improving operation by the power factor improving circuit 10 is basically as follows. In the configuration of the power factor correction circuit 10 shown in this figure, the switching output supplied to the primary parallel resonance circuit is fed back to the rectified current path via an inductive reactance (magnetic coupling) assumed to be included in the choke coil LS itself. To be.

The alternating voltage of the switching cycle is superimposed on the rectified current path by the switching output fed back as described above. The superimposed voltage of the switching cycle causes the high speed recovery type diode D1 to be superimposed. In this case, an operation of interrupting the rectified current in the switching cycle is obtained, and the apparent inductance of the choke coil LS also increases due to the interrupting action. This allows the charging current to flow to the smoothing capacitor Ci even during a period in which the rectified output voltage level is lower than the voltage across the smoothing capacitor Ci. As a result, the average waveform of the AC input current is made closer to the waveform of the AC input voltage, and the conduction angle of the AC input current is increased. As a result, the power factor is improved.

Here, in this example, as described above, the parallel resonance capacitors Cr forming the primary side parallel resonance circuit of the primary side voltage resonance type converter are capacitors Cr1 and Cr2.
Are connected in series, and the capacitors Cr1,
The connection point of Cr2 is connected to the cathode of the high-speed recovery type diode D1 of the power factor correction circuit 10. Therefore, the voltage resonance pulse voltage, which appears as the voltage across the resonance capacitor Cr (Cr1, Cr2),
1, a circuit system is formed as a voltage feedback system in which the voltage is divided by the capacitance ratio of Cr2 and fed back to the connection point between the high-speed recovery type diode D1 and the choke coil LS.

The capacitance of the capacitors Cr1 and Cr2 is
It is assumed that Cr1 <Cr2. In particular, when the capacitance of the capacitor Cr2 is increased, the power factor PF is improved. That is, during the period when the AC input voltage VAC is high, the switching frequency fs is controlled to be high, and the AC input voltage VAC
Is low, the switching frequency fs is controlled to be low. Therefore, near the peak value of the AC input voltage VAC, the voltage resonance pulse voltage is not fed back to the power factor correction circuit 10, and the AC input current IAC from the AC power supply AC is Rectifier circuit Di → soft switch circuit 13 → high-speed recovery type diode D1 → smoothing capacitor C via choke coil LS
i is charged. Then, as the AC input voltage VAC decreases, the feedback amount of the voltage resonance pulse voltage to the power factor improvement circuit 10 increases.

From this, the operation waveforms of the AC input voltage VAC, the AC input current IAC, the divided voltage V2 by the capacitors Cr1 and Cr2, and the current ILS flowing through the choke coil LS are shown in FIGS. 4 (a), 4 (b) and 4 (e). As shown in FIG. For example, here, the frequency of the commercial power supply is 50 Hz, and the AC input voltage VAC has a half cycle of 10 as shown in FIG.
A sinusoidal waveform of ms is obtained. When an AC input current IAC flows as shown in FIG. 4B as a rectified current flowing through the bridge rectifier circuit Di, the high-speed recovery diode D1 switches so as to intermittently switch the choke coil LS. As shown in FIG. 4F, the switching current ID1 and the feedback current I
The alternating current ILS of the switching cycle in which the two are combined flows. At this time, due to the series resonance of the capacitor Cr2 and the choke coil LS, the operation waveforms of the current ILS flowing through the choke coil LS and the feedback current I2 being voltage-feedback become sinusoidal.

The soft switch circuit 13 includes an NPN transistor Q3, a resistor R1, and a small-capacity electrolytic capacitor C.
10. It is composed of a diode D3. The collector of the transistor Q3 is connected to the positive electrode of the bridge rectifier circuit Di, and the emitter of the transistor Q3 is connected to the power factor correction circuit 10. The resistor R1 is arranged between the collector and the base of the transistor Q3, and the capacitor C10 is arranged between the base and the primary side ground of the transistor Q3. A diode D3 is connected between the emitter and the base of the transistor Q3 to prevent a reverse voltage of the base-emitter voltage VBE.

In the steady state, transistor Q3 is on, and collector current Ic continues to flow during the positive and negative periods of AC input voltage VAC, and the collector current
The emitter voltage VCE is in a saturation voltage state. The resistance value of the resistor R1 is selected so that Ic <hFE.multidot. (Base current IB). In a steady state where the transistor Q3 is turned on, the collector-emitter voltage VCE and the collector current Ic of the transistor Q3 are as shown in FIGS.

FIG. 5 shows that the switch PS operates when the AC input voltage V
The state of the change of the AC input current IAC when it is turned on at the peak of AC is shown. The resistance value of the resistor R1 is such that the ASO region of the transistor Q3 has a collector current Ic <16A.
, The collector current Ic is determined to be 15A. The capacitor C10 is composed of the resistor R1 and the capacitor C
The time constant of 10 has an effect of delaying the rise of the AC input current IAC after the switch PS is turned on. Therefore, the soft start function by the soft start circuit 13 works, and as shown in FIG.
, The AC input current IAC does not flow to the smoothing capacitor Ci as an excessive rush charging current. Therefore, even if a current limiting resistor is not provided in the AC input line as in the present embodiment, the AC input current IAC does not flow. The input current IAC does not exceed the maximum allowable current of the bridge rectifier circuit Di, the fast recovery type diode D1, and the smoothing capacitor Ci.

That is, since the large-capacity, low-resistance rush current limiting resistor Ri inserted into the AC line as shown in FIGS. 10 and 11 becomes unnecessary, the power loss due to the rush current limiting resistor Ri is eliminated. Thus, the power conversion efficiency is improved. In addition, naturally, a circuit portion for reducing the power loss by short-circuiting the rush current limiting resistor Ri by an electromagnetic power relay or a semiconductor switch such as a thyristor or a triac when one second elapses after the switch PS is turned on becomes unnecessary. .

As a result of an experiment conducted on such a switching power supply circuit as shown in FIG.
In the range of 0 W to 140 W as o, the power factor (PF) increases as the load power Po decreases. Especially Po
= 60 W or less, a high value of power factor PF = 0.8 or so was obtained. When the relationship between the fluctuation of the AC input voltage VAC and the power factor under the condition that the load power Po is constant at 140 W is considered, the power factor PF is expressed as the AC input voltage V
It was found that the power factor (PF) was maintained almost constant in the range of AC = 80 V to 140 V.

As described above, the power supply circuit according to the present embodiment has such characteristics that the power factor becomes almost constant with respect to the fluctuation of the AC input voltage, and the power factor rises as the load power decreases. Can be For this reason, the power supply circuit according to the present embodiment is not limited to a television receiver in which an AC input voltage or a load condition is specified, but is mounted on, for example, office equipment or a personal computer in which the load condition varies. Is practically sufficient.

The secondary side DC output voltage level EO1 of 50
The Hz ripple voltage component is also increased only about twice as much as that when the power factor correction circuit 10 is not provided, and is in a range where there is no practical problem as a power supply circuit used for a color television or the like. The operation of the power factor correction circuit 10 is low noise because a smooth sinusoidal waveform is obtained. Further, in practice, the choke coil LS has a small inductance value, so that a small and light element can be selected, so that the circuit can be reduced in size and weight and cost can be reduced.

Next, a second embodiment of the present invention will be described with reference to FIG. FIG. 6 is a circuit diagram showing a configuration of a power supply circuit according to a second embodiment of the present invention. In this figure, the same parts as those in FIG. The insulating converter transformer PIT shown in this figure has the same structure as that shown in FIG.

In this figure, the voltage resonance type converter provided on the primary side has a separately-excited configuration, and is provided with, for example, a single MOS-FET switching element Q21. The drain of the switching element Q21 is connected to the positive electrode of the smoothing capacitor Ci via the primary winding N1,
The source is connected to the primary ground. Also in this case, the parallel resonance capacitor Cr is also composed of the capacitors Cr1 and C
Although it is formed by the series connection of r2,
One end of the capacitor Cr1 is connected to the drain of the switching element Q21, and the other end is connected to a connection point between the high-speed recovery type diode D1 and the choke coil LS in the power factor correction circuit 10. Further, the capacitor Cr2 is connected in parallel with the fast recovery diode D1. A clamp diode DD is connected in parallel between the drain and source of the switching element Q21.

The switching element Q21 is driven by the oscillation / drive circuit 2 so that the switching operation described above with reference to FIG. 1 is obtained. That is, the control circuit 1 supplies a current or voltage of a level fluctuated according to the fluctuation of the secondary DC output voltage E01 to the oscillation / drive circuit 2. In the oscillation / drive circuit 2, a switching drive signal (voltage) whose cycle is varied in accordance with the output level from the control circuit 1 is switched by the switching element Q so as to stabilize the secondary side DC output voltage E01.
Output to 21 gates. Thereby, the switching frequency of the switching element Q21 is varied. In this case, as described in FIG. 1, the period during which the switching element Q21 is off is constant, and the period during which the switching element Q21 is on is varied. The switching drive signal generated as much as possible is output.

In this case, the rectified and smoothed voltage Ei obtained from the smoothing capacitor Ci is supplied to the starting circuit 3 as an operating power source.
Starting voltage obtained at winding N4 additionally wound on IT and rectified and smoothed by diode D2 and capacitor C3 causes start-up circuit 3 to generate oscillation / drive circuit 2
An operation for activating is executed.

The power factor improving circuit 10 shown in FIG.
It is the same as the power factor correction circuit 10 of FIG. 1 except that the capacitor Cr2 is connected in parallel with the fast recovery diode D1. Also in this case, the switching output supplied to the primary-side parallel resonance circuit is fed back to the rectified current path via the inductive reactance (magnetic coupling) assumed to be included in the choke coil LS itself.

With such a configuration, as in the example of FIG. 1, the parallel resonance capacitor Cr forming the primary side parallel resonance circuit of the primary side voltage resonance type converter is formed by connecting capacitors Cr1 and Cr2 in series. And
The connection point of the capacitors Cr1 and Cr2 is the power factor correction circuit 10.
Of the high speed recovery type diode D1. Therefore, the resonance capacitors Cr (Cr1, Cr
The voltage resonance pulse voltage, which appears as the voltage between the two ends, is divided by the capacitance ratio of the capacitors Cr1 and Cr2, and is fed back to the connection point between the high-speed recovery type diode D1 and the choke coil LS. A system has formed. The capacitances of the capacitors Cr1 and Cr2 are set to Cr1 <Cr2. As a result, similarly to the example described with reference to FIG. 1, a high power factor can be maintained with respect to fluctuations in the AC input voltage and load, and the present invention is not limited to a television receiver in which the AC input voltage and load conditions are specified. For example, it becomes practically sufficient for office equipment such as office equipment and personal computers in which the load condition varies.

In this example, the soft switch circuit 13A includes a PNP transistor Q4, a resistor R1,
It is composed of a small-capacity electrolytic capacitor C10 and a diode D3. The collector of the transistor Q4 is connected to the negative side of the bridge rectifier circuit Di. The resistor R1 is disposed between the collector and the base of the transistor Q4, and the capacitor C10 is disposed between the base and the primary side ground of the transistor Q4. A diode D3 is connected between the emitter and the base of the transistor Q4 to prevent a reverse voltage of the base-emitter voltage VBE.

That is, the soft switch circuit 13A
Has a configuration using a PNP transistor, and is inserted on the negative electrode side of the bridge rectifier circuit Di. The operation is the same as that of the soft switch circuit 13 of FIG. 1, and the rise of the AC input current IAC after the switch PS is turned on is delayed by the soft switch circuit 13A, so that the switch PS is turned on at the peak of the AC input voltage VAC. Even if it is, the AC input current IAC does not flow to the smoothing capacitor Ci as an excessive inrush charging current.
Therefore, it is not necessary to provide a current limiting resistor in the AC input line.

On the secondary side of the power supply circuit shown in FIG. 6, one end of the secondary winding N2 is connected to the secondary side ground, and the other end is connected to the rectifier diode via the series connection of the series resonance capacitor Cs1. It is connected to the connection point between the anode of DO1 and the cathode of rectifier diode DO2. The cathode of the rectifier diode DO1 is connected to the positive electrode of the smoothing capacitor CO1, and the anode of the rectifier diode DO2 is connected to the secondary side ground. The negative side of the smoothing capacitor CO1 is connected to the secondary side ground.

As a result, in such a connection form, a voltage doubler half-wave rectifier circuit comprising a set of [series resonance capacitor Cs, rectifier diodes DO1, DO2, smoothing capacitor CO1] is provided. Here, the series resonance capacitor Cs
Forms a series resonance circuit corresponding to the on / off operation of the rectifier diodes DO1 and DO2 by its own capacitance and the leakage inductance component of the secondary winding N2. That is, the power supply circuit of the present embodiment is provided with a parallel resonance circuit on the primary side for performing a voltage resonance type switching operation, and a series resonance circuit for obtaining a voltage doubled half-wave rectification operation on the secondary side. A complex resonance type switching converter provided with a circuit is employed.

Here, the above [series resonance capacitors Cs,
The double voltage half-wave rectification operation by the combination of the rectifier diodes DO1, DO2 and the smoothing capacitor CO1] is as follows. When a switching output is obtained on the primary winding N1 by the switching operation on the primary side, this switching output is excited by the secondary winding N2. The voltage doubler rectifier circuit performs a rectifying operation by inputting the obtained alternating voltage to the secondary winding N2. In this case, first, during the period when the rectifier diode DO1 is turned off and the rectifier diode DO2 is turned on, the primary winding N
The polarity (mutual inductance M) between 1 and the secondary winding N2 is-
M operates in the depolarization mode, and the operation of charging the rectified current rectified by the rectifier diode D02 to the series resonance capacitor Cs by the leakage inductance of the secondary winding N2 and the series resonance action of the series resonance capacitor Cs. can get.

During the period when the rectifier diode DO2 is turned off and the rectifier diode DO1 is turned on to perform the rectification operation, the polarity (mutual inductance M) of the primary winding N1 and the secondary winding N2 is + M. The operation is in the additive polarity mode, in which the smoothing capacitor C01 is charged in a state where series resonance occurs in which the potential of the series resonance capacitor Cs is added to the voltage induced in the secondary winding N2. As described above, the insulated converter transformer PIT operates in the positive polarity mode (+ M; forward operation) and the depolarization mode (-M).
M; flyback operation), the DC output voltage EO1 (rectified smoothed voltage) corresponding to almost twice the induced voltage of the secondary winding N2 in the smoothing capacitor CO1.
Is obtained. By performing the double voltage half-wave rectification in this way, the secondary side DC output voltage Eo1 is obtained.

In this embodiment, a secondary winding N2A is wound independently of the secondary winding N2, a center tap is grounded to the secondary winding N2A, and a rectifier diode A DC output voltage EO2 is generated by connecting a full-wave rectifier circuit composed of DO3, DO4 and a smoothing capacitor CO2.

FIG. 7 is a circuit diagram showing a configuration of a power supply circuit according to a third embodiment of the present invention. This is a modification of the power supply circuit of FIG. The same parts as those in FIG. In the example of FIG. 1, the soft switch circuit 13 is provided in a combination of the self-oscillation type switching frequency control type composite resonance type converter and the voltage feedback type magnetic coupling type power factor correction circuit. 7 is a combination of a self-oscillation type switching frequency control type composite resonance type converter and a voltage feedback type electrostatic coupling type power factor correction circuit, and a soft switch circuit 13.
This is an example in which is provided.

In this case, a parallel resonance capacitor Cr is connected to the collector of the switching element Q1. This parallel resonance capacitor Cr is connected to its own capacitance and the primary winding N1 of the insulation converter transformer PIT.
The primary side parallel resonance circuit of the voltage resonance type converter is formed by the leakage inductance L1 on the side. And
When the switching element Q1 is turned off, the operation of the parallel resonance circuit causes the voltage across the resonance capacitor Cr to be a sinusoidal pulse waveform, and a voltage resonance type operation is obtained. Insulating converter transformer P
One end of the IT primary winding N1 is connected to the collector of the switching element Q1, and the other end is connected to the positive electrode of the smoothing capacitor Ci (rectified smoothing voltage Ei) via a series connection of the detection winding ND.
Is connected to

In the power factor correction circuit 11, the soft switch circuit 13 is connected to the positive output terminal of the bridge rectifier circuit Di.
, A choke coil Ls and a high-speed recovery type diode D1 are connected in series and inserted. The filter capacitor CN is provided in parallel with the series connection circuit of the choke coil Ls and the high-speed recovery type diode D1, thereby forming a normal mode low-pass filter together with the choke coil Ls.

The parallel resonance capacitor C10 is provided in parallel with the high-speed recovery type diode D1. Although detailed description is omitted here, for example, the parallel resonance capacitor C10 forms a parallel resonance circuit together with, for example, the choke coil Ls. This has the effect of suppressing an increase in the rectified smoothed voltage Ei when the load is reduced. Further, for the power factor improving circuit 11, the above-described parallel resonance capacitor Cr is connected to a connection point of the choke coil Ls, the anode of the high-speed recovery type diode D1, and the parallel resonance capacitor C10. The switching output (voltage resonance pulse voltage) obtained in the side parallel resonance circuit is fed back.

The power factor improving operation by the power factor improving circuit 11 is basically as follows. In the configuration of the power factor correction circuit 11 shown in this figure, the switching output obtained in the primary side parallel resonance circuit is fed back to the rectification current path via the capacitive coupling of the parallel resonance capacitor Cr.

The switching output fed back in this way causes the alternating voltage of the switching cycle to be superimposed on the rectified current path. An operation of interrupting the rectified current at the switching cycle is obtained, and the apparent inductance of the choke coil Ls also increases due to the interrupting action. Further, a voltage is generated at both ends of the parallel resonance capacitor C10 when a current of a switching period flows through the capacitor.
The level of the rectified smoothed voltage Ei is reduced by the voltage across the parallel resonance capacitor C10. This allows the charging current to flow to the smoothing capacitor Ci even during a period in which the rectified output voltage level is lower than the voltage across the smoothing capacitor Ci. As a result, the average waveform of the AC input current is made closer to the waveform of the AC input voltage, and the conduction angle of the AC input current is increased. As a result, the power factor is improved.

As described above, the parallel resonance capacitor Cr forming the primary side parallel resonance circuit of the primary side voltage resonance type converter is connected to the cathode of the high speed recovery type diode D 1 of the power factor correction circuit 11. This is a state in which the parallel resonance capacitor Cr and the parallel resonance capacitor C10 are connected in series.
The voltage resonance pulse voltage, which appears as the voltage across
The voltage is divided by the capacitance ratio of the parallel resonance capacitor Cr and the parallel resonance capacitor C10. Then, a circuit system is formed as a voltage feedback system in which a voltage is fed back to the smoothing capacitor Ci via a parallel resonance capacitor C10 connected in parallel with the high-speed recovery type diode D1.

The capacitance of the parallel resonance capacitors Cr and C10 is Cr <C10. In particular, when the capacitance of the capacitor C10 is increased, the power factor PF is improved.
That is, the switching frequency fs is controlled to be high during the period when the AC input voltage VAC is high, and the switching frequency fs is controlled to be low during the period when the AC input voltage VAC is low. The resonance pulse voltage is not fed back to the power factor correction circuit 10, and the AC input current IAC from the AC power supply AC is supplied to the bridge rectifier circuit Di.
→ Soft switch circuit 13 → Choke coil Ls → Smoothing capacitor Ci via high speed recovery type diode D1
Is charged. Then, as the AC input voltage VAC decreases, the amount of feedback of the voltage resonance pulse voltage to the power factor improvement circuit 11 increases.

For the switching power supply circuit of this example, the maximum load power POmax = 140 W and the minimum load power POmin =
0 W, AC input voltage VAC = 80V-14
Switching frequency fs = 100K for 0V fluctuation
The experiment was performed under the control range of Hz to 200 KHz. As a result, in the state of the AC input voltage VAC = 100 V, as shown in FIG.
The power factor PF is approximately 0.
82, which was kept constant. Further, as shown in FIG. 8B, the AC input voltage VAC = 80V-
The load power PO = 140 with respect to the fluctuation in the range of 140 V
Under each condition of W to 20 W, almost the same power factor (PF = 0.
82).

As described above, in the power supply circuit of the present embodiment, a high power factor can be maintained even when the AC input voltage and the load fluctuate. For this reason, the power supply circuit according to the present embodiment is not limited to a television receiver or the like in which the AC input voltage or the load condition is specified, and is applied to office equipment such as an office equipment or a personal computer in which the load condition varies. It is practically possible to mount it.

Also, in the case of the power supply circuit of FIG.
Similarly, a soft switch circuit 13 is provided. Accordingly, since the large-capacity low-resistance inrush current limiting resistor Ri inserted into the AC line as shown in FIGS. 10 and 11 is not required, the power loss due to the inrush current limiting resistor Ri is eliminated, and the power conversion efficiency is reduced. Is improved. This also eliminates the need for a circuit section that short-circuits the inrush current limiting resistor Ri by using an electromagnetic power relay or a semiconductor switch such as a thyristor or a triac when one second elapses after the switch PS is turned on, thereby reducing power loss.

FIG. 9 is a circuit diagram showing a configuration of a power supply circuit according to a fourth embodiment of the present invention. This is a modification of the power supply circuit of FIG. The same parts as those in FIG. 6 are denoted by the same reference numerals, and description thereof will be omitted. In the example of FIG. 6, the soft switching circuit 13 is provided in a combination of the separately excited oscillation type switching frequency control type composite resonance type converter and the voltage feedback type magnetic coupling type power factor correction circuit. Reference numeral 9 denotes a combination of a separately excited oscillation type switching frequency control type composite resonance type converter and a voltage feedback type electrostatic coupling type power factor correction circuit, and a soft switch circuit 13.
This is an example in which A is provided. 1, 6, and 7 are denoted by the same reference numerals, and description thereof will be omitted.

In the case of FIG. 9, the parallel resonance capacitor Cr
Is connected to the drain of the switching element Q21, and the other end is connected to the high-speed recovery type diode D1, the choke coil LS, and the parallel resonance capacitor C10 in the power factor correction circuit 10.
Is connected to the connection point. A clamp diode DD is connected in parallel between the drain and source of the switching element Q21. The power factor improving circuit 11 shown in this figure is the same as the power factor improving circuit 11 shown in FIG.

With such a configuration, the parallel resonance capacitor Cr forming the primary side parallel resonance circuit of the primary side voltage resonance type converter is the same as the example of FIG. It is connected to the anode of D1. Therefore, the parallel resonance capacitor Cr and the parallel resonance capacitor C10 are connected in series, and the voltage resonance pulse voltage, which appears as the voltage across the parallel resonance capacitor Cr, is divided by the capacitance ratio between the parallel resonance capacitor Cr and the parallel resonance capacitor C10. Pressed. The smoothing capacitor C10 is connected via a parallel resonance capacitor C10 connected in parallel with the high-speed recovery type diode D1.
A circuit system as a voltage feedback system in which voltage is fed back to i is formed. The capacitance of the parallel resonance capacitors Cr and C10 is set to Cr <C10.

Thus, similar to the example described with reference to FIG.
High power factor can be maintained even with fluctuations in AC input voltage and load, and is not limited to television receivers and the like in which AC input voltage and load conditions are specified, such as office equipment and personal computers in which load conditions fluctuate It will be practically sufficient for office equipment.

In the case of the power supply circuit of FIG.
Similarly, a soft switch circuit 13A is provided.
Therefore, since the large-capacity, low-resistance inrush current limiting resistor Ri inserted into the AC line is not required, power loss due to the inrush current limiting resistor Ri is eliminated, and power conversion efficiency is improved. This also eliminates the need for a circuit section that short-circuits the inrush current limiting resistor Ri by using an electromagnetic power relay or a semiconductor switch such as a thyristor or a triac when one second elapses after the switch PS is turned on, thereby reducing power loss.

Although the embodiments of the present invention have been described above, various modifications are possible. The present applicant has already proposed a configuration provided with a quadruple voltage rectifier circuit using a secondary-side series resonance circuit as a complex resonance type switching converter, but such a configuration is also a modification of the present embodiment. It can be established as That is, the present embodiment is not particularly limited as the configuration of the secondary-side resonance circuit and the rectifier circuit. Further, such a secondary side voltage doubler rectifier circuit and a quadruple voltage doubler rectifier circuit, as shown in FIG.
The present invention can also be applied to a circuit having a self-excited voltage resonance type converter on the primary side.

In each of the above-described embodiments, the so-called single-ended configuration having one switching element is employed as the primary-side voltage resonance converter. The present invention can also be applied to a so-called push-pull method in which switching is alternately performed.

[0097]

As described above, according to the present invention, in a power supply circuit as a complex resonance type converter provided with a power factor improving circuit, the power factor becomes substantially constant with respect to the fluctuation of the AC input voltage, and the load is reduced. The characteristic that the power factor rises with the decrease in power is obtained, and therefore, there is an effect that a power supply circuit that can be practically used sufficiently for office equipment and personal computers in which the load conditions fluctuate can be provided. . The soft start means eliminates the need to insert an inrush current limiting resistor in the AC input line, thereby eliminating the need for a large high-temperature heat-generating component, and improving power conversion efficiency by eliminating power loss due to the inrush current limiting resistor. Note that the soft start means using an NPN-type or PNP-type transistor having a low withstand voltage and a low saturation voltage has a low loss and does not require a heat sink. Therefore, when the electromagnetic power relay is provided for short-circuiting the inrush current limiting resistor, the collector loss becomes almost equal to the driving power of 0.5 W,
The power loss due to the soft start means of the present invention is of a level that does not pose a problem in practical use. Further, since there is no need to provide an electromagnetic power relay associated with the inrush current limiting resistor, it is not necessary to take measures against secondary defects such as when the electromagnetic power relay is open and fails. Also, compared with the provision of a rush current limiting resistor and the associated resistance short circuit system using electromagnetic power relays and semiconductor switches, the circuit configuration is very simple, and it is significantly advantageous in terms of the number of components and cost. Becomes

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing a configuration of a switching power supply circuit according to a first embodiment of the present invention.

FIG. 2 is a side sectional view showing a structure of an insulating converter transformer employed in the power supply circuit of the present embodiment.

FIG. 3 is an explanatory diagram showing each operation when the mutual inductance is + M / -M.

FIG. 4 is a waveform chart showing an operation of the switching power supply circuit of the present embodiment.

FIG. 5 is an explanatory diagram of an input AC current when power is turned on for the switching power supply circuit of the present embodiment.

FIG. 6 is a circuit diagram illustrating a configuration of a switching power supply circuit according to a second embodiment.

FIG. 7 is a circuit diagram illustrating a configuration of a switching power supply circuit according to a third embodiment.

FIG. 8 is a characteristic diagram illustrating a relationship between a load power and a power factor and a relationship between an AC input voltage and a power factor of the switching power supply circuit according to the third embodiment.

FIG. 9 is a circuit diagram illustrating a configuration of a switching power supply circuit according to a fourth embodiment.

FIG. 10 is a circuit diagram showing a configuration of a power supply circuit as a prior art.

FIG. 11 is a circuit diagram showing a configuration of a power supply circuit as a prior art.

FIG. 12 is an explanatory diagram of an input alternating current at the time of power-on of a power supply circuit of the related art.

FIG. 13 is a characteristic diagram illustrating a relationship between an AC input voltage and a power factor of a power supply circuit according to the related art.

FIG. 14 is a waveform diagram showing an operation for input of a commercial AC power supply according to load power of a power supply circuit of the prior art.

[Explanation of symbols]

1 control circuit, 10, 11 power factor improvement circuit, 13, 13
A Soft start circuit, Di bridge rectifier circuit, C
i Smoothing capacitor, LS choke coil, D1 high speed recovery type diode, C3 resonance capacitor, C2 secondary parallel resonance capacitor, Cs1 secondary series resonance capacitor, PRT orthogonal control transformer, PIT isolation converter transformer, Q1, Q21 switching element

Claims (3)

[Claims] A rectifying diode for inputting a commercial AC power and rectifying the rectifying diode and a rectifying and smoothing means for generating a rectified and smoothed voltage by a smoothing capacitor; and a gap formed so as to obtain a required coupling coefficient which is loosely coupled. An insulating converter transformer provided for transmitting a primary side output to a secondary side, and switching means adapted to intermittently output the smoothed voltage by a switching element to output to a primary winding of the insulating converter transformer, A primary-side parallel resonance circuit formed by a leakage inductance component including a primary winding of the insulating converter transformer and a capacitance of a parallel resonance capacitor, and making the operation of the switching means a voltage resonance type; The switching output is fed back to the rectified current path by magnetic coupling or electrostatic coupling. Power factor improving means adapted to improve the power factor by intermittently commutating the rectified current based on the returned switching output; and a power factor improving means arranged in front of the power factor improving means. A soft switch means formed by using a transistor, a leakage inductance component of a secondary winding of the insulating converter transformer, and a capacitance of a secondary resonance capacitor so that an inrush current can be limited. The secondary side resonance circuit formed in and the secondary side resonance circuit which is formed including the secondary side resonance circuit, inputs the alternating voltage obtained in the secondary winding of the insulating converter transformer, performs rectification operation, and performs the rectification operation. DC output voltage generating means configured to generate a DC output voltage; and the switching means according to the level of the secondary DC output voltage. The switching frequency is variably controlling,
And a constant voltage control means configured to perform constant voltage control on the secondary side DC output voltage. 2. The soft switch means includes: an NPN transistor connected to a positive electrode of the rectifier diode; a resistor arranged between a collector and a base of the NPN transistor; The switching power supply circuit according to claim 1, wherein the switching power supply circuit is formed having: 3. The soft switch means includes: a PNP transistor connected to a negative electrode of the rectifier diode; a resistor arranged between a collector and a base of the PNP transistor; The switching power supply circuit according to claim 1, wherein the switching power supply circuit is formed having: