JP2020089088A - Power source device for electric motor vehicle - Google Patents

Abstract

A power supply device for an electric vehicle capable of minimizing the switching frequency of a resonance inverter and suppressing noise generated by a transformer. SOLUTION: An electric vehicle power supply device according to an embodiment includes a high-frequency transformer, a step-up chopper, an inverter, and a control circuit. The boost chopper boosts a DC voltage supplied from a DC power supply. The inverter uses the DC voltage supplied from the boost chopper to supply AC current to the high frequency transformer. A control circuit synchronizes the boost chopper and the inverter and causes the boost chopper to switch at a switching frequency that is an even multiple of the switching frequency of the inverter. [Selection diagram] Fig. 1

Description

Embodiments of the present invention relate to an electric vehicle power supply device.

An electric vehicle (moving body) is a power supply device that converts a DC voltage supplied from a high-voltage train line (for example, an overhead train line or a third rail) into a voltage according to a load and outputs the DC voltage to the load. Equipped with. For example, an electric vehicle includes, as an electric vehicle power source device, a power source device for driving a traveling electric motor and an auxiliary power source device for supplying electric power to other devices such as lighting and air conditioning.

The auxiliary power supply device is a high-frequency transformer that is excited by a high-frequency AC current, a step-up chopper that adjusts the DC voltage from the train line, and the output of the step-up chopper is converted to high-frequency AC and supplied to the high-frequency transformer. And a resonant inverter that operates. According to such a configuration, there is a problem that noise may occur due to the switching timing of the boost chopper and the resonance inverter.

JP-A-3-222866

The problem to be solved by the present invention is to provide an electric vehicle power supply device capable of suppressing the switching frequency of the resonant inverter to a minimum and suppressing the noise generated by the transformer.

The electric vehicle power supply device according to the embodiment includes a high-frequency transformer, a step-up chopper, an inverter, and a control circuit. The boost chopper boosts the DC voltage supplied from the DC power supply. The inverter supplies an alternating current to the high frequency transformer by using the direct current voltage supplied from the boost chopper. The control circuit synchronizes the boost chopper and the inverter, and switches the boost chopper at a switching frequency that is an even multiple of the switching frequency of the inverter.

FIG. 1 is a diagram for explaining an example of the configuration of the electric vehicle power supply device according to the first embodiment. FIG. 2 is an explanatory diagram for describing an example of the operation of the booster circuit of the electric vehicle power supply device according to the first embodiment. FIG. 3 is an explanatory diagram for explaining an example of the operation of the booster circuit of the electric vehicle power supply device according to the first embodiment. FIG. 4 is an explanatory diagram for describing an example of the configuration of the control circuit of the electric vehicle power supply device according to the first embodiment. FIG. 5 is an explanatory diagram for describing an example of the operation of the control circuit of the electric vehicle power supply device according to the first embodiment. FIG. 6 is an explanatory diagram for describing an example of the operation of the control circuit of the electric vehicle power supply device according to the first embodiment. FIG. 7 is a diagram for explaining an example of the configuration of the electric vehicle power supply device according to the second embodiment. FIG. 8 is an explanatory diagram for describing an example of the configuration of the control circuit of the electric vehicle power supply device according to the second embodiment. FIG. 9 is an explanatory diagram for describing an example of the operation of the control circuit of the electric vehicle power supply device according to the second embodiment.

Hereinafter, embodiments will be described with reference to the drawings.
(First embodiment)
FIG. 1 is an explanatory diagram showing a configuration example of an electric vehicle power supply device 1 according to the first embodiment. The electric vehicle power supply device 1 is mounted on a moving body such as an electric vehicle. The electric vehicle power supply device 1 receives DC power from an overhead train line or a train line 2 such as a third railroad track via a current collector 3, converts the received DC power according to the rating of the load 4, and converts it into a load 4. Supply. In the present embodiment, the electric vehicle power supply device 1 will be described as an auxiliary power supply device that supplies electric power to a load such as lighting and air conditioning of the electric vehicle. The electric vehicle includes a main power supply device (not shown) for driving the traveling electric motor. The main power supply device causes the electric vehicle to travel on the track 5 by driving the traveling electric motor with the DC power received from the train line 2 via the current collector 3.

The electric vehicle power supply device 1 as an auxiliary power supply device is connected to a device that operates at a lower voltage than a traveling electric motor. Therefore, in the electric vehicle power source device 1, the primary side to which electric power is input and the secondary side to output electric power need to be insulated.

In order to ensure insulation between the primary side and the secondary side, there is a transformer that uses a transformer provided with a pair of windings (coils) electromagnetically coupled to insulate the primary side and the secondary side. The transformer becomes larger as the excitation frequency becomes lower. For example, a transformer in which an excitation frequency corresponding to the frequency of a commercial power source is set becomes large. Therefore, the electric vehicle power supply device 1 of the present embodiment uses the high-frequency transformer to insulate the primary side and the secondary side from each other and realize miniaturization.

First, the configuration of the electric vehicle power supply device 1 will be described.
The electric vehicle power supply device 1 includes a booster circuit 11, a power conversion circuit 12, and a three-phase inverter 13. The electric vehicle power supply device 1 also includes a control circuit 14 that controls the booster circuit 11, the power conversion circuit 12, and the three-phase inverter 13.

The booster circuit 11 boosts the DC power input from the train line 2 via the current collector 3. The booster circuit 11 includes a booster reactor L, a first booster chopper 21 and a second booster chopper 22.

The first boost chopper 21 includes a first switch S1 and a first diode D1. The first boost chopper 21 controls the current flowing through the boost reactor L by turning on and off the first switch S1 based on the control of the control circuit 14. As a result, the first boost chopper 21 outputs a DC voltage boosted by the electromagnetic energy stored in the boost reactor L. Further, the first boost chopper 21 may include a filter capacitor that stabilizes the output DC voltage.

The second boost chopper 22 includes a second switch S2 and a second diode D2. The second step-up chopper 22 controls the current flowing through the step-up reactor L by turning on and off the second switch S2 based on the control of the control circuit 14. As a result, the second boost chopper 22 outputs the DC voltage boosted by the electromagnetic energy stored in the boost reactor L. In addition, the second boost chopper 22 may include a filter capacitor that stabilizes the output DC voltage.

The power conversion circuit 12 converts the DC power output from the booster circuit 11 into DC load power. The power conversion circuit 12 includes, for example, a first resonant inverter 31, a first high-frequency transformer 32, a first rectifier 33, a first capacitor C1, a second resonant inverter 34, a second high-frequency transformer 35, It has a second rectifier 36 and a second capacitor C2.

The first resonant inverter 31 is a circuit that causes an alternating current (inverter current, single-phase alternating current, or the like) to flow in the first high-frequency transformer 32 using the direct-current voltage supplied from the first boost chopper 21. .. The first resonant inverter 31 is configured as, for example, a resonant single-phase half bridge inverter. The first resonant inverter 31 includes a third switch S3, a fourth switch S4, a third capacitor C3, and a fourth capacitor C4. The first high-frequency transformer 32 is connected to the connection point between the third switch S3 and the fourth switch S4 and the connection point between the third capacitor C3 and the fourth capacitor C4. The first resonance inverter 31 supplies an alternating current to the first high-frequency transformer 32 by performing on/off control of the third switch S3 and the fourth switch S4 under the control of the control circuit 14. The third switch S3 side of the first resonant inverter 31 is referred to as the upper arm of the resonant inverter, and the fourth switch S4 side of the first resonant inverter 31 is referred to as the lower arm of the resonant inverter.

The first high-frequency transformer 32 is a secondary winding (primary winding) that generates a magnetic flux, and a secondary winding that is insulated from the primary winding and is excited by the magnetic flux generated in the primary winding. It is an insulation transformer having a side winding (secondary winding). When an alternating current is supplied from the first resonance inverter 31 to the primary winding of the first high frequency transformer 32, magnetic flux is generated in the primary winding. The magnetic flux generated in the primary winding causes an induced current in the secondary winding. As a result, the first high frequency transformer 32 supplies power to the secondary side according to the alternating current input from the primary side.

The first rectifier 33 is a circuit that rectifies the electric power generated in the secondary winding of the first high frequency transformer 32. The first rectifier 33 is configured, for example, as a rectifying bridge in which a plurality of diodes are combined.

The first capacitor C1 smoothes the positive voltage supplied from the first rectifier 33. The first capacitor C1 supplies a DC voltage to the three-phase inverters 13 connected in parallel.

The second resonance inverter 34 is a circuit that causes an alternating current (inverter current, single-phase alternating current, or the like) to flow in the second high-frequency transformer 35 using the direct-current voltage supplied from the second boost chopper 22. . The second resonant inverter 34 is configured as, for example, a resonant single-phase half bridge inverter. The second resonant inverter 34 includes a fifth switch S5, a sixth switch S6, a fifth capacitor C5, and a sixth capacitor C6. The second high frequency transformer 35 is connected to a connection point between the fifth switch S5 and the sixth switch S6 and a connection point between the fifth capacitor C5 and the sixth capacitor C6. The second resonance inverter 34 supplies an alternating current to the second high-frequency transformer 35 by controlling on/off of the fifth switch S5 and the sixth switch S6 based on the control of the control circuit 14. The fifth switch S5 side of the second resonant inverter 34 is referred to as the upper arm of the resonant inverter, and the sixth switch S6 side of the second resonant inverter 34 is referred to as the lower arm of the resonant inverter.

The second high frequency transformer 35 is a secondary winding (primary winding) that generates a magnetic flux, and a secondary winding that is insulated from the primary winding and is excited by the magnetic flux generated in the primary winding. It is an insulation transformer having a side winding (secondary winding). When an alternating current is supplied from the second resonance inverter 34 to the primary winding of the second high frequency transformer 35, magnetic flux is generated in the primary winding. The magnetic flux generated in the primary winding causes an induced current in the secondary winding. As a result, the second high-frequency transformer 35 supplies power to the secondary side according to the alternating current input from the primary side.

The second rectifier 36 is a circuit that rectifies the electric power generated in the secondary winding of the second high frequency transformer 35. The second rectifier 36 is configured, for example, as a rectifying bridge in which a plurality of diodes are combined.

The second capacitor C2 smoothes the positive voltage supplied from the second rectifier 36. The second capacitor C2 supplies a DC voltage to the three-phase inverter 13 connected in parallel. With this configuration, the DC voltage from the first capacitor C1 and the second capacitor C2 is supplied to the three-phase inverter 13.

The three-phase inverter 13 converts DC power into AC power (three-phase AC power) and outputs the AC power to the load 4. The three-phase inverter 13 includes a plurality of switches whose on/off is controlled by the control circuit 14. The three-phase inverter 13 outputs three-phase AC power whose phases are different from each other by 120 degrees from each other by turning on and off a plurality of switches by the control circuit 14.

The control circuit 14 controls the booster circuit 11, the power conversion circuit 12, and the three-phase inverter 13. The control circuit 14 is configured as a logic circuit that generates a pulse signal, for example. Further, the control circuit 14 includes a processor that is an arithmetic element that executes arithmetic processing, and a memory that stores a program and data used in the program, and the processor executes the program to generate a pulse signal. May be

The control circuit 14 controls the operations of the booster circuit 11, the power conversion circuit 12, and the three-phase inverter 13 by inputting the pulse signal to the booster circuit 11, the power conversion circuit 12, and the three-phase inverter 13, respectively. For example, the control circuit 14 performs PWM control for adjusting the ON/OFF duty ratio of the pulse signal based on the detection result of a current detector or a voltage detector (not shown). As a result, the control circuit 14 adjusts the output of the booster circuit 11, the output of the power conversion circuit 12, and the output of the three-phase inverter 13, respectively.

As described above, the control circuit 14 supplies the pulse signal to the first resonant inverter 31 and the second resonant inverter 34. As a result, the control circuit 14 converts the DC power supplied from the electric power line 2 into AC power, and outputs inverter currents from the first resonant inverter 31 and the second resonant inverter 34, respectively.

In addition, the DC voltage supplied from the train line 2 may not be stable. Therefore, the control circuit 14 supplies a pulse signal to the first boost chopper 21 and the second boost chopper 22 of the boost circuit 11. Thereby, the control circuit 14 controls so that the stable DC voltage is supplied to the first resonance inverter 31 and the second resonance inverter 34.

The first resonant inverter 31 and the second resonant inverter 34 are supplied from the first resonant inverter 31 and the second resonant inverter 34 to the first high frequency transformer 32 and the second high frequency transformer 35, respectively. It operates at the same switching frequency as the frequency of the inverter current (fundamental frequency). The fundamental frequency is determined by the excitation frequencies of the first high frequency transformer 32 and the second high frequency transformer 35. Further, the pulse width (on/off duty ratio) of the pulse signal supplied from the control circuit 14 to the first resonant inverter 31 and the second resonant inverter 34 is always constant.

Next, the relationship between the operation of the boost chopper and the inverter current will be described.
2 and 3 are explanatory diagrams for explaining the relationship between the on/off timing of the boost chopper of the booster circuit 11, the output voltage of the boost chopper, and the inverter current supplied from the resonant inverter to the high frequency transformer. The first boost chopper 21 and the second boost chopper 22, and the first resonant inverter 31 and the second resonant inverter 34 have the same configuration. Therefore, the operation timing of the first step-up chopper 21 (chopper on/off timing), the output voltage of the first step-up chopper 21 (chopper output voltage), and the first resonance inverter 31 supply the first high-frequency transformer 32. An example of the generated alternating current (inverter current) will be described.

The graph 41 of FIG. 2 shows an example in which the first boost chopper 21 is synchronized with the first resonant inverter 31 and the first boost chopper 21 is operated at the same switching frequency as that of the first resonant inverter 31. Show. In this case, the first boost chopper 21 is turned on when the first resonant inverter 31 is generating a positive current. The chopper output voltage of the first boost chopper 21 becomes higher than that when the first resonant inverter 31 is generating a negative current. As described above, the first resonance inverter 31 is always controlled by the pulse signal having the constant pulse width. Therefore, the first resonant inverter 31 cannot adjust the output voltage. Therefore, when the chopper output voltage supplied from the first boost chopper 21 decreases, the inverter current supplied from the first resonant inverter 31 to the first high frequency transformer 32 also decreases. As a result, the positive and negative amplitudes of the inverter current are unbalanced.

In this way, when the positive side amplitude and the negative side amplitude of the inverter current are unbalanced, the excitation frequency component of the resonant inverter is generated as magnetostrictive noise. Further, when the positive-side amplitude and the negative-side amplitude of the inverter current do not match, the inverter current contains a DC component. As a result, magnetic bias occurs.

The graph 42 of FIG. 3 shows that the first boost chopper 21 is synchronized with the first resonant inverter 31 and the first boost chopper 21 is switched at an even multiple (twice in this example) of the first resonant inverter 31. An example of operating at a frequency is shown.

In this case, the pulsation of the output voltage of the first boost chopper 21 is completed every half cycle of the first resonant inverter 31. Therefore, the chopper output voltage of the first step-up chopper 21 is when the first resonant inverter 31 is generating a positive side current and when the first resonant inverter 31 is generating a negative side current. It is the same as time. Therefore, the positive-side amplitude and the negative-side amplitude of the inverter current supplied from the first resonance inverter 31 to the first high-frequency transformer 32 match.

In this way, the control circuit 14 controls the first boost chopper 21 and the first resonance so that the positive-side amplitude and the negative-side amplitude of the inverter current supplied to the first high-frequency transformer 32 match. By controlling the inverter 31, magnetostrictive noise can be reduced. Further, it is possible to prevent the occurrence of magnetic bias.

When the positive-side amplitude and the negative-side amplitude of the inverter current supplied from the first resonance inverter 31 to the first high-frequency transformer 32 match, the switching frequency of the first resonance inverter 31 is twice the frequency. Noise. By reducing the frequency of this noise to a high frequency outside the audible range of human beings, noise reduction can be realized. That is, the switching frequency of the resonant inverter is set to be equal to or higher than the threshold value set based on the upper limit of the human audible range, and the switching frequency of the boost chopper is set to an even multiple of the switching frequency of the resonant inverter. The generated noise can be reduced.

The upper limit of the human audible range is generally between 15 kHz and 16 kHz, although there are individual differences due to various factors. Therefore, the switching frequency of the resonant inverter is set to 8 kHz or higher, which is half the upper limit of the human audible range, and the switching frequency of the boost chopper is set to 16 kHz or higher, which is an even multiple of the switching frequency (fundamental frequency) of the resonant inverter. By doing so, noise generated by the transformer can be reduced. That is, by setting the above threshold to 8 kHz or more, the switching frequency of the resonant inverter can be minimized and the noise generated by the transformer can be suppressed.

Next, a configuration for the control circuit 14 to operate the boost chopper and the resonant inverter will be described with reference to FIGS. 4 to 6.

FIG. 4 is an explanatory diagram for explaining a specific configuration example of the control circuit 14. FIG. 5 shows that the first boost chopper 21 and the second boost chopper 22 are synchronized with the first resonant inverter 31 and the second resonant inverter 34, respectively, and the first boost chopper 21 and the second boost chopper 22 are also provided. FIG. 5 is an explanatory diagram for explaining an example in which the first resonance inverter 31 and the second resonance inverter 34 are operated at the same switching frequency. In FIG. 6, the first boost chopper 21 and the second boost chopper 22 are synchronized with the first resonant inverter 31 and the second resonant inverter 34, and the first boost chopper 21 and the second boost chopper 22 are connected. It is explanatory drawing for demonstrating the structure for operating with the switching frequency of the even-numbered multiple (two times in this example) of the switching frequency of the 1st resonance inverter 31 and the 2nd resonance inverter 34.

In the example of FIG. 4, an example in which the control circuit 14 is configured as an analog circuit will be described. However, the control circuit 14 includes a memory in which a program that performs processing equivalent to that of the analog circuit in FIG. It may be configured by a processor that executes a program.

The control circuit 14 includes a first comparator 51, a second comparator 52, a first on-time delay (ONTD) circuit 53, a second ONTD circuit 54, and a NOT circuit 55. Further, the first carrier wave 61, the second carrier wave 62, the voltage command 63, and the fixed value 64 shown in FIGS. 5 and 6 are input to the control circuit 14.

The first carrier wave 61 and the second carrier wave 62 are triangular waves supplied from a driver's cab of an electric vehicle, a control device that controls traveling of the electric vehicle, or a circuit that outputs another carrier wave. The control circuit 14 may be configured to generate the first carrier wave 61 and the second carrier wave 62 by itself. Further, in the example of FIG. 5, the first carrier wave 61 and the second carrier wave 62 have the same frequency.

The voltage command 63 is a control signal (voltage value) supplied from a driver's cab of an electric vehicle or a control device that controls traveling of the electric vehicle.

The fixed value 64 is a signal (voltage value) triangular wave supplied from a driver's cab of an electric vehicle, a control device that controls running of the electric vehicle, or another circuit that outputs a fixed value.

The first carrier wave 61 and the voltage command 63 are input to the first comparator 51 of the control circuit 14. The first comparator 51 uses the result of comparison between the first carrier wave 61 and the voltage command 63 as a signal for turning on/off the boost chopper (boost chopper on/off command), and the first boost chopper 21 and the second booster. Output to the chopper 22. Specifically, the first comparator 51 turns on the boost chopper at the timing when the voltage command 63 is the first carrier wave 61 or more, and boosts at the timing when the voltage command 63 is less than the first carrier wave 61. Outputs a boost chopper on/off command to turn off the chopper.

The second carrier wave 62 and the fixed value 64 are input to the second comparator 52 of the control circuit 14. The second comparator 52 outputs the comparison result of the second carrier wave 62 and the fixed value 64. The fixed value 64 is set to be half the maximum value of the second carrier wave 62. As a result, the on/off duty ratio of the output signal of the second comparator 52 is controlled to 50%.

The output signal of the second comparator 52 is input to the first ONTD circuit 53 of the control circuit 14. The first ONTD circuit 53 delays the timing of turning on the output signal of the second comparator 52 and outputs it. The signal output from the first ONTD circuit 53 is a signal (resonant inverter upper arm on/off command) for turning on/off the switches of the upper arms of the first resonant inverter 31 and the second resonant inverter 34. The first ONTD circuit 53 supplies a resonance inverter upper arm ON/OFF command to the first resonance inverter 31 and the second resonance inverter 34.

A signal obtained by inverting the output signal of the second comparator 52 by the NOT circuit 55 is input to the second ONTD circuit 54 of the control circuit 14. The second ONTD circuit 54 delays the timing at which the input signal is turned on and then outputs the delayed signal. The signal output from the second ONTD circuit 54 is a signal (resonant inverter lower arm on/off command) for turning on/off the switches of the lower arms of the first resonant inverter 31 and the second resonant inverter 34. The second ONTD circuit 54 supplies a resonance inverter lower arm ON/OFF command to the first resonance inverter 31 and the second resonance inverter 34.

With the above configuration, the control circuit 14 can synchronize the boost chopper with the resonant inverter and operate the boost chopper at the same switching frequency as the resonant inverter. In order to operate the boost chopper at a switching frequency that is an even multiple of the switching frequency of the resonant inverter, as shown in FIG. 6, the frequency of the first carrier wave 61 is an even multiple of the frequency of the second carrier wave 62. I wish I had it.

As described above, the electric vehicle power source device 1 according to the first embodiment uses the high-frequency transformer, the step-up chopper for stepping up the DC voltage supplied from the DC power source, and the DC voltage supplied from the step-up chopper. A resonance inverter for supplying an alternating current (inverter current) to the high frequency transformer, and a control circuit 14. The control circuit 14 synchronizes the boost chopper and the resonant inverter, and switches the boost chopper at a switching frequency that is an even multiple of the switching frequency of the resonant inverter. As a result, the electric vehicle power source device 1 can reduce the magnetostrictive noise and prevent the occurrence of magnetic bias.

Further, the control circuit 14 sets the switching frequency of the resonant inverter to a value equal to or higher than a preset threshold value. Thereby, the frequency of noise generated in the high frequency transformer can be controlled.

The control circuit 14 also sets the threshold value based on the upper limit of the human audible range. As a result, the noise generated in the high frequency transformer can be controlled so as to be out of the human audible range.

Further, the control circuit 14 sets the threshold value to 8 kHz or higher, which is half the upper limit of the human audible range, and sets the switching frequency of the boost chopper to 16 kHz or higher, which is an even multiple of the switching frequency of the resonant inverter. As a result, the noise generated by the high frequency transformer can be controlled so as to be out of the human audible range.

(Second embodiment)
Next, the electric vehicle power supply device 1A according to the second embodiment will be described.
The electric vehicle power source device 1A is different from the electric vehicle power source device 1 according to the first embodiment in the configuration of the booster circuit and the control circuit. The same components as those in the first embodiment are designated by the same reference numerals, and detailed description thereof will be omitted.

FIG. 7 is an explanatory diagram for describing an example of the electric vehicle power source device 1A according to the second embodiment. The electric vehicle power supply device 1A includes a booster circuit 11A, a power conversion circuit 12, a three-phase inverter 13, and a control circuit 14A.

The booster circuit 11A includes two booster reactors L, a first booster chopper 21, and a second booster chopper 22. The two boosting reactors L are connected in parallel to the collector 3. The first boost chopper 21 and the second boost chopper 22 are connected to the rear stage of each boost reactor L. Further, the first resonant inverter 31 and the second resonant inverter 34 are connected in parallel to the first boost chopper 21, and the first resonant inverter 31 and the second resonant inverter 31 are connected to the second boost chopper 22. The resonant inverter 34 is connected in parallel. That is, in the booster circuit 11A, the first boost chopper 21 and the second boost chopper 22 are connected in parallel between the current collector 3 and the first resonant inverter 31 and the second resonant inverter 34. With configuration.

Next, the configuration for the control circuit 14A to operate the boost chopper and the resonant inverter will be described with reference to FIGS. 8 and 9.

FIG. 8 is an explanatory diagram for explaining a specific configuration example of the control circuit 14A. FIG. 9 is an explanatory diagram for explaining an example of operating the pressure chopper and the resonance inverter.

In the example of FIG. 8, an example in which the control circuit 14A is configured as an analog circuit will be described. However, the control circuit 14A includes a memory in which a program that performs processing equivalent to that of the analog circuit in FIG. It may be configured by a processor that executes a program.

The control circuit 14A includes a second comparator 52, a first ONTD circuit 53, a second ONTD circuit 54, a NOT circuit 55, a third comparator 56A, and a fourth comparator 57A. Further, the third carrier wave 65A, the fourth carrier wave 66A, the voltage command 63, and the fixed value 64 shown in FIG. 9 are input to the control circuit 14A. The second carrier wave 62 shown in FIGS. 5 and 6 is input to the second comparator 52 of the control circuit 14A. The second comparator 52 may be configured to receive the third carrier wave 65A or the fourth carrier wave 66A.

The third carrier wave 65A and the fourth carrier wave 66A are triangular waves supplied from a driver's cab of an electric vehicle, a control device for controlling traveling of the electric vehicle, or a circuit for outputting another carrier wave. The control circuit 14A may be configured to generate the third carrier wave 65A and the fourth carrier wave 66A by itself. For example, the third carrier wave 65A and the fourth carrier wave 66A have the same frequency as the second carrier wave 62. In the example of FIG. 9, the third carrier wave 65A and the fourth carrier wave 66A are configured as triangular waves having the same frequency and different phases. Further, in the example of FIG. 9, the third carrier wave 65A and the fourth carrier wave 66A are configured as triangular waves whose phases are different by 180 degrees.

The third carrier 56A and the voltage command 63 are input to the third comparator 56A of the control circuit 14A. The third comparator 56A sends the result of comparison between the third carrier wave 65A and the voltage command 63 to the first boost chopper 21 as a signal for turning on and off the first boost chopper 21 (boost chopper on/off command). Output. Specifically, the third comparator 56A turns on the first boost chopper 21 at the timing when the voltage command 63 is equal to or higher than the third carrier wave 65A, and the voltage command 63 is less than the third carrier wave 65A. At a timing, a boost chopper on/off command for turning off the first boost chopper 21 is output to the first boost chopper 21.

The fourth carrier wave 66A and the voltage command 63 are input to the fourth comparator 57A of the control circuit 14A. The fourth comparator 57A sends the result of comparison between the fourth carrier 66A and the voltage command 63 to the second boost chopper 22 as a signal (step-up chopper on/off command) for turning on/off the second boost chopper 22. Output. Specifically, the fourth comparator 57A turns on the second boost chopper 22 at the timing when the voltage command 63 is equal to or higher than the fourth carrier wave 66A, and the voltage command 63 is less than the fourth carrier wave 66A. At a timing, a boost chopper on/off command for turning off the second boost chopper 22 is output to the second boost chopper 22.

According to the above configuration, the first boost chopper 21 and the second boost chopper 22 are alternately turned on. As a result, the booster circuit 11A can obtain a switching frequency that is twice the frequency of the third carrier wave 65A and the fourth carrier wave 66A. That is, even when the first boost chopper 21 and the second boost chopper 22 are controlled based on the carrier wave having the same frequency as the output currents of the first resonant inverter 31 and the second resonant inverter 34, The pulsation of the output voltage of the first boost chopper 21 and the second boost chopper 22 is twice the frequency of the output current of the first resonant inverter 31 and the second resonant inverter 34.

As described above, the electric vehicle power source device 1A of the second embodiment includes a pair of high frequency transformers, a DC voltage boosted from a DC power source, and a DC power source and a pair of high frequency transformers. A pair of step-up choppers connected in parallel between the two and a pair of resonant inverters that supply an alternating current to each high-frequency transformer by using a DC voltage supplied from the pair of step-up choppers. The control circuit 14A of the electric vehicle power supply device 1A synchronizes each combination of the step-up chopper and the resonant inverter to switch the pair of step-up choppers at different phases, and switch the step-up chopper to an integral multiple of the switching frequency of the resonant inverter. Switching at frequency. As a result, the electric vehicle power source device 1A can reduce the magnetostrictive noise and prevent the occurrence of magnetic bias.

Further, the control circuit 14A switches the two boosting choppers by shifting their phases by 180 degrees. As a result, the pulsation of the output voltage of the booster circuit 11A is completed every half (half cycle) of the switching cycle of the first resonant inverter 31 and the second resonant inverter 34. Therefore, the positive side amplitude and the negative side amplitude of the output currents of the first resonance inverter 31 and the second resonance inverter 34 match. As a result, the electric vehicle power source device 1A can reduce the magnetostrictive noise and prevent the occurrence of magnetic bias.

Further, the control circuit 14A sets the switching frequency of the resonant inverter to a value equal to or higher than a preset threshold value. Thereby, the frequency of noise generated in the high frequency transformer can be controlled.

Further, the control circuit 14A sets the threshold value based on the upper limit of the human audible range. As a result, the noise generated in the high frequency transformer can be controlled so as to be out of the human audible range.

Further, the control circuit 14A sets the threshold to 8 kHz or higher, which is half the upper limit of the human audible range, and sets the switching frequency of the boost chopper to 8 kHz or higher, which is an integral multiple of the switching frequency of the resonant inverter. As a result, the noise generated by the high frequency transformer can be controlled so as to be out of the human audible range.

Although some embodiments of the present invention have been described, these embodiments are presented as examples and are not intended to limit the scope of the invention. These novel embodiments can be implemented in various other forms, and various omissions, replacements, and changes can be made without departing from the spirit of the invention. These embodiments and their modifications are included in the scope and gist of the invention, and are also included in the invention described in the claims and the scope equivalent thereto.

DESCRIPTION OF SYMBOLS 1... Electric vehicle power supply device, 1A... Electric vehicle power supply device, 2... Train line, 3... Current collector, 4... Load, 5... Line, 11... Boost circuit, 11A... Boost circuit, 12... Power conversion circuit, 13... Three-phase inverter, 14... Control circuit, 14A... Control circuit, 21... 1st step-up chopper, 22... 2nd step-up chopper, 31... 1st resonance inverter, 32... 1st high frequency transformer, 33 ... 1st rectifier, 34... 2nd resonance inverter, 35... 2nd high frequency transformer, 36... 2nd rectifier, 51... 1st comparator, 52... 2nd comparator, 53... 1st ONTD circuit, 54... Second ONTD circuit, 55... NOT circuit, 56A... Third comparator, 57A... Fourth comparator, 61... First carrier wave, 62... Second carrier wave, 63... Voltage Command, 64... Fixed value, 65A... Third carrier wave, 66A... Fourth carrier wave, C1... First capacitor, C2... Second capacitor, C3... Third capacitor, C4... Fourth capacitor, C5 ...Fifth capacitor, C6...sixth capacitor, D1...first diode, D2...second diode, S1...first switch, S2...second switch, S3...third switch, S4... Fourth switch, S5... Fifth switch, S6... Sixth switch.

Claims (8)

A high frequency transformer,
A boost chopper that boosts the DC voltage supplied from the DC power supply,
An inverter that supplies an alternating current to the high-frequency transformer using the direct-current voltage supplied from the boost chopper,
A control circuit that synchronizes the boost chopper and the inverter, and switches the boost chopper at a switching frequency that is an even multiple of the switching frequency of the inverter;
A power supply device for an electric vehicle, comprising: The electric vehicle power supply device according to claim 1, wherein the control circuit switches the boost chopper at a frequency equal to or higher than a preset threshold value. The electric vehicle power supply device according to claim 2, wherein the threshold value is set based on a human audible range. The electric vehicle power supply device according to claim 3, wherein the threshold value is a value of 16 kHz or more. A pair of high frequency transformers,
A pair of step-up choppers for boosting a DC voltage supplied from a DC power source and connected in parallel between the DC power source and the pair of high-frequency transformers;
A pair of inverters for supplying an alternating current to each of the high frequency transformers by using a direct current voltage supplied from a pair of the step-up choppers;
A control circuit that synchronizes each combination of the boost chopper and the inverter, switches a pair of the boost choppers in different phases, and switches the boost chopper at a switching frequency that is an integer multiple of the switching frequency of the inverter,
A power supply device for an electric vehicle, comprising: The electric vehicle power supply device according to claim 5, wherein the control circuit causes the pair of boost choppers to switch at phases different by 180 degrees. The electric vehicle power supply device according to claim 6, wherein the control circuit causes the boost chopper to switch at a frequency equal to or higher than a threshold value set in advance based on a human audible range. The electric vehicle power supply device according to claim 7, wherein the threshold value is a value of 8 kHz or more.