JP3394812B2 - Power supply - Google Patents

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a power supply device for inputting a pulsating current power supply obtained by rectifying an AC power supply and outputting high frequency power.

[0002]

2. Description of the Related Art Conventionally, as shown in FIG. 42, a power supply device has been proposed in which an inverter device 1 is driven by an AC power supply AC such as a commercial power supply to supply high frequency power to a load such as a discharge lamp DL. Has been done. Since the input to the inverter device 1 is a DC power supply, in the circuit of FIG. 42, the AC power supply AC is full-wave rectified through the rectifier circuit DB including a diode bridge, and the pulsating current power supply that is the output of the rectifier circuit DB is smoothed. In order to do so, a boost type chopper circuit (DC
-DC converter) 4 is used as an active smoothing filter to obtain a stabilized DC power supply.

The chopper circuit 4 connects a series circuit of a primary winding of a choke coil CH, a switching element Q ₃ composed of an FET and a resistor R ₁ between output terminals of a rectifier circuit DB,
It has a basic configuration in which a series circuit of a diode D ₁₀ and a capacitor C ₁ is connected in parallel to a series circuit of a switching element Q ₁ and a resistor R ₁ . The control circuit 5 controls ON / OFF of the switching element Q ₃ . Control circuit 5
Then, the voltage between the output terminals of the rectifier circuit DB is set to the resistances R ₂ , R ₃
Connected in series with the switching element Q ₁ and the voltage induced by the secondary winding of the choke coil CH, the voltage across the capacitor C ₁ divided by the resistors R ₄ and R ₅ , The ON period and the OFF period of the switching element Q ₁ are determined based on the relationship with the voltage across the resistor R _{1 that} is generated.

Although the operation of the chopper circuit 4 is well known, it will be briefly described. During the ON period of the switching element Q ₃ , a current flows in the primary winding of the choke coil CH, energy is accumulated in the choke coil CH, and switching is performed. The energy stored in the choke coil CH is released during the off period of the element Q ₃ to charge the capacitor C ₁ . At this time, the output voltage of the rectifier circuit DB is set to 1 of the choke coil CH.
The voltage across the capacitor C ₁ is boosted with respect to the output voltage of the rectifier circuit DB by adding the voltage across the secondary winding.

[0005] Here, when the switching element Q ₃ is turned on again before the current flowing through the diode D ₁₀ of zero, the switching element Q ₃ takes the back electromotive current of the choke coil CH is a diode D ₁₀ and the switching element Q ₃ Stress increases. Therefore, the control circuit 5 controls so that the switching element Q ₃ is turned on again after the current stops flowing through the diode D ₁₀ . By this control, the current flowing through the choke coil CH has a triangular waveform, and the average value of the input current in each switching of the switching element Q ₃ becomes half the peak value of the current flowing through the choke coil CH. Therefore, if the peak value of the current flowing through the choke coil CH is put on a curve proportional to the sine wave voltage of the AC power supply AC, the chopper circuit 4 can be treated as a resistance load with respect to the AC power supply AC, and a high input can be obtained. You can get the power factor.

In order to make the current flowing through the choke coil CH into a triangular waveform as described above, the control circuit 5 switches the threshold value set based on the product of the input voltage and the output voltage to the chopper circuit 4 and switching. The peak value of the current flowing through the choke coil CH is determined by comparing it with the voltage across the resistor R ₁ corresponding to the current flowing through the element Q ₃ , and the diode is generated based on the induced voltage in the secondary winding of the choke coil CH. The presence or absence of a current flowing through D ₁₀ is detected. By providing the step-up type chopper circuit adopting such a control system in the preceding stage of the inverter device 1, it is possible to provide a highly efficient power supply device having a high input power factor.

As a power supply device, a structure as shown in FIG. 43 has also been proposed (Japanese Patent Laid-Open No. 58-209896).
Issue). In this configuration, without using the chopper circuit 4,
The capacitor C ₁ for smoothing the output side of the rectifier circuit DB is provided, the frequency for generating a voltage across the capacitor C ₁ as an input power source of the inverter device 1, the input or output side of the rectifier circuit DB at the inverter device 1 Transformer T
₁ and the capacitor C _{2 are used for} feedback superposition. Specifically, when superposing a high frequency wave on the output side of the rectifier circuit DB, as shown in FIG. 44, two diodes D connected in series between the output end of the rectifier circuit DB and the capacitor C _{1. 1} and D ₂ are inserted, and the high frequency generated in the inverter device 1 is superposed on the connection point of both diodes D ₁ and D ₂ .

In the structure shown in FIG. 44, the discharge lamp DL is used.
If the load is almost constant, the smoothing capacitor C
₁Voltage V across_dcBecomes almost constant (see Fig. 45).
The high frequency voltage generated from the inverter device 1 is also almost constant.
It Therefore, the high frequency generated in the inverter device 1
Primary winding n₁₁Transformer T being applied to₁Secondary winding
n_{twenty one}Voltage E₁Amplitude is almost constant (peak voltage is V
₁And). Here, first, the voltage wave of the AC power supply AC
Considering the vicinity of the zero-cross point of the shape (part A in FIG. 45)
Min), and the potential of the output terminal on the negative side of the rectifier circuit DB as the reference potential.
Therefore, the output voltage of the rectifier circuit DB can be regarded as 0V.
It At this time, the transformer T₁Secondary winding n_{twenty one}Voltage across
E₁Is V on the negative side of the reference potential₁If you shake only
Sensor C₂Voltage V across_C2Is V₁become. Then the transformer
T₁Secondary winding n_{twenty one}Voltage E ₁Is V on the positive side₁Just shake
Both diodes D₁, D₂To the reference potential of the connection point of
Potential V_S(= V_C2+ E₁) Is 2V₁become. here
And capacitor C₁Voltage V across_dc2V₁Higher than
If set (V_dc> 2V₁) Diode D₂Is off
Kept in. Transformer T₁Secondary winding n_{twenty one}Voltage E
₁Is V again on the negative side₁When only swinging, both diodes D₁，
D₂Potential V at the connection point_SBecomes 0V again. After all, Figure 4
As shown in 6 (a), both diodes D₁, D₂Connection
Point potential V_SIs the voltage V₁Amplitude biased only by V₁of
It becomes a vibration waveform. In short, the output voltage of the rectifier circuit DB is
When it is near 0 V, the capacitor C
₂Is the charging / discharging current I_C2Does not flow, diode D₁Pass through
Input current I from rectifier circuit DB _D1(Fig. 46 (c)) and
Iodo D₂Capacitor C passing through₁Charging current I to
_D2(Fig. 46 (d)) does not flow either.

On the other hand, considering a portion of the voltage waveform of the AC power supply AC other than the zero-cross point (the portion B and C in FIG. 45).
Part), a voltage V _in is generated between the output terminals of the rectifier circuit DB. At this time, when the voltage E ₁ across the secondary winding n ₂₁ of the transformer T ₁ swings to the negative side by V _{1, the} voltage across the capacitor C ₁ becomes V _in + V ₁ . Next, when the voltage across E ₁ of the secondary winding n ₂₁ of the transformer T ₁ is deflected by V ₁ to the positive potential V _S of the connection point both diodes D _1, D ₂ and would V _in + 2V ₁ To do. Here, the voltage V _dc across the capacitor C ₁
Is lower than V _in + 2V ₁ , charging current I _D2 flows to the capacitor C ₁ through the diode D ₂ during the period of V _in + 2E ₁ ≧ V _dc (FIG. 47 (d), FIG. 48 (d)). During the period, the potential V _{S at} the connection point of both diodes D ₁ and D ₂
Is held at V _dc (ignoring the voltage drop across diode D ₂ ). Voltage across the secondary winding n ₂ of the transformer T ₁ E ₁
Is reduced to V _C2 + E ₁ ≤V _in , both diodes D
_Since the potential V _{S at} the connection point of ₁ and D ₂ is maintained at V _in , the current I _D1 flows from the rectifier circuit DB to the capacitor C ₂ through the diode D ₁ in this period (FIG. 47).
(C), FIG. 48 (c)). That is, looking at the charging / discharging current I _C2 to the capacitor C ₂ , FIG.
As shown in 8 (b), the discharge current flows during the period of V _in + 2E ₁ ≧ V _dc , and the charging current flows during the period of V _C2 + E ₁ ≦ V _in . After all, both diodes D ₁ and D ₂
The electric potential V _{S at} the connection point is in the form shown in FIGS. 47 (a) and 48 (a).

Therefore, as shown in FIG. 49 (c),
If the voltage E ₁ across the secondary winding n ₂₁ of the transformer T ₁ is constant, as shown in FIG. 49 (a), the diodes D ₁ , D
The voltage waveform of the potential V _S at the connection point of ₂ is ₂ of the transformer T ₁ .
A high-frequency voltage waveform having an upper limit value of V _dc and a lower limit value of V _in is obtained according to the voltage E ₁ across the secondary winding n ₂₁ . The voltage V _C2 across the capacitor C ₂ reflects the potential V _S at the connection point of the diodes D ₁ and D ₂ (that is, V _C2 = V _S −
E ₁ ), the voltage waveform becomes as shown by the solid line in FIG. 49 (b), the discharge current to the capacitor C ₁ is made to flow during the period of V _dc or more, and the rectification circuit DB is output during the period of (V _dc −V ₁ ) or less. The charging current will be applied. As a result, the input current from the rectifier circuit DB has a shape substantially similar to the current waveform of the AC power supply AC, and the input power factor is good. After all, Figure 4
In the configuration shown in FIG. 3 and FIG. 44, the input power factor is improved without using a chopper circuit. Therefore, the configuration is simplified, and a compact, lightweight and highly efficient power supply device can be obtained.

[0011]

In the conventional configuration of FIG. 42 described above, the degree of stabilization of the voltage across the capacitor C ₁ and improvement of the input power factor depends on whether the switching element Q ₃ is turned on or off by the control circuit 5. It depends on the timing. Therefore, since it is one that is to detect the voltage of the multi-point (four places) in determining the timing of the on-off switching element Q _3, the configuration of the control circuit 5 is turned very complicated It has the problem of becoming expensive. Further, since the switching element Q ₃ is turned off when the passing current of the choke coil CH reaches its peak, a large switching noise is generated. Here, since switching is also performed in the inverter device 1, switching noise is generated. Since the operating frequency is generally different between the inverter circuit 1 and the chopper circuit 4, the noise level is high and the noise frequency is wide. become. As a result, the cost of noise countermeasures increases and the wiring paths are also limited so as to reduce noise, resulting in problems such as an increase in cost and a reduction in design freedom.

On the other hand, in the conventional configuration shown in FIGS. 43 and 44, since the chopper circuit is not used, regarding the noise, only the switching noise of the inverter device 1 needs to be considered, and the configuration shown in FIG. 42. Compared to, it is advantageous in terms of cost required for noise suppression and design flexibility. However, when the output of the inverter device 1 is changed according to the load of the discharge lamp DL, the following problems will occur.

For example, when the discharge lamp DL needs to be preheated before starting, the inverter device 1 changes the operating frequency and the duty ratio of ON / OFF at the time of switching to change the power supplied to the discharge lamp DL. ,
Controls such as preheating, starting, and lighting will be performed. When performing such control, the assumption that the voltage across E ₁ of the secondary winding n ₂₁ of the transformer T ₁ as described above is substantially constant no longer satisfied, the capacitor C for the resulting smooth
There arises a problem that the voltage across ₁ does not become constant. Further, the power consumption of the discharge lamp DL is different between the preheating time and the lighting time, and the input current to the inverter device 1 is different. Consequently, the input current from the AC power supply AC changes, which improves the input power factor. There is a problem that the degree of preheating differs from that during lighting. This problem also occurs when the dimming control is performed or when the number of lighting of the discharge lamp DL changes.

Further, in the configuration shown in FIGS. 43 and 44, the input current from the AC power supply AC to the rectifier circuit DB flows only for a part of the half cycle for each cycle of the high frequency voltage which is feedback superimposed. The peak value of the input current becomes relatively high, and the inductor L ₀ provided to suppress noise
It is necessary to use a high-capacity filter circuit NF including the capacitor C ₀ and the capacitor C _0, and as a result, there is a problem that the filter circuit NF becomes large in size.

More specifically, when the high frequency voltage swings to the negative side with respect to the reference voltage, the charging current to the capacitor C ₂ flows through the diode D ₁ and this charging current becomes the input current from the AC power supply AC. Further, when the high frequency voltage swings to the positive side, a current flows through the diode D ₂ only during the period when the anode potential of the diode D ₂ is higher than the voltage across the capacitor C ₁ , and this current causes the capacitor C ₁ to be high frequency. Will be charged. Therefore, the voltage V _dc smoothed by the capacitor C ₁
Is a DC voltage with few ripple components. next,
When the high-frequency voltage swings to the negative side, the input current flows again from the AC power supply AC, but the period in which the input current flows is shorter than the half cycle period of the high-frequency voltage, so an input current with a high peak value flows. Similarly, the peak value of the charging current to the capacitor C ₁ becomes high, and
_Since the calorific value of ₁ becomes large, a capacitor having a large capacity is required as the capacitor C ₁ .

The present invention has been made in view of the above problems, and a first object thereof is to provide a power supply device in which the peak value of the input current and the charging current to the smoothing capacitor is reduced. The second object is to provide a power supply circuit that suppresses a change in the input power factor when the output of the inverter device changes.

[0017]

According to a first aspect of the present invention, a rectifier circuit for rectifying an AC power source, a smoothing capacitor for smoothing an output voltage of the rectifier circuit, and high-frequency power to a load using a voltage across the smoothing capacitor as a power source. in the power supply device including an inverter apparatus for outputting, provided a high-frequency feedback means having a plurality of paths for feeding back superimposed on the input side of the part of the inverter device of a high-frequency voltage coupled to a load, the high-frequency feedback means
Is a capacitor for feedback superposition inserted on the path.
And insert it on the charging path of the smoothing capacitor from the AC power supply
From the smoothing capacitor to the capacitor for feedback superposition
And a diode for blocking the flow .

According to a second aspect of the present invention, in the first aspect of the present invention, the high-frequency feedback means takes out the high-frequency voltage coupled to the load from one location and feeds back and superimposes the high-frequency voltage at a plurality of locations on the input side of the inverter device. Characterize. According to the invention of claim 3, in the invention of claim 1, the high frequency feedback means takes out the high frequency voltage from the inverter device and distributes the high frequency voltage so as to feed back and superimpose the high frequency voltage on a plurality of positions on the input side of the smoothing capacitor. It is characterized by comprising: a distribution means and a phase shift means for making the phases of the voltage waveforms of the high-frequency voltage distributed by the distribution means different from each other.

According to a fourth aspect of the present invention, in the first aspect of the present invention, the high-frequency feedback means extracts the high-frequency voltage coupled to the load from a plurality of points and feeds back and superimposes the high-frequency voltage to a single point on the input side of the inverter device. Characterize. According to a fifth aspect of the present invention, in the first aspect of the present invention, the high-frequency feedback means extracts high-frequency voltages from a plurality of locations of the inverter device, and adds the high-frequency voltages as appropriate to add feedback means to the input side of the smoothing capacitor. The high frequency voltage is taken out at a place where the voltage amplitude of the high frequency voltage added by the adding means changes according to the output of the inverter device.

According to a sixth aspect of the invention, in the second to fifth aspects of the invention, the high frequency feedback means feedback-superimposes a high frequency voltage between the rectifying circuit and the smoothing capacitor. The invention of claim 7 is characterized in that, in the invention of claims 2 to 5, the high-frequency feedback means feedback-superimposes a high-frequency voltage between the AC power supply and the rectifier circuit.

According to an eighth aspect of the present invention, in the second or third aspect of the invention, the high frequency feedback means is provided at each position between the AC power supply and the rectifying circuit and between the rectifying circuit and the smoothing capacitor. It is characterized in that a high frequency voltage is fed back and superimposed. According to the invention of claim 9, in the invention of claim 6, the diode is inserted in a forward direction between a rectifier circuit and a smoothing capacitor, and a high-frequency voltage is superimposed between the rectifier circuit and the diode. .

According to a tenth aspect of the invention, in the invention of the first aspect, the inverter device includes a resonance circuit including an inductor and a capacitor, and the high frequency feedback means extracts the voltage across the capacitor of the resonance circuit as a high frequency voltage. Characterize. According to the invention of claim 11, in the invention of claim 1, the inverter device includes a resonance circuit including an inductor and a capacitor, and the high frequency feedback means extracts a voltage across the inductor of the resonance circuit as a high frequency voltage. .

According to a twelfth aspect of the present invention, in the first aspect of the present invention, the inverter device includes a resonance circuit, and the high-frequency feedback means includes a current transformer having a primary winding inserted in a closed circuit through which a resonance current flows. It is characterized in that the high frequency voltage induced in the secondary winding of the current transformer is taken out. A thirteenth aspect of the present invention is characterized in that, in the first aspect of the present invention, the inverter device includes a transformer that outputs high-frequency power to a load, and the high-frequency feedback means extracts the high-frequency voltage via the transformer.

(Basic Structure) The invention of claim 3 basically adopts the structure of FIG. 1, in which a part of the high frequency power generated in the inverter device 1 is input to the rectifier circuit DB. in a conventional configuration similar to the configuration shown in FIG. 43 to return to the side or output side, capacitor C ₁ and the high-frequency voltage
Is characterized in that a plurality of paths B ₁ , ..., Bn for feedback superposition are provided on the input side of. That is, the high frequency voltage extracted from the inverter device 1 is applied to the input side of the capacitor C ₁ through the plurality of paths B ₁ , ..., Bn by appropriately distributing by the distribution means, and each path B ₁ ,
The phase shift means is provided so as to make the phase of the high frequency voltage of Bn different. Further, when the high frequency voltage is superimposed on the output side of the rectifier circuit DB, the diode D for blocking the reverse current from the capacitor C ₁ is provided.
When the high frequency voltage is superimposed on the input side of B, the diode D becomes unnecessary.

The invention of claim 5 basically adopts the configuration of FIG. 2, wherein a part of the high frequency power generated in the inverter device 1 is input to the rectifier circuit DB or to the output side. returning to the conventional configuration similar to the configuration shown in FIG. 43, the path F ₁ to return superimposed on the input side of the capacitor C ₁ by removing a high-frequency voltage from a plurality of positions of the inverter device 1, ..., a plurality of Fm It is characterized by the fact that The high frequency voltage applied to the input side of the capacitor C ₁ through each of the paths F ₁ , ..., Fn is appropriately added by the adding means, and the added voltage fluctuates according to the output of the inverter device 1. Further, when a high frequency voltage is superimposed on the output side of the rectifier circuit DB, a diode D for blocking the reverse current from the capacitor C ₁
However, when the high frequency voltage is superimposed on the input side of the rectifier circuit DB, the diode D becomes unnecessary.

According to the invention of claim 3 and the invention of claim 5, as shown in FIG. 3, the high frequency voltage is taken out from a plurality of locations of the inverter device 1 and is superimposed on a plurality of locations on the input side of the capacitor C _1. Including the configuration described above, the high-frequency voltages extracted from the inverter device 1 are appropriately added and appropriately distributed, and the phases of the distributed high-frequency voltages are made different from each other. For the diode D, is necessary in the case of superimposing a high-frequency voltage to the output side of the rectifier circuit DB, when superimposed on the input side of the rectifier circuit DB is rectified times
It is unnecessary because the path DB functions as the diode D and blocks backflow .

[0027]

The operation will be described in accordance with the basic structure shown in FIGS. When a configuration in which high frequency voltages having different phases are fed back and superimposed at a plurality of points on the input side of the capacitor C ₁ as shown in FIG. 1, one cycle of the high frequency voltage is applied to each of the paths B ₁ , ..., Bn. The input current from the AC power supply AC and the charging current to the capacitor C ₁ flow in a part of a half cycle of each of the paths B ₁ , ..., B.
Since the phase of the high frequency voltage of n is different, the input current and the charging current flow at different timings, and as a result, the energization period of the input current and the charging current can be extended as compared with the conventional configuration. is there. Therefore,
If the current consumption in the inverter device is the same, the peak values of the input current and the charging current can be reduced as compared with the conventional configuration. This leads to a reduction in the capacity of the filter circuit NF if the filter circuit NF is provided to reduce noise to the AC power supply AC, and the heat generation of the capacitor C ₁ is suppressed, so that the capacitor C ₁ is suppressed. It also leads to lower capacity.

As shown in FIG. 2, if the high frequency voltage obtained by appropriately adding the high frequency voltages of the plurality of paths F ₁ , ..., Fm is changed according to the change of the output of the inverter device 1, for example, at the time of preheating and lighting. change in the output of the inverter device 1 in the time, the output variation of the inverter device 1 when the dimming control, the time of such change in the output of the inverter device 1 in accordance with the load of severity, such as the discharge lamp DL, the capacitor C ₁ Since the high frequency voltage superimposed on the input side is changed, it is possible to suppress the fluctuation of the input power factor as a result. For example, when the output of the inverter device 1 is large, the charge in the capacitor C ₁ tends to be insufficient and the input current is about to increase. However, by increasing the superimposed high frequency voltage, the charge in the capacitor C ₁ is compensated for. In this way, the increase of the input current is suppressed. This makes it possible to suppress fluctuations in the input power factor.

[0029] in the case where the configuration of Figure 3, will have both effects of the configuration of the FIG. 2 arrangement Figure 1, the input current and the charging current to the capacitor C ₁ from the AC power source AC It is possible to suppress the peak value and suppress the change of the input power factor with respect to the fluctuation of the output of the inverter device 1.

[0030]

【Example】

(Embodiment 1) In this embodiment, as shown in FIG. 4, a filter circuit N for blocking a high frequency from an AC power supply AC such as a commercial power supply.
It is input to the rectifier circuit DB consisting of a diode bridge through F, the full-wave rectified output from the rectifier circuit DB is given to the smoothing capacitor C ₁ through the high frequency feedback circuit 2, and the direct current from both ends of this capacitor C ₁ to the inverter device 1 is applied. It has a structure adapted to obtain a power source. That is, a configuration is adopted in which the high frequency generated in the inverter device 1 is feedback superimposed on the output side of the rectifier circuit DB. Further, a discharge lamp DL is connected as a load to the output of the inverter device 1 so that the discharge lamp DL is lit at a high frequency. The filter circuit NF includes an inductor L ₀ and a capacitor C _0, and the high frequency generated in the inverter device 1 is an AC power source A.
It is prevented from being sent to the C side.

In the high frequency feedback circuit 2, the high frequency voltage generated in the inverter device 1 is applied to the primary winding n ₁₁ , and a transformer T ₁ having two secondary windings n ₂₁ and n ₃₁ and paired one by one. Four diodes D _{1 to} D _4, which are connected in series and each series circuit is inserted between the positive side output terminal of the rectifier circuit DB and the capacitor C _1, and one secondary winding n of the transformer T _1. A capacitor C ₂ connected between one end of ₂₁ and a connection point of the diodes D ₁ and D ₂ connected in series;
_The capacitor C ₃ is connected between one end of the other secondary winding n ₃₁ of ₁ and the connection point of the diodes D ₃ and D ₄ connected in series. Both secondary windings n ₂₁ and n ₃₁ of the transformer T ₁ are connected in series and the rectifier circuit DB
Is connected to the negative output terminal of. Also, the two secondary windings n
₂₁ and n ₃₁ are connected in series with one winding start end connected to the other winding end. In this way, the high-frequency feedback circuit 2 has the two feedback circuits 2A and 2B, and both high-frequency feedback circuits 2A and 2B respond to the high-frequency voltage generated in the inverter device 1.
A and 2B are configured to operate in opposite phases. In other words, a secondary winding n ₃₁ is added to the transformer T ₁ in addition to the conventional configuration shown in FIG. 44 which has a configuration corresponding to one feedback circuit 2A, and the diodes D ₃ and D ₄ and capacitors C ₂ and C ₃ are added as a feedback circuit 2B.

The individual operation of each feedback circuit 2A, 2B is shown in FIG.
44, which is similar to the conventional configuration shown in FIG.₁of
Secondary winding n_{twenty one}Voltage E₁Every half cycle of
Sensor C₂, C₃To capacitor C₁Charging current to the
And the output current of the rectifier circuit DB, the capacitor C₂，
C₃And the state of being charged are alternately repeated. Also,
In the vicinity of the zero cross point of the AC power supply AC, the rectifier circuit DB
The force current is set to stop. However, the transformer
T₁Secondary winding n_{twenty one}, N₃₁And each capacitor C₂, C₃of
By connecting the connections in reverse polarity,
SA C₂, C₃The timing of charging and discharging is shown in FIG.
As shown in (b),
Therefore, the diode D is on the upper side in FIG.₁of
Current I_D1, The lower side is diode D₂Current I_D2Shows the figure
5 (b) is the diode D on the upper side₃Current I_D3, The lower side is
Iodo D_FourCurrent I_D4, And one capacitor C
₂(C ₃) Is charged while the other capacitor
C₃(C₂) Is designed to discharge. That is,
As shown in FIG. 5C, the transformer T₁Secondary winding n_{twenty one}
Rectifier circuit D for every half cycle of high-frequency voltage induced in
B to diode D₁, D₃Through one of the
Sensor C₂, C₃Current to charge the capacitor C ₂, C
₃To diode D₂, D_FourThrough one of the
Sensor C₁Both the current to charge the will flow.
In short, the transformer T₁Is the high frequency voltage divided into two
Secondary winding n_{twenty one}, N₃₁To
Capacitor C₁Is connected to the opposite polarity with respect to
Therefore, it also functions as a phase shifting means.

According to this structure, if the power consumption of the discharge lamp DL is the same as that of the conventional structure shown in FIG. 44, the effective value of the input current from the AC power supply AC becomes the same, but the high frequency half By alternately turning on the diodes D ₁ and D _{3 in} each cycle, the period during which the input current from the AC power supply AC flows is about twice as long as that in the conventional configuration, and as a result, the peak value of the input current is suppressed. be able to. That is,
Since the peak value is small and the cycle of the input current flows is short, the filter circuit NF can be downsized. Further, since the peak value of the charging current to the capacitor C ₁ is also suppressed, the amount of heat generated by the capacitor C ₁ is reduced, and further, the ripple component of the smoothed voltage is reduced, so that the capacitance of the capacitor C ₁ can be reduced. it can.

In this embodiment, the high frequency feedback circuit 2 has
Lance T₁I am using a transformer T₁Without using
The high frequency generated by the converter device 1₂, C
₃The same effect can be obtained by giving it directly to
Wear. (Embodiment 2) This embodiment is a diagram of the configuration of Embodiment 1.
As shown in 6, the transformer T of the high frequency feedback circuit₁Is in
With the configuration that doubles as the output transformer of the barter device 1.
is there. That is, the transformer T₁Connect the discharge lamp DL to
Secondary winding n for₄₁Is provided. This inva
The data device 1 includes a pair of transistors that are switching elements.
Star Q₁, Q₂Connect in series between the emitter and collector of
High voltage side transistor Q₁Between the emitter and collector of
Capacitor C for DC cut_FiveAnd trance T₁Primary volume
Line n₁₁And inductor L₁And connect the series circuit in parallel with
Rani trance T ₁Primary winding n₁₁Capacitor C₆Average
It has a configuration of column connection. Also, each transistor Q₁，
Q₂A diode D for freewheeling between the emitter and collector of
_Five, D₆Are connected in anti-parallel. Transistor Q₁
Inverts the output signal from the control circuit 3 by the inverting circuit NT
Controlled by the signal₂From the control circuit 3
Is controlled by the output signal of
₁, Q₂Are turned on and off alternately so that they do not turn on at the same time.
It will be turned off.

The operation of the inverter device 1 will be briefly described. When the transistor Q ₂ is turned on, the capacitors C ₁ to C ₅ -C ₆ and the primary winding n ₁₁ of the transformer T ₁ are connected. Parallel circuit-inductor L ₁
-Current flows through the path of transistor Q ₂ . Also,
When the transistor Q ₁ is turned on, a current flows from the capacitor C ₅ through a path of a parallel circuit of the transistor Q ₂ -inductor L ₁ -capacitor C ₆ and the primary winding n ₁₁ of the transformer T ₁ . Since a series resonance circuit is formed by the capacitors C ₅ and C ₆ and the inductor L ₁ , a sinusoidal voltage is generated across the capacitor C _{6 and} the primary winding n ₁₁ of the transformer T ₁ is generated. Is applied with a sinusoidal voltage. In this way, the secondary windings n _{21 to} n _{41 of the} transformer T ₁ are
The high frequency voltage is output to. Here, since the discharge lamp DL is insulated from the AC power supply AC by the transformer T ₁ , it is possible to prevent an electric shock accident when the discharge lamp DL is replaced. Other configurations and operations are the same as those in the first embodiment, and thus the description thereof is omitted.

Although the series resonance circuit is connected in parallel to the transistor Q ₁ in this embodiment, a series resonance circuit may be connected in parallel to the transistor Q ₂ , and a capacitor C ₅ , which constitutes the series resonance circuit, may be formed. The order of connecting C ₆ and inductor L ₁ is not particularly limited. Further, if insulation by the transformer T ₁ is unnecessary, the following connection relationship may be used. That is, if the discharge lamp DL having a filament is used and each filament is inserted and connected between the parallel circuit of the capacitor C ₆ and the primary winding n ₁₁ of the transformer T ₁ , and the capacitor C ₅ and the inductor L _1. , By monitoring the output voltage of the secondary windings n _{21 to} n ₄₁ of the transformer T ₁ , it becomes possible to easily detect the disconnection of the filament.

(Embodiment 3) In this embodiment, as shown in FIG. 7, two inductors are used instead of the inductor L ₁ in the configuration of the second embodiment.
The transformer T ₁ is a current transformer having a number of secondary winding n _21, n ₃₁ is provided, the output to the discharge lamp DL is a in which they were taken out using another transformer T _2, the transformer T ₁ Of the primary winding n ₁₁ of the inductor L in the second embodiment.
_Acts as ₁ . In this circuit configuration, the high frequency wave from the inverter device 1 is taken out through the transformer T ₁ whose primary winding n ₁₁ also serves as the inductor of the series resonance circuit, but operates similarly to the second embodiment. Other configurations are similar to those of the second embodiment.

Here, in the present embodiment, a sinusoidal high frequency voltage is applied to the primary windings n ₁₁ and n ₁₂ of the transformers T ₁ and T ₂ using a series resonance circuit, but a parallel resonance circuit is used. It may be used, or a series resonance circuit and a parallel resonance circuit may be used in combination. (Embodiment 4) In this embodiment, as shown in FIG. 8, a primary winding of a transformer T ₁ is connected in series with an inductor L ₁ . Even with this configuration, the secondary windings n ₂₁ , n of the transformer T ₁ are
High frequency can be taken out from ₃₁ . Note that, here, the connection point between the transistors Q ₁ and Q ₂ and the inductor L ₁
While inserting the primary winding n ₁₁ of the transformer T ₁ between the insertion position of the primary winding n ₁₁ of the transformer T _1, the resonance current is within the loop circuit including the series resonant circuit Any place that flows can be used. Other configurations and operations are similar to those of the second embodiment.

(Embodiment 5) In this embodiment, as shown in FIG. 9, a series circuit of a primary winding n ₁₁ of a transformer T ₁ and a capacitor C ₇ is provided between the emitter of one of the transistors Q _{2 and} the equator. The transformer T ₁ is connected in parallel, and high frequencies are taken out from the secondary windings n ₂₁ and n _{31 of the} transformer T ₁ . Even with this configuration, the secondary windings n ₂₁ , n _{31 of the} transformer T ₁
Higher frequencies can be taken out. Moreover, the high frequency is not taken out from the series resonance circuit, but the transistor Q
_{Since the} high frequency is taken out between the emitter and collector of the _second transistor, the influence on the resonance current is reduced, and the ripple component of the voltage across the secondary winding n ₂₂ of the transformer T ₂ can be reduced. As a result, the luminous efficiency of the discharge lamp DL is improved. Other configurations and operations are similar to those of the second embodiment.

In this embodiment, the low voltage side transistor Q ₂
Although the series circuit of the primary winding of the transformer T ₁ and the capacitor C ₇ is connected in parallel with, a similar configuration may be applied to the transistor Q ₁ on the high voltage side. (Embodiment 6) In this embodiment, the inverter device 1 is of a one-stone type, and as shown in FIG. 10, the transistor Q of the inverter device 1 in the embodiments 2 to 5 is used.
Instead of _1, it is provided with a parallel resonant circuit of the primary winding n ₁₁ and capacitor C ₈ of the transformer T _1.

In this configuration, the transistor Q ₂ is turned on.
When the capacitor C ₈ is turned off, a resonance voltage is generated across the capacitor C ₈ , and the resonance voltage causes a resonance current to flow in the series resonance circuit composed of the capacitors C ₅ , C ₆ and the inductor L ₁ . Other configurations and operations are similar to those of the second embodiment.
Even in the case of using the one-stone type inverter device 1 as in the present embodiment, it is not limited to the configuration in which the high frequency voltage is taken out from the parallel resonant circuit, and the capacitors C ₆ , C ₈ and the inductor L _{1 are used.} , it may be taken out a high-frequency voltage from the transformer or T _2.

The type of the inverter device 1 is as follows.
In addition to the serial inverter circuit in which the transistors Q ₁ and Q ₂ as switching elements are connected in series as in the _second to fifth embodiments and the single-stone inverter circuit as in the sixth embodiment, what kind of inverter device 1 is used. Alternatively, it may be, for example, an L-push pull inverter circuit or a full bridge inverter circuit. In short, as long as it is a circuit configuration for converting a DC voltage into a high frequency voltage and the high frequency voltage can be taken out to the high frequency feedback circuit 2, any circuit form may be used. Further, although the resonance circuit is used on the side of the discharge lamp DL, this is not always necessary.

(Embodiment 7) In each of the above embodiments, the inverter device 1 is provided by the high frequency feedback circuit on the output side of the rectifier circuit DB.
However, in the present embodiment, as shown in FIG. 11, two rectifier circuits DB ₁ and DB ₂ are superposed.
Is provided, and a high-frequency feedback circuit for feedback-superimposing a high frequency on the input side of each of the rectifier circuits DB ₁ and DB ₂ is configured.
That is, the output side of the filter circuit NF is branched into two paths, and the rectified circuits DB ₁ and DB ₂ are connected to the branched paths via the inductors L ₂ and L ₃ , respectively.
The output of B ₁ and DB ₂ is merged again and connected to one smoothing capacitor C ₁ . Further, the secondary windings n ₂₁ and n ₃₁ of the transformer T ₁ that extracts the high frequency voltage from the inverter device 1 are commonly connected at one ends thereof having opposite polarities, and are connected to the AC power supply AC and the rectifier circuits DB ₁ and DB ₂ . Is connected to the connection point, and the other end is connected to a capacitor C ₂ ,
Each inductor L ₂ , L ₃ and rectifier circuit D via C _3.
It is connected to the connection point with B ₁ and DB ₂ .

According to the above configuration, the high frequency voltage generated in the inverter device 1 and induced in the secondary windings n ₂₁ and n ₃₁ of the transformer T ₁ is superimposed on the voltage waveform of the AC power supply AC, and the rectifier circuit DB is formed. ₁ , input to DB ₂ . Here, since the secondary windings n ₂₁ and n ₃₁ of the transformer T ₁ are connected to the AC power source AC in opposite polarities to each other, both rectifier circuits DB ₁ and D ₁ are connected.
The phases of the voltage waveforms of the high frequency voltage input to B ₂ are different by 180 degrees.

Now, pay attention to the rectifier circuit DB ₁ with the negative electrode of the capacitor C ₁ as the reference potential. Where AC power supply AC
The period of the voltage waveform is a positive side, when the induced voltage to the secondary winding n ₂₁ of the transformer T ₁ is deflected to the negative side than the reference potential, a charging current through the inductor L ₂ with the capacitor C ₂ When the current flows and the induced voltage of the secondary winding n ₂₁ swings to the positive side of the reference potential, a counter electromotive voltage is generated in the inductor L ₂ and the capacitor C ₂ is charged. Where rectifier circuit D
If the voltage drop at B ₁ is ignored, the induced voltage in the secondary winding n ₂₁ swings to the positive side of the reference potential, and the induced voltage in the secondary winding n _{21 and} the voltage across the capacitor C ₂ are summed. Is the capacitor C
During the period when the voltage across both ends of ₁ is about to be exceeded, the capacitor C
_A charging current flows from ₂ to the capacitor C ₁ through the rectifier circuit DB ₁ . Here, since the rectifier circuit DB ₁ performs full-wave rectification, almost the same operation is performed during the period when the voltage waveform of the AC power supply AC is on the negative side.

The operation of the rectifier circuit DB ₂ is similar, but the secondary windings n ₂₁ and n _{31 of the} transformer T ₁ are connected to the capacitors C ₂ and C _{3 in} opposite polarities. While one capacitor C ₂ (C ₃ ) is being charged by the AC power supply AC, the other capacitor C 2
₃ (C ₂ ) will flow the charging current to the capacitor C ₁ . Therefore, the input current from the AC power supply AC flows in every half cycle of the high frequency generated from the inverter device 1, and as in the first embodiment, the inverter device 1
If the current consumption in the same is the same, the peak value of the input current can be made smaller than that in the conventional configuration. Further, the peak value of the charging current to the capacitor C ₁ is also reduced, and the first embodiment
The same effect as can be obtained.

In this embodiment, the inductor L₂，
L₃, Capacitor C₂, C₃, Transformer T₁Secondary winding
n_{twenty one}, N₃₁High-frequency feedback circuit using
₁, DB₂It is also possible to apply to the output side of. this
If each capacitor C₂, C ₃And capacitor C₁Between
Requires a diode in. Other configurations and operations performed
Similar to Example 1.

(Embodiment 8) In this embodiment, as shown in FIG. 12, the output side of the filter circuit NF is branched into two paths, and the rectified circuits DB ₁ and D are respectively connected to the branched paths.
B ₂ is provided, and the outputs of both rectifier circuits DB ₁ and DB ₂ are merged and a smoothing capacitor C ₁ is connected. The secondary windings n ₂₁ and n ₃₁ of the transformer T ₁ are connected to a filter circuit. The capacitors C ₂ and C ₃ are inserted between the NF and the rectifier circuits DB ₁ and DB ₂ , respectively, and capacitors C ₂ and C ₃ are connected to the input sides of the rectifier circuits DB ₁ and DB ₂ , respectively. Also,
The secondary windings n ₂₁ and n ₃₁ are provided with a filter circuit NF and each rectifier circuit D.
B ₁ and DB ₂ are connected so as to have opposite polarities.

That is, the inductor L in the seventh embodiment
_{In place of 2} , L ₃ , secondary windings n ₂₁ , n ₃₁ of the transformer T ₁ are connected. Also in this configuration, the seventh embodiment
Similarly to the AC power supply AC voltage waveform, the transformer T ₁
2 in a path by superimposing the secondary winding n _21, the induced high frequency voltage n _31, and correspond to the respective rectifier DB _1, DB ₂ of
Since the secondary windings n ₂₁ and n ₃₁ are connected in opposite polarities, the input current from the AC power supply AC flows and the capacitor C simultaneously with each half cycle of the high frequency generated from the inverter device 1.
The charging current to ₁ flows. That is, as compared to the case where the input current and the charging current alternately flow every half cycle of high frequency as in the conventional configuration, the peak value of both currents can be reduced.

The configuration in which the secondary windings n ₂₁ and n ₃₁ of the transformer T ₁ are inserted in the charging path of the capacitor C ₁ as in this embodiment is arranged on the output side of the rectifier circuits DB ₁ and DB _2. Good. Further, in the seventh and eighth embodiments, the rectifier circuit DB
_{Although the} high frequency is superposed on the input side of ₁ and DB ₂ , the high frequency may be superposed on both the input side and the output side.

(Embodiment 9) In this embodiment, as shown in FIG. 13, the diode D ₃ in Embodiment 1 is replaced by an inductor L.
The configuration is the same as that of the first embodiment except that the _third configuration is replaced. By the way, the discharge lamp DL
In the case where preheating is performed by using a filament having a filament, the energy supplied to the discharge lamp DL is reduced and preheating can be performed by increasing the operating frequency of the switching element provided in the inverter device 1. It is assumed that a configuration is adopted in which the energy supplied to the discharge lamp DL is increased and the lamp is lit when lowered (if the resonance circuit is provided in the output section of the inverter device 1, the discharge lamp D
The energy supplied to L can be changed according to the output frequency of the inverter device 1). In the case of performing such control, if the frequency generated from the inverter device 1 becomes low, the period during which the input current from the AC power supply AC and the charging current to the smoothing capacitor C ₁ flow becomes long. The input current will decrease.
That is, since the high-frequency input current decreases despite the increase in the output to the discharge lamp DL, the input current is distorted and the input power factor is lower during lighting than during preheating.

In order to solve such a problem, this embodiment adopts the above-mentioned structure. That is,
A series resonance circuit is formed by the capacitor C ₃ and the inductor L _3, and the voltage of the capacitor C ₃ is changed according to the change in the frequency of the high frequency output from the inverter device 1.
It changes like. Now, assuming that the resonance frequency of the series resonance circuit of the capacitor C ₃ and the inductor L ₄ is f ₀ and the frequency of the high frequency from the inverter device 1 is higher than the resonance frequency f ₀ , the high frequency generated from the inverter device 1 is The higher the frequency is, the less energy is given to the capacitor C ₁ from the inverter device 1. Therefore, in the inverter device 1, the operating frequency is f ₁ during preheating.
And then, the operating frequency at the time of lighting the shall be set to _{f 2 (f 0 <f 2} <f 1), supply energy to the capacitor C ₁ high frequency decreases during preheating, low frequency at the time of operating capacitor C ₁ Since the energy supplied to the inverter device 1 increases, the high frequency input increases or decreases so as to correspond to the state of the discharge lamp DL that is the load of the inverter device 1. As a result, the input power factor can be improved regardless of the load of the inverter device 1.

As described above, in the case where the energy supplied to the load is changed according to the change in the operating frequency of the inverter device 1, in order to improve the input power factor, the series resonance circuit is used as described above and the parallel resonance circuit is used. A resonance circuit may be used, and in any case, an element that causes a change in current / voltage according to a change in frequency may be used. Note that not only the inductor L ₃ may be provided in place of the diode D ₃ , but an inductor may be provided in place of the diode D ₁ . That is, even if it is configured to have two series resonance circuits, the same operation is performed. Although the case has been described where the energy supplied to the load increases as the operating frequency of the inverter device 1 decreases, as long as the energy supplied to the load changes as the operating frequency of the inverter device 1 changes. By using an element whose current or voltage changes with the change of its frequency, the same effect as that of this embodiment can be obtained.

(Embodiment 10) In Embodiment 1, the path on the output side of the rectifier circuit DB to the smoothing capacitor C ₁ is branched into two paths, and the high frequency generated in the inverter device 1 is generated for each path. However, in the present embodiment, as shown in FIG. 15, in the configuration shown in FIG. 44, instead of the transformer T ₁ , high frequencies are taken out from two different places of the inverter device 1 and both high frequencies are given. It is configured such that two transformers T ₃ and T ₄ for adding are added. That is, the primary windings n _{13 of} the transformers T ₃ and T ₄ ,
n ₁₄ are respectively connected to different portions of the inverter device 1, secondary windings n ₂₃ and n ₂₄ are connected in series, and this series circuit is connected in series to the capacitor C ₂ .

Here, the primary windings n ₁₃ and n ₁₄ of the transformers T ₃ and T ₄ are connected to the inverter device 1 at locations where their phases change depending on the magnitude of the output.
That is, in the left half of FIGS. 16 and 17, the output of the inverter device 1 is small and the secondary windings n of the transformers T ₃ and T ₄ are n.
Induced voltages E ₃ and E _{4 of 23} and n ₂₄ ((a) and (b) in each figure)
Shows a state where the voltage becomes a relatively low voltage ((c) in each figure), and the right half shows a state where the added voltage (E ₃ + E ₄ ) of the induced voltages E ₃ and E ₄ becomes a relatively high voltage. Show. In order to obtain such a phase relationship, the transformers T ₃ , T
_By connecting ₄ to the inverter device 1, even if the output of the inverter device 1 changes, the period during which the charging / discharging current flows through the capacitor C ₂ can be stabilized so as to substantially correspond to the change in the input voltage. . In other words, even if the current consumption in the inverter device 1 changes, the capacitor C
The excess and deficiency of the charging / discharging current of ₂ is reduced, and the capacitor C ₁
The period during which the charging current flows to the capacitor is optimized, and as a result, the voltage across the capacitor C ₁ can be stabilized. As a result, even if the current consumption of the inverter device 1 changes,
The voltage added to the voltage across capacitor C ₂ (E ₃ + E
₄ ) can be maintained almost constant, and the input power factor can be maintained. The adding means is realized by connecting the secondary windings n ₂₃ and n ₂₄ of the transformers T ₃ and T ₄ in series here.

Needless to say, both transformers T ₃ and T ₄
18, the added voltage (E ₃ + E ₄ ) of the voltages E ₃ and E ₄ induced in the secondary windings n ₂₃ and n ₂₄ is minimum when the phase difference between the voltage waveforms is 180 degrees. And the phase difference is 0
Maximum when degree. Therefore, regarding the voltage phases of the voltages E ₃ and E _{4, when} the phase difference approaches 0 degrees as the output of the inverter device 1 increases, the secondary windings n ₂₃ and n ₂₄ of both transformers T ₃ and T ₄ are connected. Voltages induced in the secondary windings n ₂₃ and n ₂₄ of both transformers T ₃ and T ₄ when the phase difference becomes closer to 0 degree as the output of the inverter device 1 becomes smaller by connecting and adding the same polarity. The voltage phase of one of E ₃ and E ₄ is shifted by 90 degrees and added. If such a configuration is adopted, if the output of the inverter device 1 is reduced so as to preheat the discharge lamp DL which is a load, the voltage (E ₃ + E ₄ ) added to the capacitor C ₂ is also reduced and the capacitor C ₂ is reduced.
The charge / discharge current of ₂ can be made relatively small, and
If the output of the inverter device 1 is increased so that the discharge lamp DL is turned on, the voltage (E ₃ + E ₄ ) rises and the charging / discharging current of the capacitor C ₂ can be increased. In this way, the charge / discharge current of the capacitor C ₂ can be controlled so as to adapt to the output of the inverter device 1, and as a result the input power factor becomes stable. In the present embodiment as well, as in the first embodiment, the transformers T ₃ and T ₄ are not always necessary, and the place where a high frequency is generated in the inverter device 1 may be directly connected to the capacitor C ₂ .

(Embodiment 11) In this embodiment, as shown in FIG. 19, the diode D ₁ in the structure of the embodiment 10 is replaced with an inductor L ₂ . With this configuration,
Since the capacitor C ₂ and the inductor L ₂ form a series resonance circuit, the same operation as in Example 9 can be performed. That is, when using the inverter device 1 that lowers the frequency when the discharge lamp DL is preheated as compared to when it is preheated,
The resonance frequency of the series resonance circuit constituted by the capacitor C ₂ and the inductor L ₂ is set to be lower than the frequency of the high frequency generated in the inverter device 1. Therefore,
At the time of preheating the discharge lamp DL, the frequency of the high frequency generated from the inverter device 1 is high, and the charging / discharging current of the capacitor C ₂ is small.
The frequency of the high frequency output from the inverter device 1 is low, and the charging / discharging current of the capacitor C ₂ is large.
In this way, as in the tenth embodiment, the voltage (E ₃ + E ₄ ) added to the capacitor C ₂ is controlled according to the output of the inverter device 1, and the capacitor is also controlled by the relationship with the resonance frequency of the series resonance circuit. Since the charging / discharging current of C ₂ is controlled, both effects act synergistically, and the fluctuation of the input power factor due to the change of the operating frequency of the inverter device 1 can be significantly improved. Other configurations and operations are similar to those of the tenth embodiment.

(Embodiment 12) In this embodiment, as shown in FIG. 20, an example in which the high frequency wave output from the inverter device 1 is fed back and superimposed on the input side of the rectifier circuit DB is shown. That is, the inductor L ₂ is inserted between the filter circuit NF and the rectifier circuit DB, and the secondary windings n _{23 of} the transformers T ₃ and T ₄ are
It has a configuration in which a series circuit of n ₂₄ and a capacitor C ₂ is connected between the input ends of the rectifier circuit DB. Here, one end of the capacitor C ₂ is connected to the connection point between the inductor L ₂ and the rectifier circuit DB. Even in this configuration, the diode D ₂ can be omitted because it has the same function as that of the eleventh embodiment and the backflow from the capacitor C ₁ is blocked by the rectifier circuit DB. Other configurations and operations are the same as the tenth embodiment.
Is the same as.

(Embodiment 13) In this embodiment, as shown in FIG. 21, the secondary winding n of the transformers T ₃ and T ₄ is provided between the rectifier circuit DB and the diode D ₂ on the output side of the rectifier circuit DB. A series circuit of ₂₃ and n ₂₄ is inserted, a capacitor C ₉ is connected between the output terminals of the rectifier circuit DB, and a capacitor C ₉ is connected to the series circuit of the diode D ₂ and the capacitor C _1.
It has a configuration in which ₁₀ are connected in parallel. In this configuration, the function of the capacitor C _{2 in} each of the above-described embodiments is the same as that of the capacitor C ₉ ,
We share in C ₁₀ .

That is, since the high frequency voltage induced in the secondary windings n ₂₃ and n ₂₄ of the transformers T ₃ and T ₄ is superimposed on the output voltage waveform of the rectifier circuit DB, as in the tenth embodiment, The charging / discharging current of the capacitors C ₉ and C ₁₀ can be controlled at high frequencies. Other configurations and operations are similar to those of the tenth embodiment. (Example 14) This example, as shown in FIG. 22, Example 12 and similarly the inductor L ₂ provided between the filter circuit NF a rectifier circuit DB, a rectifier circuit inductor L ₂ at the input side of the DB A series circuit of secondary windings n ₂₃ and n ₂₄ of the transformers T ₃ and T ₄ between the rectifier circuit DB and the rectifier circuit DB,
Further, the capacitor C ₁₀ is connected between the input terminals of the rectifier circuit DB.

In this configuration, since the high frequency voltage induced in the secondary windings n ₂₃ and n ₂₄ of the transformers T ₃ and T ₄ is superimposed on the voltage waveform of the AC power supply AC, as in the twelfth embodiment, The charging / discharging current of the capacitor C ₁₀ can be controlled at high frequencies. Other configurations and operations are the same as the first embodiment.
Same as 2. (Fifteenth Embodiment) In the tenth to fourteenth embodiments, in the inverter device 1, the high frequency is obtained from two locations where the phase difference changes between the preheating and the lighting of the discharge lamp DL. Then, one of the locations where the high frequency is taken out from the inverter device 1 is a location where the change in the high-frequency voltage amplitude is small between preheating and lighting, and the other is the location where the change in the voltage amplitude is large between preheating and lighting. Now, for one of the high frequencies taken out from the inverter device 1, FIG.
As shown in FIG. 3 (a), the voltage amplitudes during preheating and lighting are substantially equal (the left half of FIG. 23 indicates preheating, the right half indicates lighting), and the other high frequency is shown in FIG. ), The voltage amplitude during preheating is smaller than the voltage amplitude during lighting. Here, it is assumed that the phase difference does not change between preheating and lighting. In this case, by adding the two high frequencies in phase, as shown in FIG. 23 (c), it is possible to obtain a high frequency in which the voltage amplitude is larger when the lamp is lit than when it is preheated. The high frequency thus obtained can be used for controlling the charge / discharge currents of the capacitors C ₂ , C ₉ and C _{10 in} the _{tenth to} fourteenth embodiments.

By the way, as shown in FIG. 24 (a), one high frequency has substantially the same voltage amplitude during preheating and during lighting, and as shown in FIG. 24 (b), the other high frequency is higher than during preheating. When the voltage amplitude becomes smaller during lighting, both high frequencies are added as opposite phases, and the added voltage amplitude during lighting is larger than that during preheating, as shown in FIG. growing. Even if this configuration is adopted, it can function similarly to the configurations of the tenth to fourteenth embodiments. However, there is no change in the high-frequency phase difference between preheating and lighting.

In the tenth to fifteenth embodiments,
Although the case where the high frequency is obtained from two places of the inverter device 1 has been described, the high frequencies may be obtained from more places and added. In this case as well, it goes without saying that the added voltage must be set so that the voltage amplitude is small when the output energy of the inverter device 1 is small and the voltage amplitude is large when the output energy is large. Further, the state in which the output energy of the inverter device 1 changes is described with the preheating and the lighting. However, when the discharge lamp DL is dimmed, the voltage amplitude to which a high frequency is added depends on the dimming degree. Therefore, the input power factor can be improved even during dimming control.

(Embodiment 16) In this embodiment, as shown in FIG. 25, the diode D ₁ connected between the rectifier circuit DB and one end of the capacitor C ₂ in the embodiment 10 is omitted. The inverter device 1 having the same configuration as that of the second embodiment is used, and is provided for outputting to the discharge lamp DL.
Winding transformer T ₄ with a n ₃₄ is provided with another secondary winding n ₂₄ (the transformer T ₁ in the second embodiment), the one transistor Q ₂ emitter - 1 of the transformer T ₃ between the collector A series circuit of a secondary winding n ₁₃ and a capacitor C ₇ is connected in parallel. That is, there is the derived is a high frequency voltage from two places of the inverter device 1, connected in series and the secondary winding n ₂₄ of the secondary winding n ₂₃ and the transformer T ₄ of the transformer T _3, the series The circuit is connected in series with the capacitor C ₂ . Further, the control circuit 3 controls the ratio of the ON period and the OFF period of the transistors Q ₁ and Q ₂ to adjust the energy supplied to the discharge lamp DL, thereby controlling the preheating and lighting.

[0065] In this configuration, the rectangular wave voltage waveform across the transistor Q ₂ to which having substantially equal amplitude voltage across the capacitor C ₁ for smoothing with the switching occurs, 2 of the transformer T ₃ Tsugimaki Induced voltage E on line n _23
As shown in FIGS. 26B and 27B, ₃ has a substantially constant voltage amplitude. Further, the secondary winding n ₃₄ of the transformer T ₄ connected to the discharge lamp DL
The voltage E ₄ induced in the next winding n ₂₄ changes according to the output energy to the discharge lamp DL. Here, the voltage E ₄ induced in the secondary winding n ₂₄ is shown in FIG.
As shown in (a), if it is set to be lower during lighting than during preheating, it corresponds to the case described with reference to FIG. 24 in the fifteenth embodiment. Therefore,
By connecting the secondary windings n ₂₃ and n ₂₄ of the transformers T ₃ and T ₄ so that the voltage waveforms of the voltages E ₃ and E ₄ are almost in opposite phases, the added voltage of the voltages E ₃ and E ₄ is added. (E ₃ +
The E _4), can also be made to be higher towards the lit than during preheating of the discharge lamp DL, the voltage (E ₃ + E ₄₎
Is added to the voltage across capacitor C ₂ ,
The input power factor can be improved.

Further, the voltage E ₄ induced in the secondary winding n ₂₄
As shown in FIG. 27 (a), if it is set to be higher during lighting than during preheating, it corresponds to the case described in the fifteenth embodiment with reference to FIG. 23. Therefore,
By connecting the secondary windings n ₂₃ and n ₂₄ of the transformers T ₃ and T ₄ so that the voltage waveforms of the voltages E ₃ and E ₄ are almost in phase, the added voltage of the voltages E ₃ and E ₄ ( E ₃ + E
₄ ) can be made higher when the discharge lamp DL is preheated than when it is preheated. By adding this voltage (E ₃ + E ₄ ) to the voltage across the capacitor C ₂ , the input power factor can be increased. It can be stabilized. The other structure is similar to that of the tenth embodiment.

(Embodiment 17) In this embodiment, as shown in FIG. 28, in the inverter device 1 having the same configuration as that of Embodiment 16, the transformer T ₁ for obtaining the output to the discharge lamp DL is used alone. , Transformer T instead of inductor L _1
4 is connected to the primary winding n _14, and the primary winding n ₁₃ of the transformer T ₃ is connected between the emitter and collector of the transistor Q _1.
And a capacitor C ₇ in a series circuit connected in parallel. Further, the present embodiment has a configuration corresponding to Example 12, the series circuit of the secondary winding n _23, n ₂₄ and the capacitor C ₂ of the transformer T _3, T ₄ between the input ends of the rectifier circuit DB It is connected. Other configurations and operations are similar to those of the sixteenth embodiment.

(Embodiment 18) As shown in FIG. 29, this embodiment is based on the construction of Embodiment 16 except that the transformer T ₄ is omitted. That is, in the inverter device 1, the capacitor C ₅ , the inductor L _1, and the discharge lamp DL are connected in series, and this series circuit is connected to the transistor Q.
_{Since the two} emitters and collectors are connected in parallel, a capacitor C ₆ is connected in parallel to the discharge lamp DL.
Also, one end of the secondary winding n ₂₃ of the transformer T ₃ is connected to one end of the capacitor C ₆ via the capacitor C _2, and the other end of the secondary winding n ₂₃ is connected to the anode of the diode D _2. Have Here, similarly to the tenth embodiment, the diode D ₁ is inserted between the anode of the diode D ₂ and the output end of the rectifier circuit DB.

In the above-described inverter device 1, when the transistor Q ₁ is turned on, the transistor Q ₁ -capacitor C ₅ -inductor L ₁ -discharge lamp DL and capacitor C ₆
When a current flows through a path called a parallel circuit of the transistor Q ₂ and the transistor Q ₂ is turned on, a current flows through a path called a parallel circuit of the capacitor C ₅ -transistor Q ₂ -discharge lamp DL and the capacitor C ₆ . Therefore, an alternating current flows through the discharge lamp DL, and by setting the operating frequencies of the transistors Q ₁ and Q ₂ to be high frequency, it is possible to supply high frequency power to the discharge lamp DL. However, it is possible to provide a high frequency to one end of a capacitor C _2, it is possible to add the voltage induced in the secondary winding n ₂₃ of the high-frequency voltage and the transformer T ₃ across capacitor C _2. Other configurations and operations are similar to those of the sixteenth embodiment.

The configurations of the sixteenth to eighteenth embodiments can be applied not only to the control of preheating and lighting but also to the dimming of the discharge lamp DL. That is, in the discharge lamp DL, the lamp voltage generally rises as the lamp current decreases as shown in FIG. 30, so that the discharge lamp DL is obtained from two locations in the inverter device 1 as described with reference to FIG. 24 in the fifteenth embodiment. It suffices to add the high frequencies so that they have opposite phases. With this configuration, even if dimming control is performed, the input power factor is stabilized regardless of the dimming degree. Needless to say, the configuration of the inverter device 1 is not limited to the one described above.

(Embodiment 19) As shown in FIG. 31, this embodiment is the same as that of the embodiment 16 except that transformers T ₃ , T
_The number of secondary windings n ₂₃ , n ₄₃ , n ₂₄ , two on each output side of the inverter device 1,
Two transformers T ₃ and T ₄ having n ₄₄ are provided, and a series resonance circuit of a capacitor C ₄ and an inductor L ₄ is provided between the two transformers T ₃ and T ₄ . The series circuit of the capacitor C ₄ and the inductor L ₄ is connected across the secondary winding n ₄₃ of the transformer T ₃ , and the capacitor C ₄ is connected across the primary winding n ₁₄ of the transformer T _4. There is. The primary winding n ₁₃ of the transformer T ₃ is connected in parallel with the capacitor C ₆ , and the discharge lamp DL is connected to the secondary winding n ₄₄ of the transformer T ₄ .

According to the above configuration, the high frequency output from the inverter device 1 is applied to the transformer T ₃ and the series resonant circuit including the capacitor C ₄ and the inductor L ₄ and the transformer T 3.
It will be supplied to the discharge lamp DL through ₄ . Here, by the series resonant circuit between the two transformer T _3, T ₄ is inserted, at the time of preheating both the transformer T _3, T
₄ of the primary winding n _13, the voltage phase of n ₁₄ is approximately equal, one at the time of lighting both transformer T _3, T ₄ winding n _13,
The voltage phase of n ₁₄ is shifted. Therefore, FIG.
As already explained with reference to No. 7, both transformers T ₃ , T
The secondary windings n ₂₃ and n ₂₄ of ₄ are connected in series with a polarity such that the added voltage is larger when lighting than when preheating, and this series circuit is connected in series to the capacitor C ₂ to obtain the input power factor. Can be stabilized. The other structure is the same as that of the sixteenth embodiment.

(Embodiment 20) In this embodiment, as shown in FIG. 32, the discharge lamps DL ₁ and DL ₁ of the construction of Embodiment 10 are used.
₂ is a plurality of lights (two lights in the embodiment). In such a configuration, when two lights are normally turned on,
The output energy of the inverter device 1 changes when only one lamp is normally turned on by turning off the lamp or removing one lamp. Therefore, in this embodiment, an example is shown in which the input power factor does not change significantly even if the number of lighting of the discharge lamps DL ₁ and DL ₂ changes.

The basic idea of this embodiment is the same as that of the tenth embodiment, but in the tenth embodiment, the input power factor is stabilized against changes between preheating and lighting and changes during dimming control. In contrast to this, the discharge lamp DL is used in this embodiment.
_The objective is to stabilize the input power factor against changes in the number of lighting of ₁ and DL ₂ . Therefore, the discharge lamps DL ₁ , DL
_The high frequency is extracted from the portion where the high frequency voltage amplitude changes and the portion where the high frequency voltage does not change when the number of lighting of ₂ changes, and the high frequency voltage obtained by adding both is added to the voltage across the capacitor C ₂ .

[0075] Now, the voltage E ₄ induced in the secondary winding n ₂₄ of the transformer T ₄ in FIG. 32, the discharge lamp DL _1, DL
If the higher the number of lights of _{2 is,} the higher the number of lights is, the phases of the voltage waveforms of the induced voltages of the secondary windings n ₂₃ and n ₂₄ of both transformers T ₃ and T ₄ are added in the same phase, and conversely the discharge lamp DL ₁ , The secondary windings n ₂₃ and n ₂₄ are connected so that if the number of lighting of DL ₂ decreases as the number of lighting increases, the phases are added in reverse phase. By employing such a connection relationship, the transformer T _3, T ₄ of the secondary winding n _23, the induced voltage of the n ₂₄ E _3, E ₄ voltage obtained by adding the (E ₃
+ E ₄ ) increases as the number of discharge lamps DL ₁ and DL ₂ turned on increases. On the other hand, the output of the inverter device 1 is the discharge lamp DL.
_{Since the} larger the number of lights of ₁ and DL _{2 is,} the larger the number of lights is, so that the charging / discharging current to the capacitor C ₂ can be adjusted according to the output of the inverter device 1, and the discharge lamps DL ₁ and DL 2 can be adjusted.
It is possible to stabilize the input power factor regardless of the number of lighting of ₂ .

In this embodiment, the discharge lamps DL ₁ , DL
_The secondary winding n ₂₄ of one transformer T ₄ according to the number of lighting of 2
The case has been described in which the high-frequency voltage amplitude induced in the transformer changes, but the phase of the induced voltage in the secondary winding n ₂₃ of the transformer T ₃ changes in response to the change in the number of lighting of the discharge lamps DL ₁ and DL _2. Instead, the technical idea of this embodiment can be applied even if the phase of the induced voltage to the secondary winding n ₂₄ of the transformer T ₄ changes. In addition, the discharge lamps DL ₁ , DL
₂ lights number is not limited to two-lamp, it may be a further multi-lamp.

(Embodiment 21) This embodiment shows a specific construction of Embodiment 20. As shown in FIG. 33, a different transformer is provided from the inverter device 1 to each of the discharge lamps DL ₁ and DL ₂ . The electric power is supplied through T ₃ and T ₄ . The inverter device 1 has two transistors Q.
The emitter and collector of ₁ and Q ₂ are connected in series, and this series circuit is connected between both ends of the capacitor C ₁ .
Two inductors L ₁₁ and L ₁₂ have transformers T ₃ and
The primary windings n ₁₃ and n ₁₄ of T ₄ are connected in series, each series circuit is connected in parallel, and a capacitor C _{5 is} connected to this parallel circuit.
Is connected in series between the emitter and collector of one transistor Q ₂ . Further, diodes D ₅ and D ₆ are connected in antiparallel between the emitters and collectors of the transistors Q ₁ and Q ₂ , respectively. Transformer T _3, T ₄ has a secondary winding _{n 23, n 33, n 24} , n 34 of two each, the discharge lamp DL ₁ to the secondary winding n _33, n ₃₄ each one, DL ₂ is connected. Each of the discharge lamps DL ₁ and DL ₂ is of a preheating type having a filament, and the secondary windings n ₃₃ and n _{34 of} the transformers T ₃ and T ₄ are connected to the power source side of each filament and are connected to the non-power source side of the filament. Are connected to capacitors C ₆₁ and C ₆₂ , respectively. The other secondary windings n ₂₃ and n _{24 of the} transformers T ₃ and T ₄ are connected in series, and this series circuit is connected in series to the capacitor C ₂ . Each transistor Q of the inverter device 1
_The control circuit 3 controls ₁ and Q ₂ to be turned on and off alternately.

In the above configuration, a series resonance circuit is formed between the inductors L ₁₁ and L ₁₂ and the corresponding capacitors C ₆₁ and C ₆₂ via the transformers T ₃ and T ₄ , respectively, and the discharge lamp D
A sinusoidal high frequency voltage is applied to L ₁ and DL ₂ . That is, if the discharge lamps DL ₁ and DL ₂ are connected, the secondary windings n _{23 of} the transformers T ₃ and T ₄ ,
A sinusoidal high frequency voltage is also induced in n ₂₄ . When the discharge lamps DL ₁ and DL ₂ are removed, the secondary windings n ₂₃ and n ₂₄ of the corresponding transformers T ₃ and T _{4 have} a voltage between the emitter and collector of the transistors Q ₁ and Q ₂ , respectively. A similar rectangular wave voltage waveform is generated. Therefore, the connection polarities of the secondary windings n ₂₃ and n ₂₄ are set according to the change in voltage amplitude depending on the attachment / detachment of the discharge lamps DL ₁ and DL ₂ , and the secondary windings n ₂₃ and n ₂₄ are connected in series. The voltage across the circuit is the discharge lamp DL ₁ , D
The input power factor can be made almost constant regardless of the number of lighting of the discharge lamps DL ₁ and DL ₂ by setting the L ₂ to be higher than the one lighting when lighting two lights. Other configurations and operations are similar to those of the twentieth embodiment.

(Embodiment 22) In this embodiment, as shown in FIG. 34, in the configuration of embodiment 21, resonance capacitors C ₆₁ and C ₆₂ are connected to primary windings n _{13 of} transformers T ₃ and T _4. , N
It is connected in parallel to ₁₄ . In this configuration, setting the connection polarity of the secondary winding n _23, n ₂₄ in response to the discharge lamp DL _1, the change of the voltage amplitude of the magnitude corresponding to the attachment and detachment of the DL _2, 2
The voltage across the series circuit of the secondary windings n ₂₃ and n ₂₄ is the discharge lamp D.
When the two lamps of L ₁ and DL ₂ are lit higher than the one lamp is lit, the input power factor can be substantially constant regardless of the number of lighting of the discharge lamps DL ₁ and DL _2. . Other configurations and operations are similar to those of the twenty-first embodiment.

[0080] (Example 23) This example, as shown in FIG. 35, the configuration of the embodiment 22, the transistor Q ₂
The primary winding n of the transformer T ₅ between the emitter and collector of
_{A configuration in} which a series circuit of ₁₅ and a capacitor C ₇ is connected in parallel, and the secondary winding n ₂₅ of the transformer T ₅ is connected in series to the capacitor C ₂ together with the secondary windings n ₂₃ and n ₂₄ of the transformers T ₃ and T _4. have. In this configuration, the voltage applied to the primary winding n ₁₅ of the transformer T ₅ is coercive because becomes rectangular wave having an amplitude of approximately the voltage across the capacitor C _1, a substantially constant voltage across the capacitor C ₁ Due to the slack, a high frequency wave having a substantially constant voltage amplitude is induced in the secondary winding n ₂₅ of the transformer T ₅ regardless of the number of lighting of the discharge lamps DL ₁ and DL ₂ . As a result, the operation is the same as that of the twenty-second embodiment, and the other configuration is the same as that of the twenty-first embodiment.

(Embodiment 24) In this embodiment, as shown in FIG. 36, the embodiment 1 and the embodiment 10 are combined. That is, the path between the rectifier circuit DB and the smoothing capacitor C ₁ is branched, and a pair of diodes D _{1 to} D ₄ is inserted in each branch path. In addition, a plurality of (here, two) transformers T are provided from the inverter device 1.
₆ and T ₇ are used to extract high frequency, and a plurality (here, 2) of each transformer T ₆ and T ₇ are provided.
Secondary windings n ₂₆ , n ₃₆ , n ₂₇ , n ₃₇ are provided. Secondary windings n ₂₆ , n ₃₆ , n ₂₇ , n of transformers T ₆ , T _{7 which} are mutually different
₃₇ is the counterpart (ie, n ₂₆ and n ₂₇ , n ₃₆
And n ₃₇ ) are connected in series, and capacitors C ₂ and C ₃ are connected in series to each series circuit. Secondary winding n ₂₆ ,
a series circuit of n _27, n _36, n ₃₇ and a capacitor C _2, C _3, the one end of each branch of the diode _{D 1, D 2, D 3} , D 4 capacitor C ₂ to the connection point between, C ₃ The other end is connected to the negative electrode of the rectifier circuit DB. put it here,
The waveform of the voltage across the series circuit of the secondary windings n ₂₆ and n ₂₇ and the waveform of the voltage across the series circuit of the secondary windings n ₃₆ and n ₃₇ are almost at the connection point to the capacitors C ₂ and C ₃ . The connection polarities of the secondary windings n ₂₆ , n ₂₇ , n ₃₆ , and n ₃₇ are set so as to have opposite phases. In addition, the secondary windings n ₂₆ , n ₂₇ , connected in series,
n ₃₆ and n ₃₇ are 2 when the output of the inverter device 1 is large.
The high-frequency extraction point from the inverter device 1 is set so that the voltage across the series circuit of the secondary windings n ₂₆ , n ₂₇ , n ₃₆ , and n ₃₇ rises. Such extraction points have been described in the tenth to nineteenth embodiments.

By adopting the above configuration, when the output of the inverter device 1 changes between preheating and lighting or changes due to dimming control, the charging / discharging currents of the capacitors C ₂ and C ₃ change. The input power factor can be maintained substantially constant regardless of the output of the inverter device 1, and the input power factor can be maintained in a good state. Moreover, since the charging and discharging of the capacitors C ₂ and C ₃ are reversed, the charging currents (that is, input currents) and discharging currents (charging currents to the smoothing capacitor C ₁ ) of the capacitors C ₂ and C ₃ are reversed. The peak value of can be reduced. This is
This leads to downsizing of the filter circuit NF, suppression of heat generation of the capacitor C ₁ and reduction of capacitance. After all, Example 1
It is possible to achieve the effects of the above and Example 10 at the same time. Other configurations are similar to those of the first embodiment. Here, the number of branch paths from the rectifier circuit DB to the capacitor C ₁ and the number of locations where high frequencies are taken out are not particularly limited and need not be the same.

(Embodiment 25) This embodiment is a specific example of Embodiment 24 as shown in FIG. 37. In the inverter device 1 shown in Embodiment 2, the transformer T ₆ and the capacitor C ₇ are connected to each other. A series circuit of the primary winding n ₁₆ of the transformer T ₆ and the capacitor C ₇ is provided to form an emitter-transistor of the transistor Q ₂ .
Connected in parallel between collectors, and a transformer T for output
_It has a configuration in which ₁ is replaced with a transformer T ₇ . Then, the corresponding secondary winding n in the transformers T ₆ and T _7
26 , n ₂₇ , n ₃₆ and n ₃₇ are connected in series, and capacitors C ₂ and C ₃ are connected in series to the series circuit of the secondary windings n ₂₆ , n ₃₆ , n ₂₇ and n ₃₇ . Here, the capacitor C _2, C ₃ of 2 as charging and discharging are reversed winding n _26,
The connection polarities of n ₂₇ , n ₃₆ , and n ₃₇ are set.

This configuration is similar to the sixteenth embodiment when the connection forms of the capacitors C ₂ , C ₃ and the secondary windings n ₂₆ , n ₂₇ , n ₃₆ , n ₃₇ of the transformers T ₆ , T ₇ are individually observed. The input power factor can be kept stable regardless of the output of the inverter device 1, as in the sixteenth embodiment. Further, as in the second embodiment, the peak value of the input current and the charging current of the capacitor C ₁ can be reduced.

Regarding the place where the high frequency is taken out from the inverter device 1 and the place where the high frequency is fed back and superimposed,
The configuration is not limited to the above, but the transformers T ₆ , T
It is also possible to adopt a form that does not use ₇ and to adopt various forms as to the presence or absence of the resonance circuit as shown in each of the above embodiments. Moreover, the configuration of the inverter device 1 is not particularly limited.

(Embodiment 26) In this embodiment, as shown in FIG. 38, two rectifier circuits DB ₁ and DB ₂ are provided, and a high frequency is fed back and superposed on the output side of each rectifier circuit DB ₁ .
The rectifier circuit DB ₂ has a high frequency feedback superimposed on the input side. More specifically, two transformers T ₆ and T ₇ for extracting high frequencies are connected to the inverter device 1, and each of the transformers T ₆ and T ₇ has two secondary windings n ₂₆ and n ₃₆ . n ₂₇ and n ₃₇ are provided, and the corresponding secondary windings n ₂₆ , n ₂₇ , n ₃₆ and n ₃₇ are connected in series. The series circuits of the secondary windings n ₂₆ , n ₂₇ , n ₃₆ , and n ₃₇ are connected in series with the capacitors C ₂ and C ₃ , respectively. An inductor L ₂ is provided between the rectifier circuit DB ₁ and the capacitor C _1.
And a diode D _{2 connected} in series, and an inductor L ₃ is provided between the filter circuit NF and the rectifier circuit DB _2.
Is inserted. The series circuit of the secondary windings n ₂₆ and n ₂₇ and the capacitor C ₂ is connected in parallel to the series circuit of the diode D ₂ and the capacitor C ₁ and is connected to the secondary windings n ₃₆ and n ₃₇ and the capacitor C ₃ . The series circuit is connected between the input terminals of the rectifier circuit DB ₂ . Here, the series circuit of the secondary windings n ₂₆ and n _{27 and} the series circuit of the secondary windings n ₃₆ and n ₃₇ are connected so that the voltage waveforms of the voltages at both ends of the series circuits have opposite phases. . Further, in both series circuits, the location and connection relationship for extracting the high frequency are set so that the voltage across the inverter device 1 becomes higher as the output of the inverter device 1 becomes larger.

Also in this configuration, since the charging / discharging currents of the capacitors C ₂ and C ₃ are adjusted according to the output of the inverter device 1, the output of the inverter device 1 can be kept stable and the inverter device 1 can Since the generated high frequencies are applied to the capacitors C ₂ and C ₃ so as to have opposite phases to each other, the peak values of the input current and the charging current of the capacitor C ₁ can be reduced.

[0088] In Examples 24 and superimposes a high frequency on the output side of the Example 25, the rectifier circuit DB, superimposing each frequency on the output side of Example 26 in the rectifier circuit DB ₁ and the input side of the rectifier circuit DB ₂ Example 7
Similarly to the above, two rectifier circuits may be provided, high frequencies may be superimposed on the input side of each rectifier circuit, and the outputs of the rectifier circuits may be merged to charge the capacitor C ₁ .

(Embodiment 27) This embodiment is shown in FIG.
As described above, a plurality of (here, two) discharge lamps DL₁, DL
₂Is an example in the case of using
have. However, each transformer T₆, T₇Carried out
Similar to Example 20, the discharge lamp DL₁, DL₂Depending on the number of lights
Secondary winding n₂₆, N₂₇, N₃₆, N₃₇Both of the series circuit of each other
It is connected to the point where the terminal voltage changes. That is, lighting
The larger the number, the more the secondary winding n₂₆, N₂₇, N ₃₆, N₃₇Direct of each other
Take out high frequency so that the voltage across the column circuit becomes large
The location is set. The other structure is the same as that of the twenty-fourth embodiment.
Therefore, in this embodiment, the input current and the capacitor C₁of
As the peak value of the charging current becomes smaller, the discharge lamp D
L₁, DL₂Suppress changes in input power factor due to number of lights
be able to.

(Embodiment 28) This embodiment is a concrete example of the construction of the embodiment 27. As shown in FIG. 40, the basic construction is the same as that of the embodiment 21. ₁ , output transformer T for feeding DL _2
3 and T ₄ are provided, and transformers T ₃ and T ₄ each having two secondary windings n ₂₃ , n ₃₃ , n ₂₄ and n ₃₄ are used. Further, as described in Example 24, the corresponding secondary winding n ₂₃ of the transformer T _3, T _4, n _24,
n _33, with n ₃₄ are connected to each other in series, each series circuit is respectively connected in series with the capacitor C _2, C _3. The series circuit of the secondary windings n ₂₃ and n ₂₄ and the capacitor C ₂ is connected in parallel to the series circuit of the diode D ₂ and the capacitor C ₁ and is connected to the secondary windings n ₃₃ and n ₃₄ and the capacitor C ₃ . The series circuit is connected in parallel with the series circuit of the diode D ₄ and the capacitor C ₁ . The configuration of the inverter device 1 is the same as that of the twenty-first embodiment, and the other configurations and operations are the same as those of the twenty-seventh embodiment.

(Embodiment 29) This embodiment is a concrete example of the structure of the embodiment 27. As shown in FIG. 41, the basic structure is the same as that of the embodiment 22, and the embodiment 28 is for resonance. Only the positions of the capacitors C ₆₁ and C ₆₂ are different. That is, Example capacitor C ₆₁ for resonance at 28, C ₆₂ the discharge lamps DL ₁ to, DL whereas was connected to the non-power supply side of the _second filament, in this embodiment the transformer T _3, T
_The only difference is that they are connected in parallel to the _four primary windings n ₁₃ and n ₁₄ . Other configurations and operations are similar to those of the twenty-eighth embodiment.

[0092]

According to the present invention, a high frequency feedback means part including a plurality of paths for feeding back superimposed on the input side of the smoothing capacitor of the high frequency voltage generated by the inverter device provided, the high-frequency feedback
The means is a condenser for feedback superposition inserted on the path.
And the smoothing capacitor charging path from the AC power supply.
From the input smoothing capacitor to the feedback superimposing capacitor
Since it has a diode that blocks reverse current, the peak value of the input current is reduced by making the high-frequency voltage of each path different from each other, and the charging current to the smoothing capacitor is adjusted according to the output of the inverter device. You can

The high frequency voltage is taken out from the inverter device, and the distribution means distributes the high frequency voltage so as to feed back and superimpose the high frequency voltage on a plurality of points on the input side of the smoothing capacitor, and the voltage waveform of the high frequency voltage distributed by the distribution means. With a phase shifter that makes the phases different from each other, if there is only one path on which the high frequency voltage is superimposed, the input current from the AC power supply and the charging current to the smoothing capacitor are synchronized with one cycle of the high frequency current. However, since multiple paths are used and the phases are shifted from each other, the input current from the AC power supply and the charging current to the smoothing capacitor are compared during one cycle of high frequency current. For a long time,
The peak value of these currents can be reduced. This leads to the effect that it becomes easy to take measures to suppress the mixing of noise into the AC power supply (specifically, the capacity of the filter circuit can be reduced), and the capacity of the smoothing capacitor can also be reduced. .

The high frequency voltage is taken out from a plurality of points of the inverter device, and the adding means for adding the high frequency voltage appropriately and feedback-superimposing it on the input side of the smoothing capacitor is provided. The high frequency voltage is taken out by the adding means. If the voltage amplitude of the voltage is set at a position that changes according to the output of the inverter device, the charging current to the smoothing capacitor changes according to the change of the output of the inverter device. If the charge of the smoothing capacitor becomes insufficient due to an increase in the voltage, the high frequency voltage will be increased to increase the charging current to the smoothing capacitor, and the input power factor will be kept almost constant regardless of the output of the inverter device. There is an effect that can be.

[Brief description of drawings]

FIG. 1 is a schematic configuration diagram corresponding to the invention of claim 3.

FIG. 2 is a schematic configuration diagram corresponding to the invention of claim 5.

FIG. 3 is a schematic configuration diagram when the invention of claim 3 and the invention of claim 5 are combined.

FIG. 4 is a circuit diagram showing a first embodiment.

FIG. 5 is an operation explanatory view showing the first embodiment.

FIG. 6 is a circuit diagram showing a second embodiment.

FIG. 7 is a circuit diagram showing a third embodiment.

FIG. 8 is a circuit diagram showing a fourth embodiment.

FIG. 9 is a circuit diagram showing a fifth embodiment.

FIG. 10 is a circuit diagram showing a sixth embodiment.

FIG. 11 is a circuit diagram showing a seventh embodiment.

FIG. 12 is a circuit diagram showing an eighth embodiment.

FIG. 13 is a circuit diagram showing a ninth embodiment.

FIG. 14 is an explanatory diagram of the operation of the ninth embodiment.

FIG. 15 is a circuit diagram showing a tenth embodiment.

FIG. 16 is an operation explanatory diagram of the tenth embodiment.

FIG. 17 is an operation explanatory diagram of the tenth embodiment.

FIG. 18 is an operation explanatory diagram of the tenth embodiment.

FIG. 19 is a circuit diagram showing an eleventh embodiment.

FIG. 20 is a circuit diagram showing a twelfth embodiment.

FIG. 21 is a circuit diagram showing a thirteenth embodiment.

FIG. 22 is a circuit diagram showing a fourteenth embodiment.

FIG. 23 is an operation explanatory view of the fifteenth embodiment.

FIG. 24 is an operation explanatory view of the fifteenth embodiment.

FIG. 25 is a circuit diagram showing an example 16;

FIG. 26 is an explanatory diagram of the operation of the sixteenth embodiment.

FIG. 27 is an explanatory diagram of the operation of the sixteenth embodiment.

FIG. 28 is a circuit diagram showing an example 17;

FIG. 29 is a circuit diagram showing an eighteenth embodiment.

FIG. 30 is an explanatory diagram of the operation of the eighteenth embodiment.

FIG. 31 is a circuit diagram showing an example 19;

FIG. 32 is a circuit diagram showing an twentieth embodiment.

FIG. 33 is a circuit diagram showing an example 21.

FIG. 34 is a circuit diagram showing an example 22.

FIG. 35 is a circuit diagram showing an example 23.

FIG. 36 is a circuit diagram showing an example 24.

FIG. 37 is a circuit diagram showing an example 25.

FIG. 38 is a circuit diagram showing an example 26.

FIG. 39 is a circuit diagram showing an example 27.

FIG. 40 is a circuit diagram showing an example 28.

FIG. 41 is a circuit diagram showing an example 29.

FIG. 42 is a circuit diagram showing a conventional example.

FIG. 43 is a schematic configuration diagram showing another conventional example.

FIG. 44 is a circuit diagram showing a specific configuration of the conventional example shown in FIG. 43.

45 is an operation explanatory diagram of the conventional example shown in FIG. 44;

FIG. 46 is an operation explanatory view of the conventional example shown in FIG. 44.

FIG. 47 is an operation explanatory view of the conventional example shown in FIG. 44.

48 is an operation explanatory diagram of the conventional example shown in FIG. 44;

FIG. 49 is an operation explanatory view of the conventional example shown in FIG. 44.

[Explanation of symbols]

1 Inverter device 2 High frequency feedback circuit AC AC power supply B ₁ , ..., Bn Path C ₁ Capacitor C ₅ Capacitor C ₆ Capacitor C ₆₁ Capacitor C ₆₂ Capacitor D ₂ Diode D ₄ Diode DB Rectifier circuit DB ₁ Rectifier circuit DB ₂ Rectifier circuit DL load F ₁ , ..., Fn path L ₁ inductor L ₁₁ inductor L ₁₂ inductor T ₁ transformer T ₃ transformer T ₄ transformer T ₆ transformer T ₇ transformer

─────────────────────────────────────────────────── ─── Continuation of the front page (72) Inventor Kenichi Sachi 1048, Kadoma, Kadoma City, Osaka Prefecture Matsushita Electric Works, Ltd. (56) References JP-A 5-161366 (JP, A) JP-A 5-219756 ( (58) Fields investigated (Int.Cl. ⁷ , DB name) H02M 7/48 H05B 41/282

Claims (13)

(57) [Claims] 1. A power supply device comprising a rectifier circuit for rectifying an AC power supply, a smoothing capacitor for smoothing an output voltage of the rectifier circuit, and an inverter device for outputting high-frequency power to a load using a voltage across the smoothing capacitor as a power supply. In the above, a high-frequency feedback means is provided which has a plurality of paths for feedback-superimposing a part of the high-frequency voltage coupled to the load on the input side of the inverter device , and the high-frequency feedback means includes the feedback weights inserted on the path.
Tatami capacitor and smoothing capacitor from AC power supply
Inserted on the charging path and used for feedback superposition from the smoothing capacitor.
A power supply device comprising: a diode that blocks a reverse current to a capacitor . 2. The power supply device according to claim 1, wherein the high-frequency feedback means takes out the high-frequency voltage coupled to the load from one location, and feeds back and superimposes it at a plurality of locations on the input side of the inverter device. 3. The high-frequency feedback means extracts the high-frequency voltage from the inverter device, and distributes the high-frequency voltage so that the high-frequency voltage is feedback-superimposed on a plurality of locations on the input side of the smoothing capacitor, and the distribution means distributes the high-frequency voltage. 2. The power supply device according to claim 1, further comprising a phase shifter that makes the phases of the voltage waveforms of the high-frequency voltage different from each other. 4. The power supply device according to claim 1, wherein the high-frequency feedback means takes out the high-frequency voltage coupled to the load from a plurality of locations and feeds back and superimposes the high-frequency voltage at one location on the input side of the inverter device. 5. The high-frequency feedback means includes adding means for extracting high-frequency voltages from a plurality of locations of the inverter device, adding the high-frequency voltages as appropriate, and performing feedback superimposition on the input side of the smoothing capacitor. 2. The power supply device according to claim 1, wherein the voltage amplitude of the high-frequency voltage added by the adding means is set at a position where it changes according to the output of the inverter device. 6. The power supply device according to claim 2, wherein the high frequency feedback means feedback-superimposes a high frequency voltage between the rectifier circuit and the smoothing capacitor. 7. The power supply device according to claim 2, wherein the high frequency feedback means feedback-superimposes a high frequency voltage between the AC power supply and the rectifier circuit. 8. The high-frequency feedback means feedback-superimposes a high-frequency voltage on each location between the AC power supply and the rectifier circuit and between the rectifier circuit and the smoothing capacitor. Item 3. The power supply device according to item 3. 9. claims, characterized in that said diode is inserted in Junkata direction between the rectifying circuit and a smoothing capacitor, it superimposes a high frequency voltage between the rectifier circuit and the diode 6
The power supply described. 10. The power supply device according to claim 1, wherein the inverter device includes a resonance circuit including an inductor and a capacitor, and the high frequency feedback means extracts a voltage across the capacitor of the resonance circuit as a high frequency voltage. 11. The power supply device according to claim 1, wherein the inverter device includes a resonance circuit including an inductor and a capacitor, and the high frequency feedback means extracts a voltage across the inductor of the resonance circuit as a high frequency voltage. 12. The inverter device comprises a resonant circuit, and the high frequency feedback means comprises a current transformer having a primary winding inserted in a closed circuit through which a resonant current flows, and the high frequency induced in the secondary winding of the current transformer. The power supply device according to claim 1, wherein a voltage is taken out. 13. The power supply device according to claim 1, wherein the inverter device includes a transformer that outputs high-frequency power to a load, and the high-frequency feedback means takes out a high-frequency voltage via the transformer.