JP4635964B2 - Synchronous motor drive device - Google Patents

Description

  The present invention relates to control of a synchronous motor, and more particularly, to a control method for realizing a high-precision motor drive device with high accuracy without using a sensor for detecting the speed and position of the motor.

  The conventional technique for controlling the synchronous motor without detecting the magnetic pole position is a method based on the vector control with the speed / position sensor of the synchronous motor shown in FIG. They are a vector control type sensorless system in which the estimator 14P and the speed estimator 33 are provided, and a method for controlling the synchronous motor in an open loop, called V / F control shown in FIG.

  The vector control type sensorless system shown in FIG. 21 has the same configuration as the vector control system with a sensor, except for the magnetic pole position detection unit and the speed detection unit.

In FIG. 21, reference numeral 1 is a generator of ωr ^* which is a rotational speed command, 2P is a motor control device, 3 is a PWM generator for converting a voltage command into a PWM pulse, 4 is an inverter, 5 is a synchronous motor, and 6 is Current sensor of synchronous motor, 7 is a conversion gain for converting mechanical angular frequency to electrical angular frequency, 9 is a dq coordinate converter for converting three-phase alternating current to a value on rotational coordinates, and 10 generates a d-axis current command Id ^* Id ^* generator, 12 is a voltage command calculator, 13 is a coordinate converter that converts a value on the dq axis to a three-phase AC value, 14P is a magnetic pole position estimator that estimates the magnetic pole axis of the motor, and 16 is a signal 21P is a speed controller that adjusts Iq ^* so that the estimated speed value matches the speed command, and 24 is the current detection values Idc and Iqc, which are the command values Id ^* and Iq, respectively. Corrections are made to the voltage commands Vdc ^* and Vqc ^* so that they match ^* The current controller 33 is a speed estimator for estimating the rotational speed of the electric motor.

  In FIG. 21, the magnetic pole position estimator 14P corresponds to a magnetic pole position sensor, and the speed estimator 33 corresponds to a speed sensor. Further, the speed controller 21P and the current controller 24 are provided in the same manner as the vector controller with the speed / position sensor, and the speed and current are automatically adjusted so as to match each command value. Such a vector control type sensorless system is described as “Permanent Magnet Synchronous Motor Based on Direct Estimation Calculation of Axis Error,” 2000 Annual Meeting of the Institute of Electrical Engineers of Japan, Proceedings [III], No.97, pp.963-966. Is described in “Position Sensorless Control of”. Another sensorless drive technology for synchronous motors is "Embedded magnet type synchronous motor and drive technology: Kaitani, Matsubara, Dekai, Mitsubishi Electric Technical Report, Vol. 73, No. 9, 1999, pp. 68-71" Is disclosed.

In the V / F control, as shown in FIG. 22, the voltage to be applied to the motor is directly determined from the speed command without having an automatic speed / current adjustment unit. In FIG. 22, reference numeral 2Q is a V / F control device, 15 is a zero generator that always makes Vdc ^* zero, and 125 is a power generation coefficient gain corresponding to the power generation coefficient Ke of the motor. In the V / F control, unlike the vector control type sensorless system, since the magnetic pole axis is not estimated, the control configuration is extremely simple. However, if the load suddenly changes during driving, transient vibrations may occur. In order to suppress this, a method of adding a control loop for correcting the speed from the detected current value is disclosed in Japanese Patent Laid-Open No. 2000-236694.

JP 2000-236694 A

  In the case of the vector control type sensorless system, there are automatic adjustment units such as a speed controller and a current controller, and the control performance of the motor can be derived by adjusting these control gains to appropriate values. However, for that purpose, the magnetic pole position estimator and the speed estimator need to function sufficiently in place of the position sensor and the speed sensor. However, actual estimation calculations are affected by variations in the motor constants, delays in calculation processing, and the like, and accuracy equivalent to that of the position / speed sensor cannot be realized, and there is always an estimation error.

  The estimation error of the magnetic pole axis will be described with reference to FIG. If the magnetic pole axis in the motor is defined as the d axis and the axis orthogonal thereto as the q axis, and the estimated coordinate axis in the controller is defined as the dc-qc axis, the axis error Δθ is between them. If the axial error can be made zero by estimating the magnetic pole position, the relationship shown in FIG. 24 is obtained, and ideal vector control can be realized. Here, “ideal” means a state in which the motor current is orthogonal to the motor magnetic flux and all current components contribute as torque.

  However, in reality, since there is an axis error, sufficient performance cannot be obtained even if speed control or current control is performed by the vector control type sensorless method, and it is difficult to adjust the control gains. In addition, when an unstable phenomenon occurs, it is difficult to identify whether the direct cause is the influence of the estimation error or the gain setting of the control unit, and the cause is difficult to investigate. Furthermore, in the case of the vector control type sensorless system, since high-speed arithmetic processing is required to drive the motor at high speed, a low-cost microcomputer with low processing capability cannot cope with it.

  On the other hand, in the case of V / F control, since there is no adjustment part like the vector control type sensorless system, the motor can be controlled at a variable speed without adjustment. However, since the dq axis and the dc-qc axis do not match, high-level control is difficult. In the case of V / F control, a vector diagram of the relationship between voltage and current is shown in FIG. In the V / F control, the voltage axis becomes the qc axis, and the axial error increases as the load increases. For this reason, troubles such as vibration and overcurrent may occur due to disturbance such as load torque fluctuation.

  An object of the present invention is to provide a motor drive device having a performance equivalent to that of a conventional vector control type sensorless system by simplifying the control configuration, reducing the number of adjustment points, and stabilizing the control system.

The synchronous motor driving apparatus according to the present invention calculates the voltage applied to the motor on the coordinate axis (on the dc-qc axis) with respect to the magnetic pole axis, as in the vector control. The synchronous motor drive device of the present invention does not have an automatic adjustment unit such as a speed controller or a current controller, and uses command values such as a rotation speed command and a current command for voltage command calculation. However, since Iq ^* corresponding to the torque current command changes depending on the load state of the motor, it is calculated and given based on the detected current value.

  According to the motor drive system of the present invention, high-performance, high-precision position / speed sensorless drive can be realized without complicating the configuration of the motor control device. As a result, the reliability and stability of the motor drive system are improved.

  Embodiments of the present invention will be described below with reference to the drawings. In each figure, the same reference numeral indicates the same component.

Example 1
FIG. 1 shows a configuration diagram of this embodiment. In FIG. 1, reference numeral 1 is a speed command generator for giving a rotational speed command ωr ^* to the motor, 2 is a controller for calculating the applied voltage of the motor, and 3 is a pulse for driving the inverter 4 based on the voltage command V1 ^*. Generated PWM (pulse width modulation) generator, 4 is an inverter for driving the motor, 5 is a synchronous motor to be controlled (hereinafter abbreviated as “motor”), 6 is a current detector for detecting the current of the motor 5, and 7 is a rotational speed. Conversion gain (P is the number of poles of the motor) for converting the command ωr ^* into the electrical angular frequency command ω1 ^* of the motor, and 8 is an integration for calculating the AC phase θc inside the control device based on the electrical angular frequency command ω1 ^*. , 9 is a dq coordinate converter for converting the current value on the three-phase AC axis into a component on the dc-qc axis, which is the rotational coordinate axis, and 10 is an Id ^* generator for giving a current command Id ^* of the magnetic pole axis component of the motor. , 11 is an Iq ^* generator, 12 is ω1 ^* , I A voltage command calculator for calculating voltage commands Vdc ^* and Vqc ^* on the dc-qc axis based on d ^* and Iq ^* , and 13 is a dc−
A dq inverse converter that converts voltage commands Vdc ^* and Vqc ^* on the qc axis into values on a three-phase AC axis, 41 is a DC power supply unit that constitutes the main circuit power supply of the inverter 4, and 42 is a main circuit unit of the inverter , 43 is a gate driver for generating a gate signal to the main circuit, 411 is a three-phase AC power source for supplying power to the inverter 4, 412 is a diode bridge for rectifying the three-phase AC power source, and 413 is a pulsation in the DC power source. It is a smoothing capacitor that suppresses components.

  FIG. 2 shows a schematic configuration diagram of this embodiment. The present embodiment includes an AC power supply unit 411, a control / inverter unit, and an electric motor 5. As shown in FIG. 2, the control / inverter control board is equipped with a speed command generator 1, a control device 2 for calculating the applied voltage of the motor, and a PWM generator 3, which are based on a microprocessor. It is realized with a digital circuit. Further, the inverter 4, the current detector 6 and the like are also mounted in one device.

Next, the operation of this embodiment will be described with reference to FIG. Based on the speed command ωr ^* , the electrical angular frequency ω1 ^* of the motor is obtained as the output of the conversion gain 7. The phase calculator 8 integrates ω1 ^* to obtain an AC phase θc inside the controller. Based on the AC phase θc, the detected value of the three-phase AC current is coordinate-converted to obtain Idc as the dc axis component and Iqc as the qc axis component. An Iq ^* generator calculates Iq ^* based on Iqc. The Id ^* generator generates a predetermined Id ^* . Here, when the rotor structure of the motor is a non-salient pole type, Id ^* = 0 is given. The voltage command calculator 12 calculates Vdc ^* and Vqc ^* , which are voltages applied to the motor 5, based on the rotational speed ω1 ^* and the current commands Id ^* and Iq ^* . The arithmetic expression is
(Equation 1)
Vdc ^* = R · Id ^* −ω1 · Lq · Iq ^*
Vqc ^* = ω1 ^* · Ld · Id ^* + R · Iq ^* + Ke · ω1 ^* (Equation 1)
Where R: motor resistance, Ld: d-axis inductance,
Lq: q-axis inductance, Ke: power generation constant of the motor. The arithmetic expression of Expression (1) is the same as that used in normal vector control. For example,
"The theory and design of AC servo system: Hidehiko Sugimoto, General Electronic Publishing, p. 78, Formula (4.6)".

The coordinates of Vdc ^* and Vqc ^* obtained by the equation (1) are converted into a voltage command value V1 ^* on the three-phase AC axis. Next, the PWM generator 3 converts the voltage command V1 ^* into a pulse width. In the gate driver, the switching element is driven based on the pulse signal, and voltages corresponding to Vdc ^* and Vqc ^* are applied to the electric motor 5.

  Next, the operation of each unit will be described. The voltage command calculator 12 calculates a voltage command based on the formula (1). Expression (1) is represented in a block diagram as shown in FIG. In FIG. 3, reference numeral 121 is a gain corresponding to the resistance value (R) of the motor, 122 is a gain corresponding to the d-axis inductance (Ld), 123 is a gain corresponding to the q-axis inductance (Lq), and 124 is a multiplier. , 125 are gains corresponding to the power generation constant (Ke).

  As shown in Equation (1) and FIG. 3, the voltage command is calculated using R, Ld, Lq, and Ke that are constants of the motor. If these motor constants are accurate, the motor will operate at the rotational speed and current value as commanded. The vector diagram of the voltage and current at that time is as shown in FIG. 24, and the dq axis of the motor coincides with the dc-qc axis for control.

The command values ω1 ^* and Id ^* given to the voltage command calculator 12 can be given regardless of the load state of the motor, but the command value Iq ^* must be given according to the torque required by the motor. If there is a difference between the command value Iq ^* and the actual torque load, the magnetic pole axis of the motor is displaced from the control axis, causing instability and insufficient torque.

In this embodiment, how to create the command value Iq ^* is important. In the vector control type sensorless system of the prior art shown in FIG. 21, the output of the speed controller is set to Iq ^* . However, the vector control type sensorless system requires a speed estimator and a speed controller, and the control configuration is It becomes complicated. In order to solve these problems, in this embodiment, a current command Iq ^* is created using the current detection value Iqc. Reference numeral 11 in FIG. 1 denotes an Iq ^* generator. Here, Iq ^* is calculated according to the following equation.

ここで、Ｔｒ：時定数、ｓ：ラプラス演算子

Where Tr: time constant, s: Laplace operator

Equation (2) is a first-order lag filter, but a moving average value or the like may be used in addition to this. If Iqc is directly set to Iq ^* , the control system becomes unstable because it operates positively. Therefore, a delay element is provided to suppress the pulsation component, and the control system is stabilized. However, since the fundamental wave component (that is, direct current component) of Iqc and Iq ^* coincide with each other in a steady state, the dc-qc axis and the dq axis finally coincide with each other. Therefore, in the present embodiment, as a result, the relationship shown in the vector diagram of FIG. 24 is obtained, and a stable electric motor drive device in which no shaft error occurs can be realized without requiring adjustment of gain setting or the like. In this embodiment, there is no speed controller. However, due to the inherent characteristics of the synchronous motor, the rotational speed of the motor is controlled to coincide with the command value, and the steady speed deviation becomes zero. As described above, according to the present embodiment, the shaft error in the motor and control can be made zero with a simple control configuration.

(Example 2)
In this embodiment, the control device 2B of FIG. 4 is used instead of the control device 2 of FIG. In FIG. 4, reference numeral 14 is an axis error estimator for estimating and calculating an axis error between the dq axis of the motor and the control axis dc-qc axis, 15 is a zero generator for giving a zero command to the axis error, and 16 is an input signal. An adder 17 for adding (or subtracting) is a magnetic pole axis estimation gain for calculating a correction amount to the electrical angular frequency command ω1 ^* using an axis error.

Next, the operation of this embodiment will be described. The axis error estimator 14 estimates and calculates an error Δθ between the dq axis and the dc-qc axis. Δθ is the dc− observed from the dq axis, as shown in FIG.
This is an error component of the qc axis. The calculation of the estimated value Δθc of Δθ is performed by equation (6).

のようになる。ただし、上式において、Ｌは、Ｌ＝Ｌｄ＝Ｌｑであって、非突極型の電動機を想定している。式（３）は、電動機定数と電動機への印加電圧指令、電流検出値
（ｄｃ−ｑｃ軸上での観測値）に基づいて、直接Δθを推定演算する。式（３）は、非突極型の同期電動機を対象にしているが、突極型の場合にも、同様の演算式で軸誤差が得られることが知られていて、例えば、「電気学会半導体電力変換／産業電力電気応用合同研究会資料、Ｎo.ＳＰＣ−００−６７，軸誤差の直接推定によるＩＰＭモータの位置センサレス制御」に記載がある。

become that way. However, in the above formula, L is L = Ld = Lq, and a non-salient pole type motor is assumed. Expression (3) directly estimates and calculates Δθ based on the motor constant, the applied voltage command to the motor, and the current detection value (observed value on the dc-qc axis). Equation (3) is intended for a non-saliency type synchronous motor, but it is known that a shaft error can be obtained with a similar arithmetic expression even in the case of a salient pole type. As described in "Semiconductor Power Conversion / Industrial Electric Power Application Joint Study Group Material, No. SPC-00-67, IPM Motor Position Sensorless Control by Direct Estimation of Axis Error".

When the axis error Δθc is positive, the control axis dc-qc axis advances from the dq axis from FIG. 23, so that the correction amount Δω1 (in this case, Δω1 <0) so as to reduce the electrical angular frequency ω1 ^*. ) To decrease Δθ. Conversely, when the axial error Δθc is negative, the correction amount Δω1 is added so as to increase the electrical angular frequency ω1 ^* . These operations (PLL operations) are realized by the blocks 14 to 17 in FIG. The magnetic pole axis estimation gain 17 is a coefficient that determines the convergence time of the axis error Δθc. Basically, it may be a proportional gain, but may be a combination of proportional / integral or differential elements.

  According to the present embodiment, even if an axis error occurs due to a load variation or the like, the axis error Δθc can be converged to zero within the set response time. Since the magnetic pole axis estimation gain of this embodiment can be increased in speed, the response performance to torque fluctuations such as load disturbance can be improved.

In this embodiment, the followability to the speed command is also improved. If the axis error Δθc becomes zero at a high response speed by the PLL operation, the output Δω1 of the magnetic pole axis estimation gain 17 also becomes zero. Therefore, the drive frequency of the motor matches the electrical angular frequency ω1 ^* and the speed deviation is short. Will converge to zero, improving the followability to the speed command.

In the present embodiment, the method for directly obtaining the axis error Δθ has been described. However, the method for obtaining the state quantity corresponding to the axis error indirectly can also be applied without any problem. For example, there is no problem even if only the numerator of Expression (3) is obtained by calculation and the electrical angular frequency ω1 ^* is corrected based on the value. In addition, as a method of estimating the magnetic pole position, harmonics may be injected to estimate the magnetic pole position.

(Example 3)
In this embodiment, the control device 2C of FIG. 5 is used instead of the control device 2 of the first embodiment. Reference numeral 18 in FIG. 5 is a torque controller that calculates the q-axis current command Iq ^* based on the estimated value Δθc of the axis error. The difference between the present embodiment and the second embodiment is that the Iq ^* generator 11 is deleted and a torque controller 18 is added.

Next, the operation of this embodiment will be described. In this embodiment, the current command Iq ^* is determined using the estimated value Δθc of the axis error. Since the phase θc of the AC voltage applied to the electric motor is mainly given by the integration of the electrical angular frequency ω1 ^* , when the load suddenly changes, the estimated value Δθc of the shaft error is changed first. Of course, the electrical angular frequency ω1 ^* is corrected via the magnetic pole axis estimation gain 17, but it takes time according to the PLL setting response until it matches the actual speed. Therefore, by quickly determining the current command Iq ^* from the change in the estimated value Δθc of the axis error, torque control with a quick response can be realized. Since the estimated value Δθc of the axis error is constantly zero, the torque controller 18 requires an integral element. Therefore, the torque controller 18 is basically configured based on PI (proportional / integral) control, PID (proportional / integral / differential) control, or the like.

  As described above, in this embodiment, the torque control block 18 is added, and a current command corresponding to the torque change can be obtained quickly, and a motor drive device with higher performance can be realized.

Example 4
In this embodiment, the control device 2D of FIG. 6 is used instead of the control device 2 of the first embodiment. In FIG. 6, reference numeral 19 is an inversion gain for inverting the sign of the signal, 20 is a conversion gain for converting the electrical angular frequency into a mechanical angular frequency, and 21 is a speed controller for controlling the speed of the motor to be constant. The difference between the present embodiment and the third embodiment is that the torque controller 18 is deleted and reference numerals 19 to 21 are added.

Next, the operation of this embodiment will be described with reference to FIG. In the third embodiment, the torque current command Iq ^* is created using the estimated value Δθc of the shaft error. In this embodiment, the torque current command Iq ^* is created using the output Δω1 of the magnetic pole shaft estimated gain 17. Motor drive systems have applications where speed accuracy is important. For example, it is a case where uniform speed of a plurality of electric motors such as a rolling auxiliary machine of a steel rolling system is strongly required. In such applications, speed followability is more important than torque disturbance response.

In this embodiment, in order to quickly converge the speed deviation to zero, the speed controller 21 is provided to create the torque current command Iq ^* . However, if a speed estimator as in the conventional technique (FIG. 21) is added, the control apparatus becomes complicated. In this embodiment, speed control is performed using Δω1 which is the correction amount of ω1 ^* .

The output Δω <b> 1 of the magnetic pole axis estimation gain 17 has a positive value when the actual speed of the motor is higher than the speed command in the control device, and therefore the sign is inverted via the inversion gain 19. This is equivalent to the input (speed deviation) in the conventional speed controller (speed controller 21P in FIG. 21). Next, by the conversion gain 20, Δω1 is divided by the number of pole pairs of the electric motor to convert it into a mechanical angular frequency deviation. Finally, Iq ^* is calculated using the speed controller 21. In the present embodiment, for example, the control (PI control, PID control, etc.) used for speed control of the prior art may be used as it is for the speed controller 21. According to the present embodiment, by configuring the speed controller based on Δω1 that is the correction amount of the AC phase, it is possible to realize an electric motor drive system with good speed followability.

(Example 5)
In this embodiment, a control device 2E shown in FIG. 7 is used instead of the control device 2 of the first embodiment. Reference numeral 18E in FIG. 7 denotes a torque controller that does not include an integral element.

In this embodiment, an Iq ^* generator 11 is added to the fourth embodiment. As described above, since the Iq ^* generator always converges to Iqc = Iq ^* , the steady state characteristic can be compensated. In this embodiment, the Iq ^* generator 11 is combined to improve both steady state and transient characteristics.

Since the constant current command Iq ^* is all contributed by the Iq ^* generator 11, the torque controller 18E of FIG. 7 only needs to function during a transient response. Therefore, these controllers are not provided with an integral element, but may be combined with elements of proportional control or differential (incomplete differentiation) control, and may not have a gain with respect to DC. According to the present embodiment, it is possible to realize an electric motor drive system that can satisfy both steady-state characteristics and transient characteristics.

(Example 6)
In this embodiment, a control device 2F shown in FIG. 8 is used instead of the control device 2E of the fifth embodiment. The rest is the same as in Example 5. In FIG. 8, 21F is a speed controller that does not include an integral element. In the present embodiment, as in the fifth embodiment, an electric motor drive system that can satisfy both steady characteristics and transient characteristics can be realized.

(Example 7)
In this embodiment, the control device 2G of FIG. 9 is used instead of the control device 2 of the first embodiment. In FIG. 9, reference numeral 22 denotes a q-axis damping gain that is corrected to the electrical angular frequency ω1 ^* using the qc-axis current detection value Iqc. In this embodiment, a q-axis damping gain 22 is added to the embodiment 2 of FIG.

The operation of this embodiment will be described. In the first to sixth embodiments, as a countermeasure against load fluctuation or speed fluctuation, Iq ^* is corrected and the control side follows the load (outputs torque corresponding to the load). It is. This embodiment does not place importance on followability to load fluctuations, etc., and is intended to give top priority to trip prevention due to overcurrent during load fluctuations, that is, some speed reduction and torque reduction can be tolerated. This is useful for applications where stopping must be avoided.

  The voltage equation of the motor is

であり、Ｉｑは、

And Iq is

となる。よって、負荷が加わり、電動機速度が低下すると、電動機の誘起電圧の項（ω１・Ｋｅ）が低下し、結果的にＩｑが増加する。負荷変動による速度低下が大きい場合、
Ｉｑが過大に流れ、過電流による不具合（インバータ停止，素子破壊）が生じることも考えられる。そこで、Ｉｑの急変は、負荷変動が生じているということであるので、Ｉqcに変動があった場合に、即座に制御器内のω1^*を修正し、Ｉｑが過大を抑制する。つまり、誘起電圧ω１・Ｋｅが低下した場合、式（５）のＶｑも低減すれば、電流の増加を抑制できる。ω1^*の修正量をΔω1qとし、Ｉqcの変化率に比例した量を、ω1^*から差し引けばよい。具体的には、Ｉqcの微分、あるいは不完全微分等の進み要素で補償すればよい。

It becomes. Therefore, when a load is applied and the motor speed is decreased, the term of the induced voltage (ω1 · Ke) of the motor is decreased, and as a result, Iq is increased. If the speed drop due to load fluctuation is large,
It is also conceivable that Iq flows excessively and malfunctions due to overcurrent (inverter stop, element destruction) occur. Therefore, since a sudden change in Iq means that a load change has occurred, when Iqc changes, ω1 ^* in the controller is immediately corrected, and Iq suppresses an excessive amount. That is, when the induced voltage ω1 · Ke decreases, an increase in current can be suppressed by reducing Vq in Expression (5). The correction amount of ω1 ^* is Δω1q, and an amount proportional to the rate of change of Iqc is subtracted from ω1 ^* . Specifically, it may be compensated by a leading element such as Iqc differentiation or incomplete differentiation.

  As a result, the torque decreases when the load fluctuates, and the electric motor decelerates, but problems due to overcurrent can be suppressed. Even if the electric motor decelerates, in this embodiment, the shaft does not deviate, so that step-out does not occur. According to the present embodiment, it is possible to realize a stable motor drive system in which an overcurrent trip or the like hardly occurs.

(Example 8)
In this embodiment, the control device 2H of FIG. 10 is used instead of the control device 2 of the first embodiment. In FIG. 10, reference numeral 23 denotes a d-axis damping gain that is corrected to the electrical angular frequency ω 1 ^* using the dc-axis current detection value Idc. In this embodiment, a d-axis damping gain 23 is added to the seventh embodiment.

Next, the operation of this embodiment will be described. Similarly to the seventh embodiment, this embodiment also suppresses overcurrent during load fluctuation or speed fluctuation. In the present embodiment, not only Iqc but also Idc is used to correct ω1 ^* . In FIG. 10, the d-axis damping gain 23 receives Idc, outputs a compensation amount Δω1d corresponding to the fluctuation component, and adds it to ω1 ^* .

  When the load fluctuation of the electric motor occurs, for example, as shown in the vector diagram of FIG. 11, an axial error Δθ occurs between the d axis that is the real axis of the electric motor and the control axis dc axis. Δθ converges to zero by the PLL function of the present invention, but the current fluctuates / increases during that time, which may cause an overcurrent trip. The effect of Δθ appears not only on the q-axis current but also on the d-axis. The relationship between the voltage vector V1 when there is no axis error and V1 ′ immediately after the occurrence of the axis deviation Δθ is as shown in FIG. The d-axis voltage component Vd ′ decreases due to the influence of the axis deviation. Due to this influence, Id largely changes to the minus side. In particular, the reverse salient pole type motor (Ld <Lq) has a small value of Ld, so that the influence of Id accompanying the change in Vd becomes large.

Using the above phenomenon, the magnitude of the load fluctuation can be found from the change in Idc. Therefore, by using the value of Idc, an amount proportional to the differential or incomplete differentiation can be added to ω1 ^* to reduce the axis error and to suppress problems such as overcurrent. Since Id decreases to minus when the load increases, Δω1d is added as it is for compensation of ω1 ^* . In addition, in the present embodiment using the Iq ^* generator and the like, there is no steady axis deviation, so it is not necessary to strictly adjust the d-axis damping gain setting, and d-axis damping is performed to suppress overcurrent. What is necessary is just to set a gain. As described above, according to the present embodiment, the countermeasure against overcurrent at the time of load fluctuation can be performed with higher sensitivity, and a stable motor drive system can be realized.

Example 9
In this embodiment, the control device 2J of FIG. 12 is used instead of the control device 2 of the first embodiment. In FIG. 12, reference numeral 10J denotes an Id ^* generator that determines the value of Id ^* based on Iq ^* .

  Next, the operation principle of this embodiment will be described. Some permanent magnet type motors generate motor torque by combining the torque of a permanent magnet and the reluctance torque due to the saliency (reverse saliency) of the motor. In the case of this type of motor, the maximum torque point of the motor is at the point where Id is set to a negative value, and it is not a good idea in terms of efficiency to control to Id = 0. Therefore, when the electric motor is always driven with the maximum efficiency, the electric motor may be driven in a state where the maximum torque is always obtained. The condition for obtaining the maximum torque is expressed by equation (6),

ただし、Φｍ：永久磁石磁束、Ｌｄ≠Ｌｑ
となり、Ｉｑが定まれば、最大トルクを得るＩｄが決定する。本実施例では、Ｉd^*発生器１０Ｊで、Ｉq^*を用いて式（６）の演算処理をする。その結果、常に最大トルク（最大効率）で電動機駆動ができる。

Where Φm: permanent magnet magnetic flux, Ld ≠ Lq
If Iq is determined, Id for obtaining the maximum torque is determined. In the present embodiment, the Id ^* generator 10J performs calculation processing of Expression (6) using Iq ^* . As a result, the motor can always be driven with the maximum torque (maximum efficiency).

It is possible to use Iqc instead of Iq ^* for the calculation of equation (6). However, since the fluctuation of Iqc during transition is severe, the entire control system may be destabilized. Further, maximization of efficiency, if function in the steady state, because it is energy saving devices, there is no problem with Iq ^* which is the output of the Iq ^* generator. According to the present embodiment, it is possible to realize an electric motor drive system capable of operating at the maximum electric motor efficiency.

(Example 10)
In this embodiment, the control device 2K of FIG. 13 is used instead of the control device 2 of the first embodiment. In FIG. 13, reference numeral 24 denotes a current controller for making the value of Idc coincide with Id ^* .

The operation principle of this embodiment will be described. As described in the first to ninth embodiments, the speed controller required in the prior art can be deleted in the present invention, so that the control system can be simplified and an electric motor drive system with few adjustment points can be realized. The problem is how to create a command for the torque current Iq. In the above embodiment, Iq ^* is mainly created using the current detection value Iqc. On the other hand, since Id ^* can be given regardless of speed and load torque, an arbitrary command current can be set. (It is set to zero except when the efficiency maximization in FIG. 12 is performed.) Therefore, a conventional current controller can be added for Id.

As shown in the equation (4), there is an interference term between the d-axis and the q-axis of the motor, and as ω1 increases, the interference between the d-q axes becomes stronger. The vibration is likely to occur. Since Iq ^* is passed through a large filter in Iqc, there is no ability to suppress this vibration. However, when the current controller 24 is added to Idc, a function of suppressing the interference term between the dq axes occurs. That is, in order to make Idc coincide with Id ^* (a constant value), vibration is to be suppressed. As a result, the stability and response performance of the entire control system are improved. In this embodiment, an electric motor drive system that can improve the response of the entire control system can be realized by adding a current controller to the d-axis.

(Example 11)
In this embodiment, the control device 2L of FIG. 14 is used instead of the control device 2 of the first embodiment. In FIG. 14, reference numeral 12L denotes a voltage command calculator that can change the control parameter with an external signal, and 24L denotes a current controller that changes the control parameter in order to make the value of Idc coincide with Id ^* . FIG. 15 shows the internal configuration of the voltage command calculator 12L and the current controller 24L, where 121L is a resistance value setter, 241 is a current control proportional gain, and 242 is a current control integral gain.

  Next, the operation of this embodiment will be described. As described in the tenth embodiment, the Id current controller has a function of improving the control system response. In this embodiment, a current controller is used as a constant setting error corrector. If the actual motor constant matches the motor constant set value inside the controller, the integral element of the current controller will be zero in steady state, so the output of the current controller should be zero. is there. However, if there is an error in the set value of the constant, the current controller continues to hold the value in an attempt to correct that amount. On the contrary, if the output of the current controller is used, the deviation of the set value can be corrected.

In the case of an electric motor, an error may be included in the resistance value due to wiring routing, the resistance of the inverter, or the like. In this embodiment, the resistance setting deviation is automatically adjusted. In FIG. 15, the deviation between Id ^* and Idc is calculated and input to a current controller 24L comprising a proportional / integral element. The output of the current controller 24L is set as a correction value ΔR of the resistance R, and the resistance value in the voltage command calculator 12L is corrected. The correction of the resistance R is completed when Id ^* and Idc match. Since the voltage command calculator 12L corrects not only the d-axis but also the q-axis resistance, the voltage command calculation accuracy is improved for both the d-q axes. Thus, in this embodiment, an electric motor drive system that can automatically adjust electric motor constants can be realized.

(Example 12)
In this embodiment, the control device 2M of FIG. 14 is used instead of the control device 2 of the first embodiment. In FIG. 16, reference numeral 10M is an Id ^* generator that switches an internal current command value by an external signal RF, 24M is a current controller that can stop the current control function by an external signal RF, and 25 is a speed command ωr ^* . An R adjustment signal generator for setting the output (RF) to “1” when it is below a predetermined value set in advance, and “0” above the predetermined value, and 26 sets a current value for correcting the resistance set value. It is a current value setter. FIG. 17 shows the internal configuration of the Id ^* generator 10M and the current controller 24M. In FIG. 17, reference numeral 27 denotes a switch that is switched by a signal RF.

Next, the operation of this embodiment will be described. In Example 11, the set value of the resistance R in the control device was automatically adjusted using a current controller. In this embodiment, not only the resistance of the motor but also the resistance set value is corrected by taking into account the inductance and the power generation constant, which change due to the influence of magnetic saturation and ambient temperature. The R adjustment signal generator 25 sets a range in which the resistance set value is adjusted in advance, and outputs a signal with RF = 1 for a speed command below a predetermined value and RF = 0 at a predetermined value or higher. The Id ^* generator 10M receives the RF signal and switches the current command with the switch 27. In the case of RF = 1 (when the resistance set value is corrected), Id ^* = Id0, and the resistance value is adjusted by supplying a current for adjusting the resistance value. When RF = 0, the motor is driven using normal Id ^* (= 0). The current controller 24M operates the current controller to correct the resistance set value by setting the switch 27 to the “1” side during the period of RF = 1. When RF = 0, the switch 27M On the “0” side, the input of the current controller is set to zero. Even after the input of the current controller becomes zero, the value of the integrator 8 remains, so that the correction value ΔR continues to be output.

  In the present embodiment, the resistance set value in the controller is corrected at a predetermined speed or less to tune the resistance value with higher accuracy. As a result, highly accurate voltage command calculation can be realized, and control performance is improved.

(Example 13)
The present embodiment will be described with reference to FIGS. In this embodiment, the control device 2N of FIG. 14 is used instead of the control device 2 of the first embodiment. In FIG. 18, reference numeral 1N denotes an ωr ^* generator that outputs a speed command ωr ^* , 11N denotes an Iq ^* generator that switches an internal current command value by an external signal SF, and 17N denotes an AC phase correction by an external signal SF. 25L is an R adjustment signal generator that generates an adjustment signal for a resistance setting value, 28 is a frequency command ωr ^* , and ωr ^* exceeds a preset predetermined frequency. A switching signal generator for changing the operation mode switching signal SF from 0 to 1, 29 is a starting signal generator for outputting a signal necessary for starting the motor, and 30 is a starting Id for outputting Id ^* at the time of starting. ^* Generator 31 is an activation processor that controls the sequence of activation of the electric motor.

FIG. 19 shows details of the Iq ^* generator 11N and the magnetic pole axis estimation gain 17N. Both blocks have a built-in switch 27. When the switching signal SF = 0, Iq ^* = 0 and Δω1 = 0 respectively. When the switching signal SF = 1, the Iq ^* generator 11N and the magnetic pole axis estimation gain 17N are switched to the operations of the Iq ^* generator 11 and the magnetic pole axis estimation gain 17 in the above embodiments.

Next, the operation of this embodiment will be described. When the synchronous motor is driven without a speed / position sensor, for example, Δθ is calculated from the relational expression shown in Expression (3) to estimate the magnetic pole axis. However, in order to calculate Equation (3) with high accuracy, the motor speed must be at least about 5-10% of the rating at the minimum, and Vdc ^* and Vqc ^* are too small during stop and low speed. Sufficient calculation accuracy cannot be obtained. In the present invention, the synchronous motor is started in the three operation modes shown in FIG.

First, in the “DC positioning mode”, a direct current is supplied to the dc axis for control, and the rotor is moved so that the magnetic pole axis (d axis) and the dc axis coincide. Actually, as shown in FIG. 20C, Id ^* is increased in a ramp shape to position the rotor. Id ^* receives a signal from the activation processor 31 and outputs it from the activation Id generator 30. During this time, simultaneously with the DC positioning, the resistance set value in the control is corrected. A signal of RF = 1 is output from the R adjustment signal generator 25L, and the resistance value correction described in the twelfth embodiment is performed using the current controller 24L and the voltage command calculation 12L.

Next, in the “synchronous start mode”, the switching signal generator 28 continues to output SF = 0,
The motor is started under the conditions of Iq ^* = 0, Δω1 = 0, Id ^* = Im (Im: set current value for synchronous start). As shown in FIG. 20A, simultaneously with the synchronous start, the output of the R adjustment signal generator is set to RF = 0, and the adjustment of the resistance R is completed. The driving method in the synchronous start mode is equivalent to the conventional V / F control, and the motor is accelerated with the shaft misalignment remaining as shown in FIG. In the V / F control, the applied voltage of the motor is simply increased according to the frequency command ωr ^* , and the motor can be started without being stepped out by setting the acceleration rate gently.

Next, when the speed command ωr ^* reaches 5 to 10% of the rated speed of the motor, the output of the switching signal generator 28 is switched to SF = 1 to enter the sensorless drive mode. When SF = 1, the switch 27 that switches Id ^* , the switch 27 in the Iq ^* generator 11N, and the magnetic pole axis estimation gain 17N are switched to the “1” side, and control equivalent to the second embodiment is performed. It becomes a composition. As a result, the axis error Δθ converges to zero and stable vector control driving is achieved.

  A series of startup processing of this embodiment is automatically performed according to a program preset in the startup processor 31. According to the present embodiment, the synchronous motor can be quickly started from the stop time to the high speed rotation without using the speed / position sensor.

1 is a configuration diagram of Example 1. FIG. It is the structure schematic of the apparatus which mounted this invention. 3 is a configuration diagram of a voltage command calculator in Embodiment 1. FIG. FIG. 6 is a configuration diagram of Example 2. FIG. 6 is a configuration diagram of Example 3. FIG. 6 is a configuration diagram of Example 4. FIG. 10 is a configuration diagram of Example 5. FIG. 10 is a configuration diagram of Example 6. FIG. 10 is a configuration diagram of Example 7. FIG. 10 is a configuration diagram of Example 8. It is a vector diagram showing the influence of the voltage command when the axis error Δθ occurs. 10 is a configuration diagram of Example 9. FIG. FIG. 10 is a configuration diagram of Example 10. FIG. 16 is a configuration diagram of Example 11. It is a block diagram of the current controller and voltage command calculator of Example 11. FIG. 16 is a configuration diagram of Example 12. It is a block diagram of the current controller and voltage command calculator of Example 12. It is a block diagram of 13th Embodiment by this invention. It is a block diagram of the Iq ^* generator and magnetic pole axis estimation gain of Example 13. It is an operation | movement waveform diagram of Example 13. It is a block diagram of the synchronous motor vector control drive system of a prior art. It is a block diagram of the synchronous motor drive system by another prior art V / F control. It is a vector diagram which shows the relationship between the dq axis | shaft in an electric motor, the dc-qc axis | shaft in a control apparatus, and axial error (DELTA) (theta). It is a vector diagram showing current and voltage when there is no axis error in vector control. It is a vector diagram which shows the relationship between the electric current and voltage in V / F control.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 ... Speed command generator, 2 ... Control apparatus, 3 ... PWM (pulse width modulation) generator, 4 ... Inverter, 5 ... Synchronous motor, 6 ... Current detector, 7 ... Conversion gain (P is the number of poles of the motor) , 8 ... integrator, 9 ... dq coordinate converter, 10 ... Id ^* generator, 11 ... Iq ^* generator, 12 ... voltage command calculator, 13 ... dq inverse converter, 41 ... DC power supply unit, 42 ... inverter Main circuit section 43... Gate driver 411. Power supply 412 diode bridge 413 smoothing capacitor.

Claims (3)

A control device for controlling a motor based on a detected value of a motor current,
Pulse width control for creating a current command value to be applied to the motor from a value obtained by adding a delay element to the torque axis component (q-axis component) of the detected value of the motor current , and driving the inverter based on the current command value A control device that generates a signal and applies the pulse width control signal to the inverter to drive the motor. A motor control method for controlling a motor based on a detected value of a motor current,
Pulse width control for creating a current command value to be applied to the motor from a value obtained by adding a delay element to the torque axis component (q-axis component) of the detected value of the motor current , and driving the inverter based on the current command value A motor control method, comprising: generating a signal and applying the pulse width control signal to the inverter to drive the motor. A module comprising an inverter and a control device that controls the motor based on the detected value of the motor current,
Pulse width control for creating a current command value to be applied to the motor from a value obtained by adding a delay element to the torque axis component (q-axis component) of the detected value of the motor current , and driving the inverter based on the current command value A module that generates a signal and applies the pulse width control signal to the inverter to drive the motor.

Images