JP5157574B2 - Switching power supply - Google Patents

Description

  The present invention relates to a switching power supply that improves a power factor by creating a stable DC power supply from an AC power supply and operating the input voltage and input current of a chopper circuit so as to be similar in phase and in phase.

As a control method of such a switching power supply, and the critical mode control method for controlling to turn on the switching element, a coil (inductor) current is controlled so as not to become zero when the coil (inductor) current becomes 0 There is a continuous current mode control system.
In general, the former critical mode control method performs switching when the coil ( inductor) current is 0. Therefore , the switching noise is low, which is advantageous for low noise. However, since the current ripple becomes large, the coil ( inductor) is effective. In addition, it is difficult to apply to applications where the stress on the diode and the diode is large and the load is large.

On the other hand, the latter continuous current mode control method has larger switching noise than the critical mode control method, but the current ripple is small and the stress applied to the coil (inductor) and diode is small. is there.
Since general control ICs operate only in either the critical mode or the continuous current mode, it is necessary to properly use the control ICs according to the load, and development costs such as requiring a power supply system design change are required. To increase.

  FIG. 11 shows an example of a continuous current mode control switching power supply disclosed in Patent Document 1, for example. In the figure, 1 is an AC power source, 2 is a rectifier circuit, 3 and 6 are capacitors, 4 is an inductor, 5 is a diode, 7 is a switching element such as a MOSFET (metal oxide field effect transistor), 8 is a voltage error amplifier, 9 is a multiplier, 10 is a comparator, 11 is a monostable multivibrator, 12 is a current detection resistor, and 13 is a drive circuit.

  The AC power supply 1 is full-wave rectified by a rectifier circuit 2 composed of a diode bridge, and after high frequency noise is removed by a capacitor 3, current is supplied to the capacitor 6 through the inductor 4 and the diode 5, thereby being smoothed. A DC voltage Vout is output. The switching element 7 is connected between the inductor 4 and the diode 5, and turns on / off the current flowing from the inductor 4 to the diode 5.

Since the voltage error amplifier 8 amplifies the error between the output voltage Vout and the reference voltage Vref and supplies the amplified error voltage to the multiplier 9, the multiplier 9 multiplies the amplified error voltage Verr and the input voltage Vin to obtain the input voltage Vin. A threshold (threshold) signal Vth having the same phase and similar shape and having an amplitude proportional to the error voltage Verr is generated.
On the other hand, the current flowing through the inductor 4 is converted into the current detection signal Vi by the current detection resistor 12 and compared with the threshold signal Vth by the comparator 10. Since the output of the comparator 10 is input to the trigger input of the monostable multivibrator 11, the monostable multivibrator 11 keeps the output at a low level for a certain period after the trigger signal is input, and then changes the output to a high level. Let The output of the monostable multivibrator 11 is input to the drive circuit 13, and the drive circuit 13 turns on the switching element 7 when the input is at a high level and turns it off when the input is at a low level.

  When the switching element 7 is turned on with such a configuration, the current from the inductor 4 increases and the current detection signal Vi rises. When the current detection signal Vi exceeds the threshold signal Vth, the output of the comparator 10 becomes high level, a trigger signal is input to the monostable multivibrator 11, the monostable multivibrator 11 becomes low level, and the drive circuit 13 is turned on. Accordingly, the switching element 7 is turned off. As a result, the current from the inductor 4 gradually decreases, but since the low level period of the monostable multivibrator 11 is set so that the current does not become zero, it is monostable when the current decreases to some extent. The output of the multivibrator 11 changes to a high level, and the switching element 7 is turned on via the drive circuit 13.

  FIG. 12 illustrates the above operation, and shows a state in which the peak of the current flowing through the inductor 4 is controlled so as to coincide with the threshold signal Vth having the same phase as that of the input voltage Vin. At this time, since the on-time is changed and the off-time is fixed, the switching frequency changes and the frequency of the generated noise also changes. As a result, since the noise spectrum is dispersed, the noise can be reduced. FIG. 12A is a diagram for explaining the relationship between the threshold signal Vth and the current detection signal Vi, and FIG. 12B shows an ON / OFF waveform of the switching element.

In order to improve the power factor, the input current needs to have the same phase and similar shape as the input voltage, and for that purpose, the on / off duty ratio needs to be varied widely from 0% to nearly 100%. The voltage across the inductor differs depending on whether the input voltage is 100V or 200V, and also constantly changes during one cycle of the AC input voltage, so the time constant (di / dt) of the current flowing through the inductor varies greatly. doing. As a result, the current change in a certain time greatly changes depending on the value of the input voltage, the phase and the state of the load. Therefore, when the off-time is fixed as in the conventional example, the necessary current change becomes excessive and insufficient, and there is a problem that there is a limit to the power factor improvement.
In the case of the critical mode control method, it is necessary to turn on the current when the current becomes 0. When the off time is fixed as in the conventional example, the critical mode control is difficult.

  On the other hand, there is a method as in Patent Document 2. This is because a second threshold signal Vth2 proportional to the threshold signal Vth1 is provided, the switching element is turned off when the current flowing through the inductor reaches the threshold signal Vth1, and the current flowing through the inductor is There has been proposed a power factor improving method for turning on the switching element when it falls below the threshold signal Vth2. According to this, the on time and the off time are not fixed, and the on / off can be automatically performed in an optimum time depending on the state of the input voltage and the state of the load, and the power factor can be improved.

Japanese Patent Laid-Open No. 04-168975 JP 2006-067671 A

FIG. 13 shows an example of the control method of Patent Document 2. As shown in FIG. 13, when the second threshold signal Vth2 changes in proportion to the first threshold signal Vth1, the input voltage Vin is a full-wave rectified waveform of the AC power supply and changes every moment. As a result, the first threshold value Vth1 also changes and changes from the reference potential to the peak potential. In the vicinity of the reference potential, the second threshold signal Vth2 also decreases, the difference between the first threshold signal Vth1 and the second threshold signal Vth2 decreases, and the amplitude of the coil (inductor) current decreases. Therefore, there is a problem that the switching frequency becomes very high, and noise and switching loss increase. FIG. 13A is a diagram for explaining the relationship between the first and second threshold signals Vth1 and Vth2 and the current detection signal Vi, and FIG. 13B shows the ON / OFF waveforms of the switching elements.

  In addition, when the second threshold signal Vth2 is set to a constant ratio with respect to the first threshold signal Vt1, the first current is reduced when the average current flowing through the inductor is small due to the input voltage and load conditions. Since the width between the threshold signal Vt1 and the second threshold signal Vth2 becomes small as shown in FIG. 14B, there is a problem that the switching frequency becomes high, and noise and switching loss increase. FIG. 14A shows a case where the first threshold signal Vt1 and the second threshold signal Vth2 are relatively wide.

  Therefore, an object of the present invention is to reduce noise and loss to improve power factor, in particular to suppress increase in switching frequency to reduce noise and loss and improve power factor, and further to a critical mode control system. The current continuous mode control system can be easily switched to simplify the power supply system and reduce development costs.

In order to solve such a problem, in the invention of claim 1, a rectifier circuit that obtains a pulsating current output by full-wave rectifying an AC power source;
A chopper circuit comprising a switching element connected to the rectifier circuit for turning on and off the current flowing through the inductor, and a capacitor for smoothing the current supplied from the inductor to obtain a DC output;
First signal generating means for generating a first threshold signal having the same phase as the input voltage of the chopper circuit and having a waveform similar in shape and having an amplitude proportional to the error of the output voltage of the chopper circuit;
A series circuit of a first resistor and a variable resistor formed by connecting a transistor in series or in parallel to at least one set of second resistors is connected between the first threshold signal and the reference potential, The transistor is driven by at least one signal of an error of an input voltage, a current flowing through the inductor, or an output voltage, and the first threshold signal is output from a connection point between the first resistor and a variable resistor. Second signal generating means for generating a second threshold signal in which the difference between them is substantially constant ;
Current detecting means for detecting a current flowing through the inductor; and when the detected current value reaches a level of the first threshold signal, the switching element is turned off and the current flowing through the inductor is the second current. And switching control means for turning on the switching element when the threshold signal level is lowered.

In the first aspect of the invention, the variable resistor of the second signal generating means is a circuit in which at least one set of resistors and switches are connected in series, or a plurality of series circuits of resistors and switches are connected in parallel. 3. The circuit according to claim 2, wherein the switch can be turned on / off by at least one signal of an error of an input voltage, a current flowing through the inductor, or an output voltage .

In the invention of claim 3, a rectifier circuit that obtains a pulsating flow output by full-wave rectifying an AC power source;
A switching element connected to the rectifier circuit for turning on and off the current flowing through the inductor;
A chopper circuit comprising a capacitor that obtains a DC output by smoothing the current supplied from the inductor;
First signal generating means for generating a first threshold signal having the same phase as the input voltage of the chopper circuit and having a waveform similar in shape and having an amplitude proportional to the error of the output voltage of the chopper circuit;
Second signal generating means for generating a second threshold signal proportional to the first threshold signal;
Modulating means for modulating said second threshold signal,
Current detecting means for detecting a current flowing through the inductor;
When the detection current value of the current detection means reaches the level of the first threshold signal, the switching element is turned off, and the current flowing through the inductor is lower than the level of the second threshold signal. And switching control means for turning on the switching element.

In the third aspect of the invention, the second signal generating means is formed of a series circuit of first and second resistors connected between the first threshold signal and a reference potential, The output of the connection point of the first and second resistors can be the second threshold signal (invention of claim 4 ).
In the invention of claim 3 or 4, the modulating means includes a transistor connected between the second threshold signal and a reference potential, and the transistor based on the first threshold signal. (The invention according to claim 5 ), or the modulation means includes a transistor connected between the second threshold signal and a reference potential, and the first the threshold signal and the transistor can be composed of a drive circuit for driving the on the basis of the difference between the second threshold signal (the invention of claim 6), or the modulation means, the second And a drive circuit that drives the transistor based on a difference between the first threshold signal and the control signal. Item 7).

According to the present invention, the switching element is controlled to be turned on / off so that the current flowing through the inductor is between two threshold values having the same phase as the input voltage, a similar shape, and an amplitude proportional to the voltage error signal. The on / off time is not fixed and can be automatically turned on / off in an optimum time depending on the state of the input voltage and the state of the load, and the power factor can be improved.
In addition, since the switching frequency changes due to changes in the on-time and off-time and the generated noise spectrum is dispersed, noise can be reduced. Furthermore, when the threshold signal is close to the reference potential, the amplitude of the coil (inductor) current is not reduced by setting one threshold value as the reference potential, thereby suppressing an increase in switching frequency. Thus, high frequency noise can be reduced.
In addition, by making the control signal input below the reference potential, it can be easily changed to the critical mode control method, so that the design change becomes easy and the development cost can be reduced.

FIG. 1 is a circuit diagram showing an embodiment of the present invention. This is a modification of the conventional example shown in FIG. 11, and the same components as those in FIG.
As can be seen from FIG. 1, a series circuit of a diode 14G and a resistor 14B between the first threshold signal Vth1 and the reference potential 14C, and 2 instead of the comparator 10 is different from that shown in FIG. A feature is that two comparators 10A and 10B and a flip-flop 15 in place of the monostable multivibrator 11 are added. Hereinafter, differences will be mainly described.

Since the voltage error amplifier 8 amplifies an error between the output voltage Vout and the reference voltage Vref and supplies the amplified error voltage to the multiplier 9, the multiplier 9 multiplies the amplified error voltage Verr and the input voltage Vin to obtain the input voltage Vin. The process is the same as that of FIG. 11 until the first threshold value signal Vth1 having the same phase and similar shape as the above and having an amplitude proportional to the error voltage Verr.
The first threshold signal Vth1 is supplied with a second threshold voltage by a resistor 14B connected between the cathode of the diode 14G whose anode is connected to the first threshold signal Vth1 and the reference potential 14C. A value signal Vth2 is generated. The second threshold signal Vth2 may be obtained by using a circuit in which a resistor is connected in series to a parallel circuit of a diode and a resistor.

  The current flowing through the inductor 4 is converted into a current detection signal Vi by a current detection circuit 12 including a resistor, compared with a threshold signal Vth1 by a comparator 10A, and compared with a threshold signal Vth2 by a comparator 10B. The output of the comparator 10A is input to the reset terminal of the flip-flop 15, and the output of the comparator 10B is input to the set terminal of the flip-flop 15. The output of the flip-flop 15 is input to the drive circuit 13, and the drive circuit 13 turns on the switching element 7 when the input is at a high level and turns it off when the input is at a low level. The current detection circuit 12 is provided with a low-pass filter (not shown), whereby the harmonic component of the current detection signal Vi is removed, and only the low frequency component is supplied to the comparators 10A and 10B.

  When the switching element 7 is turned on with such a configuration, the current from the inductor 4 increases and the current detection signal Vi rises. When the current detection signal Vi exceeds the first threshold signal Vth1, the output of the comparator 10A becomes high level, the flip-flop 15 is reset, and the output of the flip-flop 15 becomes low level, via the drive circuit 13 Thus, the switching element 7 is turned off. As a result, when the current from the inductor 4 gradually decreases and the current detection signal falls below the second threshold signal Vth2, the output of the comparator 10B becomes high level, and the flip-flop 15 is set. The output of 15 becomes high level, and the switching element 7 is turned on via the drive circuit 13.

  FIG. 2 illustrates the operation of FIG. 1, and the peak of the current (current detection signal Vi) flowing through the inductor 4 coincides with the first threshold signal Vth1 having the same phase and similar shape as the input voltage Vin. It shows a state in which the bottom of the current is controlled so as to coincide with the second threshold signal Vth2 having the same phase and similar shape as the input voltage Vin. For this reason, the spectrum of noise is dispersed, and the noise can be reduced. 2A is a diagram for explaining the relationship between the first and second threshold signals Vth1 and Vth2 and the current detection signal Vi, and FIG. 2B shows an ON / OFF waveform of the switching element.

That is, as shown in FIG. 13, when the second threshold signal Vth2 changes in proportion to the first threshold signal Vth1, the input voltage Vin is sometimes a full-wave rectified waveform of the AC power supply. Along with this change, the first threshold signal Vth1 changes accordingly, and changes from the reference potential to the peak potential. In the vicinity of the reference potential, the second threshold signal Vth2 also decreases, the difference between the first threshold signal Vth1 and the second threshold signal Vth2 decreases, and the amplitude of the coil (inductor) current decreases. Therefore, the switching frequency becomes very high, and noise and switching loss increase.

On the other hand, in FIG. 1, the second threshold signal Vth2 is output as a connection point between the diode 14G and the resistor 14B connected between the first threshold signal Vth1 and the reference potential 14C. Thus, the value is lower than the first threshold signal Vth1 by the forward voltage of the diode 14G. That is, the difference between the first threshold signal Vth1 and the second threshold signal Vth2 can be kept substantially constant (see FIG. 2A), and the amplitude of the coil (inductor) current even near the reference potential. Does not decrease and the switching frequency does not increase. Thereby, increase of noise and switching loss can be prevented.

FIG. 3 shows another embodiment of the present invention. A driving circuit for driving the transistor 14D by connecting a series circuit of the first resistor 14A and the second resistor 14B between the first threshold signal and the reference potential, and connecting the transistor 14D to both ends of the resistor 14B. A feature is that 14E is provided, and the others are the same as in FIG.
That is, the voltage error amplifier 8 amplifies the error between the output voltage Vout and the reference voltage Vref and gives it to the multiplier 9, so that the multiplier 9 multiplies the amplified error voltage Verr and the input voltage Vin to obtain an input. The process is the same as in FIG. 1 until the first threshold signal Vth1 having the same phase as the voltage Vin and a similar shape and having an amplitude proportional to the error voltage Verr.

The first threshold signal Vth1 is divided by resistors 14A and 14B connected between the first threshold signal Vth1 and the reference potential 14C, and is proportional to the first threshold signal Vth1. Vth2 is generated.
Further, a transistor 14D such as a P-channel field effect transistor (FET) is connected to both ends of the resistor 14B, the gate of the transistor 14D is connected to the transistor drive circuit 14E, and the transistor drive circuit 14E receives the threshold signal Vth1. Is output to the transistor 14D.

  The current flowing through the inductor 4 is converted into a current detection signal Vi by a current detection circuit 12 including a resistor, compared with a threshold signal Vth1 by a comparator 10A, and compared with a threshold signal Vth2 by a comparator 10B. The output of the comparator 10A is input to the reset terminal (R) of the flip-flop 15, and the output of the comparator 10B is input to the set terminal (S) of the flip-flop 15. The output of the flip-flop 15 is input to the drive circuit 13, and the drive circuit 13 turns on the switching element 7 when the input is at a high level and turns it off when the input is at a low level.

  When the switching element 7 is turned on with such a configuration, the current from the inductor 4 increases and the current detection signal Vi rises. When the current detection signal Vi exceeds the first threshold signal Vth1, the output of the comparator 10A becomes high level, the flip-flop 15 is reset, and the output of the flip-flop 15 becomes low level, via the drive circuit 13 Thus, the switching element 7 is turned off. As a result, when the current from the inductor 4 gradually decreases and the current detection signal falls below the second threshold signal Vth2, the output of the comparator 10B becomes high level, and the flip-flop 15 is set. The output of 15 becomes high level, and the switching element 7 is turned on via the drive circuit 13.

  When the first threshold signal Vth1 is large, the output from the transistor drive circuit 14E is large, the transistor 14D is in an insulated state, and the second threshold signal Vth2 has a value determined by the voltage dividing ratio of the resistors 14A and 14B. Become. When the first threshold signal Vth1 decreases, the transistor 14D shifts from the insulated state to the conductive state, and the second threshold signal Vth2 decreases accordingly. Further, when the first threshold signal Vth1 decreases, the transistor 14D becomes conductive, and the second threshold signal Vth2 becomes approximately the reference potential 14C.

FIG. 4 explains the operation of FIG. 3, and the peak of the current (current detection signal Vi) flowing through the inductor 4 coincides with the first threshold signal Vth1 having the same phase and similar shape as the input voltage Vin. It shows a state in which the bottom of the current is controlled so as to coincide with the second threshold signal Vth2 having the same phase and similar shape as the input voltage Vin. Even when the first threshold signal Vth1 is smaller than the point A, the difference between the first threshold value Vth1 and the second threshold signal Vth2 does not become small, so the amplitude of the coil (inductor) current does not become small. Since the upper limit of the switching frequency is limited, it is possible to suppress an increase in noise and switching loss. 4A is a diagram for explaining the relationship between the first and second threshold signals Vth1 and Vth2 and the current detection signal Vi, and FIG. 4B shows an ON / OFF waveform of the switching element.

FIG. 5 shows a first modification of FIG. The difference from FIG. 3 is that the transistor drive circuit 14E amplifies the difference between the first threshold value signal Vth1 and the second threshold value Vth2, and drives the transistor 14D.
As a result, when the amplitude of the coil (inductor) current is reduced, the transistor 14D is turned on so that the amplitude of the coil (inductor) current is not reduced and the upper limit of the frequency is limited. An increase in switching loss can be suppressed.

FIG. 6 shows a second modification of FIG. As is apparent from FIG. 3, the difference from FIG. 3 is that a control signal (selection signal) 14F is introduced into the transistor drive circuit 14E.
In this circuit, when a potential equal to or lower than the reference potential 14C is input as the control signal 14F, the transistor 14D is turned on, the second threshold signal Vth2 is always fixed to the reference potential 14C, and the coil (inductor) current is The critical mode control in which the switching element 7 is turned on at 0 is achieved, and the noise is reduced.

FIG. 7 shows still another embodiment of the present invention. In addition, about the part same as the example so far, the same code | symbol is attached | subjected and description is abbreviate | omitted.
The AC power source 1 is full-wave rectified by a rectifier circuit 2 composed of a diode bridge, high-frequency noise is removed by a capacitor 3, current is supplied to the capacitor 6 via an inductor 4 and a diode 5, and smoothed DC The voltage Vout is output. A switching element 7 such as a MOSFET is connected between the inductor 4 and the diode 5 and turns on and off the current flowing from the inductor 4 to the diode 5.

The voltage error amplifier 8 amplifies the error between the output voltage Vout and the output reference voltage Vref, and outputs the amplified error voltage to the multiplier 9. The multiplier 9 multiplies the amplified error voltage Verr and the input voltage Vin to obtain the input voltage Vin. A first threshold signal Vth1 that is similar in phase and has an amplitude proportional to the error voltage Verr is generated. The first threshold signal Vth1 is divided by a resistor 14A and a variable resistor 14H connected in series between the first threshold signal Vth1 and the reference potential 14C, and the first threshold signal Vth1 is based on the first threshold signal Vth1. 2 threshold signal Vth2 is generated.
Here, the variable resistor 14H changes the resistance value according to the magnitude of the input voltage Vin. Details will be described later.

  The current flowing through the inductor 4 is converted into the current detection signal Vi by the current detection circuit 12 including the current detection resistor 12A, compared with the threshold signal Vth1 by the comparator 10A, and compared with the threshold signal Vth2 by the comparator 10B. Is done. The output of the comparator 10A is connected to the reset terminal of the flip-flop 15, and the output of the comparator 10B is connected to the set terminal of the flip-flop 15. The output of the flip-flop 15 is input to the drive circuit 13, and the drive circuit 13 turns on the switching element 7 when the input is at a high level and turns it off when the input is at a low level.

  When the switching element 7 is turned on with such a configuration, the current from the inductor 3 increases and the current detection signal Vi increases. When the current detection signal Vi exceeds the first threshold signal Vth1, the output of the comparator 10A becomes high level, the flip-flop 15 is reset, the output Q of the flip-flop 15 becomes low level, and switching is performed via the drive circuit 13. Element 7 is turned off. As a result, the current from the inductor 4 gradually decreases, and when the current detection signal Vi falls below the second threshold signal Vth2, the output of the comparator 10B becomes high level, the flip-flop 15 is set, and the output of the flip-flop 15 Q becomes a high level, and the switching element 7 is turned on via the drive circuit 13.

  FIG. 9 is a detailed view of the variable resistor 14H. The resistor 14J connected to the resistor 14A and the drain terminal of the transistor 14D are connected in series, and the source terminal of the transistor 14D is connected to the reference potential 14C. The gate terminal of the transistor 14D is connected to the transistor drive circuit 14E. The transistor drive circuit 14E generates a transistor drive signal from the output of the peak voltage detection circuit 14K of the input voltage Vin and the bias voltage Vbias, and outputs the transistor drive signal to the transistor 14D. For the transistor 14D, it is preferable to use an element having linearity such as an N-channel field effect transistor, and by using the N-channel field effect transistor in the linear region, its on-resistance is used as a variable resistance. Can do.

  Here, even when the peak voltage of the input voltage Vin is small, the bias voltage Vbias is set to a voltage that allows the transistor drive circuit 14E to output a voltage that does not turn off the transistor 14D. Further, when the peak voltage of the input voltage Vin is large, the transistor drive circuit 14E outputs a sufficiently large voltage that causes the transistor 14D to be in a low resistance state. The transistor drive circuit 14E is composed of an operational amplifier or the like. The second threshold signal Vth2 has a value determined by the voltage dividing ratio of the resistor 14A and the variable resistor 14H. When the input voltage Vin is high and the peak voltage is large, the output from the transistor drive circuit 14E is high, and the transistor 14D Enters a low resistance state, and the voltage division ratio between the first threshold signal Vth1 and the second threshold signal Vth2 increases.

  Here, in the example of FIG. 9, the variable resistor 14H is configured by a series circuit of the resistor 14J and the transistor 14D, but the linear region of the transistor 14D is wide, and the first threshold signal Vth1 and the second threshold signal When a desired voltage division ratio with Vth2 is obtained, the resistor 14J can be omitted. The minimum required voltage dividing ratio between the first threshold signal Vth1 and the second threshold signal Vth2 is determined by the resistor 14A and the resistor 14J, and the transistor 14D takes charge of the variable component. Thus, a small transistor can be selected as the transistor 14D, which is advantageous in terms of cost.

  FIG. 8 shows the operation of FIG. 7. The peak of the current flowing through the inductor 4 coincides with the first threshold signal Vth1 having the same phase as the input voltage Vin, and the bottom of the current is the second peak. Control is made to coincide with the threshold signal Vth2. Since the voltage division ratio between the first threshold signal Vth1 and the second threshold signal Vth2 is larger than that when the ratio is constant, the current amplitude of the inductor 4 is increased, and the switching frequency is suppressed. Switching loss can be reduced.

FIG. 10 is a block diagram showing another specific example of the variable resistor of FIG. In the figure, the same parts as those in FIG. 9 differs from FIG. 9 in that a resistor 14M connected to a resistor 14A and a switch 14N are connected in series, and the other end of the switch 14N is connected to a reference potential 14C. The resistor 14L is connected between the resistor 14A and the reference potential 14C. The switch 14N is connected to the determination circuit 14P. The determination circuit 14P compares the output of the peak voltage detection circuit 14K of the input voltage Vin with a predetermined voltage V1, and outputs an on / off signal to the switch 14N. It is configured.

With such a configuration, when the peak voltage of the input voltage Vin is large, the voltage dividing ratio between the first threshold signal Vth1 and the second threshold signal Vth2 is increased by turning on the switch 14N. When the peak voltage of the input voltage Vin is small, the voltage dividing ratio between the first threshold signal Vth1 and the second threshold signal Vth2 is reduced by turning off the switch 14N, and the voltage is divided according to the input voltage. By changing the pressure ratio so that the amplitude of the coil (inductor) current becomes substantially constant and suppressing the frequency, noise and switching loss can be reduced.

In the above, the example of changing the value of the variable resistor 14H by controlling the transistor or the switch based on the input voltage Vin has been described. However, the peak value of the current flowing through the inductor or the command value thereof is described. It is also possible to change the value of the variable resistor 14H by controlling the transistor or the switch based on the peak value of the certain first threshold signal Vth1 or the output voltage error.
In addition, it is possible to connect a plurality of resistors and transistors in the variable resistor 14H or a plurality of resistors and switches in parallel so as to cope with a large number of input voltage conditions and load conditions.

Configuration diagram showing an embodiment of the present invention FIG. 1 is an explanatory diagram of the operation. The block diagram which shows another embodiment of this invention Operational explanatory diagram of FIG. The block diagram which shows the 1st modification of FIG. The block diagram which shows the 2nd modification of FIG. The block diagram which shows another embodiment of this invention Operation explanatory diagram of FIG. Configuration diagram showing a specific example of the variable resistor of FIG. The block diagram which shows another specific example of the variable resistance of FIG. Configuration diagram showing a conventional example First operation explanatory diagram of FIG. Second operation explanatory diagram of FIG. Third operation explanatory diagram of FIG.

DESCRIPTION OF SYMBOLS 1 ... AC power source, 2 ... Rectifier circuit, 3, 6 ... Capacitor, 4 ... Inductor, 5, 14G ... Diode, 7 ... Switching element, 8 ... Voltage error amplifier, 9 ... Multiplier 10, 10A, 10B ... Comparator , 11 ... Monostable multivibrator, 12 ... Current detection resistor (current detection circuit), 13 ... Drive circuit, 14A, 14B, 14J, 14L, 14M ... Resistance, 14C ... Reference potential, 14D ... Transistor (FET), 14E ... Transistor drive circuit, 14F ... Control signal (selection signal), 14H ... Variable resistor, 14K ... Peak voltage detection circuit, 14N ... Switch, 14P ... Determination circuit, 15 ... Flip-flop.

Claims (7)

A rectifier circuit that obtains pulsating output by full-wave rectification of the AC power supply
A chopper circuit comprising a switching element connected to the rectifier circuit for turning on and off the current flowing through the inductor, and a capacitor for smoothing the current supplied from the inductor to obtain a DC output;
First signal generating means for generating a first threshold signal having the same phase as the input voltage of the chopper circuit and having a waveform similar in shape and having an amplitude proportional to the error of the output voltage of the chopper circuit;
A series circuit of a first resistor and a variable resistor formed by connecting a transistor in series or in parallel to at least one set of second resistors is connected between the first threshold signal and the reference potential, The transistor is driven by at least one signal of an error of an input voltage, a current flowing through the inductor, or an output voltage, and the first threshold signal is output from a connection point between the first resistor and a variable resistor. Second signal generating means for generating a second threshold signal in which the difference between them is substantially constant ;
Current detecting means for detecting a current flowing through the inductor; and when the detected current value reaches a level of the first threshold signal, the switching element is turned off and the current flowing through the inductor is the second current. And a switching control means for turning on the switching element when the level drops below a threshold signal level. The variable resistor of the second signal generating means is configured by a circuit in which at least one pair of resistors and switches are connected in series, or a circuit in which a plurality of series circuits of resistors and switches are connected in parallel, and the switches 2. The switching power supply according to claim 1 , wherein the switching power supply is turned on / off by at least one of an input voltage, a current flowing through the inductor, or an error of an output voltage . A rectifier circuit that obtains pulsating output by full-wave rectification of the AC power supply
A switching element connected to the rectifier circuit for turning on and off the current flowing through the inductor;
A chopper circuit comprising a capacitor that obtains a DC output by smoothing the current supplied from the inductor;
First signal generating means for generating a first threshold signal having the same phase as the input voltage of the chopper circuit and having a waveform similar in shape and having an amplitude proportional to the error of the output voltage of the chopper circuit;
Second signal generating means for generating a second threshold signal proportional to the first threshold signal;
Modulating means for modulating said second threshold signal,
Current detecting means for detecting a current flowing through the inductor;
When the detection current value of the current detection means reaches the level of the first threshold signal, the switching element is turned off, and the current flowing through the inductor is lower than the level of the second threshold signal. And a switching control means for turning on the switching element. The second signal generating means is formed of a series circuit of first and third resistors connected between the first threshold signal and a reference potential, and the first and third resistors are connected. The switching power supply according to claim 3 , wherein an output of a point is used as the second threshold signal. The modulation means includes a transistor connected between the second threshold signal and a reference potential, and a drive circuit that drives the transistor based on the first threshold signal. The switching power supply according to claim 3 or 4 . The modulating means includes a transistor connected between the second threshold signal and a reference potential, and the transistor based on a difference between the first threshold signal and the second threshold signal. 5. The switching power supply according to claim 3 , wherein the switching power supply comprises: The modulation means includes a transistor connected between the second threshold signal and a reference potential, and a drive circuit for driving the transistor based on a difference between the first threshold signal and a control signal. The switching power supply according to claim 3 or 4 , characterized by comprising :

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