JPH09191699A - Control method of induction motor - Google Patents

Abstract

(57) Abstract: In an induction motor speed control device, the influence of fluctuations in the motor constant is compensated. A current detector (4) detects a d-axis component and a q-axis component of a primary current of an induction motor (2), identifies a motor constant according to a variation of a detected d-axis component from a reference value, and identifies this. The output voltage of the inverter 1 is modified based on the value. [Effect] The fluctuation amount of the primary resistance and the fluctuation amount of the reactance can be correctly identified, whereby the fluctuation amount of the d-axis voltage and the fluctuation amount of the q-axis voltage can be accurately compensated. . Therefore, there is no inconvenience such as fluctuation of the induced electromotive force of the induction motor 2, and the speed of the induction motor 2 can be precisely controlled.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an induction motor speed control method, and more particularly to an induction motor control method suitable for high-performance speed control.

[0002]

2. Description of the Related Art As a conventional technique relating to a speed control method for an induction motor by vector control without using a speed sensor, it is described, for example, in IEEJ Transactions, Vol. 107, No. 2, pp. 191 to 198 (sho 62). There is something.

[0003]

In the above-mentioned prior art, there is a problem that the control characteristics of speed and torque are deteriorated mainly in low frequency operation due to the fluctuation of the motor constant, especially the fluctuation of the primary resistance of the induction motor. .

An object of the present invention is to provide a control method for an induction motor which can compensate the influence of fluctuations in the motor constant.

[0005]

The first aspect of the present invention for achieving the above object is a method for controlling the speed of an induction motor by an inverter whose output voltage is controlled according to a voltage command signal. The d-axis component (corresponding to the exciting current) and the q-axis component (corresponding to the torque current) of the primary current are detected, the motor constant is identified according to the variation from the reference value of the d-axis component, and based on this identified value To correct the output voltage of the inverter.

[0006]

BEST MODE FOR CARRYING OUT THE INVENTION As described above, the control characteristic is deteriorated by the fluctuation of the electric motor constant, especially the fluctuation of the primary resistance due to the temperature change of the induction motor (hereinafter, abbreviated as an electric motor). In particular, in low frequency operation, the ratio of the voltage drop due to the fluctuation of the primary resistance with respect to the induced electromotive force of the induction motor increases, and the control characteristics are greatly affected.

Therefore, by compensating for the influence of the fluctuation of the motor characteristics, it becomes possible to control the speed and torque with higher accuracy over a wide speed range. Specifically, the d-axis component (exciting current component) and q-axis component (torque current component) of the primary current of the induction motor are detected, and the primary resistance of the induction motor varies depending on the variation from the reference value. It suffices to identify the voltage drop due to the fluctuation of the voltage drop and the leakage reactance, and correct the magnitude and phase of the output voltage of the frequency conversion device by this.

That is, the output voltage of the frequency converter for driving and controlling the induction motor is corrected based on the identification signal of the voltage drop due to the fluctuation of the primary resistance of the induction motor and the voltage drop due to the fluctuation of the leakage reactance. It is possible to compensate for the influence of variations in the primary resistance and the leakage inductance, and it is possible to perform highly accurate motor control.

Embodiments of the present invention will be described in detail below with reference to the drawings.

(Embodiment 1) A control apparatus for an induction motor according to a preferred embodiment of the present invention will be described with reference to FIGS. 1 and 2.

In FIG. 1, a voltage type PWM control inverter 1 drives and controls an induction motor 2 to generate a voltage command V _u *,
The instantaneous value of the fundamental wave component of the output voltage is controlled in proportion to V _v * and V _w *. The phase angle calculator (integrator) 3 is
The frequency command ω ₁ ** is integrated, the phase angle θ * of the rotating magnetic field coordinate is calculated and output, and the current detector (coordinate converter) 4 calculates the induction motor currents i _u , _iv , and i _w with the phase angle θ *. Using,
Exciting current components i _1d and q that are d-axis components of the rotating magnetic field coordinate amount
It is converted into a current component i _1q which is an axial component. The slip calculator 50 performs a calculation for estimating the slip frequency ω _s of the induction motor from the above-described torque current component i _1q , and the subtractor 206
Is the estimated value ω of the rotational speed of the induction motor from the frequency command ω ₁ ** and the estimated slip frequency ω _s of the induction motor.
_{Find r} . A speed controller (ASR) 207 amplifies the deviation between the speed command ω _r * and the estimated rotation speed ω _r, and outputs a q-axis current command, that is, a torque current command i _1q *.
The current regulator (ACR) 208 uses this current command i _1q *
And the deviation from the torque current component i _1q is amplified and the frequency command ω ₁ * is output. The differentiator 219 calculates the torque current component i
Differentiation of _1q and subtraction of its output from the frequency command ω ₁ * prevent fluctuations in the induction motor magnetic flux during transients. The coefficient units 210 to 212 are for setting the resistance and the inductance value of the induction motor 2, and the coefficient unit 21
An amount proportional to the primary resistance drop (reference value) is output from 0 and 212, and an amount proportional to the leak reactance drop (reference value) is output from the coefficient unit 211. Also, multiplier 2
13 outputs the value obtained by multiplying the output of the coefficient unit 211 by the frequency command ω ₁ *, and the multiplier 214 outputs the induced electromotive force command value. The identifier 215 identifies the voltage drop due to the fluctuation of the primary resistance and the voltage drop due to the fluctuation of the leakage reactance. The adder 216 adds the output signals of the coefficient unit 210, the multiplier 213, and the identifier 215, and outputs the d-axis voltage command V _1d *, and the adder 217 outputs the coefficient unit 212 and the multiplier 214.
And the output signals of the identifier 215 are added to output the q-axis voltage command V _1q *. Three-phase voltage command calculator (coordinate converter)
7 calculates a stator coordinate amount using the d-axis voltage command V _1d *, the q-axis voltage command V _1q * and the phase angle θ *, and calculates the three-phase voltage commands V _u *, V _v *, V _w. Output *. Identifier 21
In FIG. 2 showing the configuration of No. 5, the subtractor 151 outputs the difference between the command value i _1d * of the exciting current and the detected value i _1d . The state determiner 152 outputs an “H” signal when the frequency command ω ₁ * is less than or equal to a predetermined value or when the q-axis current i _1q is less than or equal to a predetermined value, and controls the switch 153 to be “closed”. When the frequency command ω ₁ * is greater than or equal to the predetermined value and the q-axis current i _1q is greater than or equal to the predetermined value, the “L” signal is output and the switch 154 is controlled to be “closed”. When the switch 153 is “closed”, the integrator 155 integrates the output of the subtractor 151 and outputs a signal corresponding to the variation Δr ₁ of the primary resistance.
When 53 is “open”, the signal Δr ₁ calculated while the switch 153 is “closed” is held. The integrator 156 integrates the polarity inversion signal of the output of the subtractor 151 when the switch 154 is “closed”, and corresponds to the variation Δ (l ₁ + l ₂ ′) of the leakage inductance of the winding of the induction motor 2. When the switch 154 is “open”, the signal calculated while the switch 154 is “closed” is held. The multipliers 157 and 158 multiply the variation Δr _{1 of the} primary resistance and the exciting current component i _1d or the torque current component i _1q , respectively, and calculate the voltage drop due to the variation of the d-axis and q-axis resistances. In addition, the multipliers 160 and 161 have a variation Δ (l ₁ +) of the leakage inductance which is the output of the integrator 156.
l ₂ ′) and the frequency command ω ₁ * multiplied by the multiplier 159, Δ (l ₁ + l ₂ ) ω ₁ *, and the exciting current component i
_1d or torque current component i _1q and
The voltage drop due to the fluctuation of the reactances of the d-axis and the q-axis is calculated. The adders 162 and 163 add the above-mentioned voltage drops and output an identification signal corresponding to the fluctuation of the leakage impedance drop.

Next, the operation of the above-described embodiment will be described. Inverter output voltage command value V for the coordinate converter 7
_1d * and _V1q * are calculated according to the following equations by the control configuration described above.

[0013]

[Equation 1]

In (Equation 1), r ₁ * i ₁ , ω ₁ * (l ₁
* + L ₂ ′ *) i ₁ is the estimated value of the voltage drop due to the fluctuation of the primary resistance and the voltage drop due to the fluctuation of the leakage reactance, and ω ₁
* Φ _1d is the command value of the induced electromotive force, ΔV _d , ΔV _q are the output values of the identifier 215, the set values of the coefficient units 210 to 212 match the electric constants of the induction motor 2, and φ _1d * =
If φ _1d * and i _1d * are set in the relation of (M + l ₁ ) i _1d *, the output values ΔV _d and ΔV _q become zero in the steady state.

The output voltage command values V _1d *, V _1q * calculated as described above are converted into three-phase voltage commands V _u *, V _v *, V _w * in the coordinate converter 7. Since the three-phase voltage command has different phases by 120 degrees, the U-phase command _Vu * can be calculated by the following (Equation 2) as a representative.

[0016]

[Equation 2]

In the voltage type PWM inverter 1, the output voltage of each phase is controlled according to the pulse width modulation signal obtained by comparing the sine wave voltage commands V _u *, V _v *, V _w * with the carrier signal. Therefore, the instantaneous value of the fundamental wave component is controlled in proportion to these voltage commands. As a result, the output voltage vector of the inverter 1 is controlled according to the d-axis and q-axis output voltage command values V _1d *, V _1q * and the phase angle θ *. At this time, if the leakage impedance drop estimated using the coefficient units 210 to 212 and the like match the actual value, the magnitude of the induced electromotive force of the induction motor 2 is:
This command value (= ω ₁ * φ _1d *) matches. Under this condition, the phase angle θ * output from the phase angle calculator (integrator) 3 represents the rotation angle θ of the electric motor magnetic flux vector from the stator U-phase axis. At this time, in the coordinate converter 4, the d-axis current component i _1d and the q-axis current component i _1q which are calculated and detected from the induction motor currents i _u , _iv , and i _{w according} to (Equation 3).
Are proportional to the exciting current and the torque current of the induction motor 2, respectively.

[0018]

(Equation 3)

As described above, since the q-axis current component i _1q is proportional to the induction motor torque current, the slip calculator 50
Is the slip frequency ω _{s of the} induction motor based on i _1q.
Further, by subtracting the estimated slip frequency ω _s from the inverter frequency command ω ₁ ** in the subtractor 206, an estimate value ω _r of the rotational speed of the induction motor can be estimated. You can The speed adjuster 207 outputs the current command i _1q * in accordance with the speed deviation value (ω _r * −ω _r ) between the speed command value ω _r * and the estimated value ω _r , and the current adjuster 208 further outputs the current command i _1q *. Current command i _1q *
The frequency command ω ₁ * is output according to the current deviation (i _1q * -i _1q ) between the q-axis current component i _1q and the q-axis current component i _1q . The slip frequency of the induction motor 2 is controlled according to the change of the frequency command ω ₁ *, and the q-axis, that is, the torque current component i _1q , which is the output of the coordinate converter 4, is the current command i _1q described above. Controlled to match *. Since the torque of the induction motor is proportional to the torque current component i _1q , the torque can be controlled according to the speed deviation by the control as described above.
The rotation speed can be controlled so as to match the speed command value.

The above-described operation is the basic operation of the embodiment shown in FIG. 1 of the present invention. Next, the operation when the motor constant fluctuates will be described.

The values set in the coefficient units 210 to 212 are
If the actual value of the motor constant does not match due to temperature fluctuations inside the induction motor, etc., the estimated voltage drop due to the fluctuation of the primary resistance of the induction motor and the voltage drop due to the fluctuation of the leakage inductance are calculated based on this. , It will not match the actual value. As a result, the actual value of the induced electromotive force does not match the command value. At this time, since the induction motor magnetic flux fluctuates from the reference value and the gain of ω _s / i _1q fluctuates, an error occurs in the slip frequency ω _s estimated by the slip calculator 50, and as a result, the estimated rotation speed value ω
_An error also occurs in _r, and the speed accuracy of the induction motor deteriorates. Further, the value of torque / i _1q also fluctuates and decreases due to the fluctuation of the magnetic flux. This tendency becomes remarkable especially when the operating frequency is low and the influence of the voltage drop due to the fluctuation of the primary resistance becomes large.

In the present embodiment shown in FIG. 1, the influence as described above can be prevented as follows.

That is, the output voltage of the inverter 1 is controlled in accordance with the above-mentioned (Equation 1).
Induction motor voltage is shown below (equation 4) in steady state.
Can be indicated by

[0024]

(Equation 4)

However, φ _1d = (M + l ₁ ) i _1d .

The output voltage of the inverter 1 is
If the output voltage saturation and the non-linear distortion are ignored, the induction motor voltage will be the same, and the d
Axial variation [Delta] V _q of (exciting current component) variation [Delta] V _d and q-axis voltages (secondary current component) voltage is represented by the following equation (5).

[0027]

(Equation 5)

However, Δr ₁ = r ₁ −r ₁ *, Δ (l ₁ +
l ₂ ′) = (l ₁ + l ₂ ′) − (l ₁ * + l ₂ ′ *).

That is, in relation to the fluctuations Δr ₁ and Δ (l ₁ + l ₂ ′) of r ₁ and l ₁ + l ₂ ′ from the reference value, ΔV _{d with} respect to the voltage commands V _1d * and V _1q * , ΔV _q must be compensated. Therefore, the embodiment of the present invention shown in FIG.
15 and using the identifier 215, the above-mentioned variation Δ
V _d , ΔV _q are identified, and the result is the voltage command V _1d *, V _1q *
To give compensation. The operation of the identifier 215 will be described next.

If the variation ΔV _d described above is not compensated, the output i _1d of the coordinate converter 4 varies from the reference value, and conversely, ΔV _d can be estimated from the variation. However, as shown in (Equation 5), the fluctuations have mixed effects of Δr ₁ and Δ (l ₁ + l ₂ ′), and
Since the polarities related to ΔV _d and ΔV _q are different, the voltage drop due to the resistance change and the voltage drop due to the reactance change must be identified separately.

As can be understood from (Equation 5), when ω ₁ or i _1q is small, Δ (l ₁ +
The influence of l ₂ ′) is small, and Δr ₁ is dominant. Therefore,
Under such a condition, the fluctuation of i _1d can be considered to be caused by Δr ₁ . On the contrary, when ω ₁ is large and i _1q is large, the influence of Δ (l ₁ + l ₂ ′) is dominant, and under such conditions, the variation of i _1d is Δ (l ₁ + l ₂ ′).

Therefore, the identifier 215 is provided with a state discriminator 152 for discriminating the above-mentioned conditions. Under the former condition, the state discriminator 152 closes the switch 153, and the identifier 215 thereby obtains the resistance obtained by multiplying the output of the integrator 155 by the exciting current component i _1d and the torque current component i _1q. The d-axis voltage command V _1d * and the q-axis voltage command V _1q * are corrected by using the identification signal of the voltage drop due to the fluctuation of the voltage fluctuations ΔV _d and ΔV _q . At this time, the output of the integrator 155 is the variation Δr _{1 of the} primary resistance.
Is equivalent to. Next, under the latter condition, the identifier 215 determines that the switch 154 is the state determiner 15
The output of the integrator 156 is multiplied by ω ₁ *, and further, i _1d and i _1q are multiplied, and the identification signal of the voltage drop due to the fluctuation of the reactance is used to obtain ΔV _d ,
Compensate for ΔV _q . At this time, the output of the integrator 156 is
It corresponds to the variation Δ (l ₁ + l ₂ ′) of the inductance. In the above, the switch 153 is “open”
In the case of, the output of the integrator 155 holds the value immediately before the switch 153 becomes “open”, and the output of the switch 154
Is “open”, the output of the integrator 156 is the switch 15
It holds the value immediately before 4 becomes "open".

In the present embodiment, the variation Δr _{1 of} the primary resistance and the variation Δ (l ₁ + l ₂ ′) of the reactance can be correctly identified while the above-mentioned conditions are alternately changed. , D-axis voltage variation ΔV _d , q-axis voltage variation Δ
The V _q can be compensated with high accuracy, and the induction motor 2 can be precisely speed-controlled without inconvenience such as fluctuation of the induced electromotive force (magnetic flux) of the induction motor 2.

Japanese Unexamined Patent Publication (Kokai) No. 62-126894 discloses that the voltage drop due to impedance is calculated and the output voltage of the inverter is determined based on the calculated voltage drop. However, in the above calculation, the voltage drop is calculated using the resistance and the inductance, which are preset values, and therefore the voltage drop that occurs when the resistance and the inductance themselves change cannot be compensated. On the other hand, in the present embodiment, as described above, the variation of the resistance and the inductance itself is obtained, and the variation of the voltage due to the variation is compensated.

(Embodiment 2) A control device for an induction motor according to another embodiment of the present invention will be described with reference to FIGS. 5 is a frequency controller, 21 is a slip calculator, 22 is a differentiator, 6 is a voltage command calculator, 25, 30, 36 are coefficient multipliers, 23, 26, 32, 38 are adders, 27, 31, 3
Reference numerals 4 and 37 are multipliers, 28 is a coefficient unit, 29, 33 and 35 are function units, and other reference numerals are the same as those in FIGS. 1 and 2.

In the embodiment shown in FIG. 3, the frequency controller 5 controls the speed command ω _r *, the output of the slip calculator 21,
The adder 23 adds the output of the differentiator 22 for stable control of the torque current component i _1q and outputs the frequency command ω ₁ *. The voltage command calculation unit 6 uses the frequency command ω ₁ * and the torque current component i _1q from the frequency control unit 5 to output the inverter output voltage, that is, the absolute value command V _1a * of the induction motor voltage and the induction motor internal phase difference. The angle command δ * is calculated and output.

It should be noted that, although the description of the identifier is omitted in FIG. 3, the embodiment shown in FIG. 3 is also provided with an identifier, and the output thereof is the multipliers 27, 31 ,. Three
7 and coefficient unit 28. The identifier used in this embodiment only needs to be able to identify r ₁ and (l ₁ + l ₂ ′), and does not need to calculate the voltage drop due to the resistance variation and the voltage drop due to the reactance variation. 157 to 161 are unnecessary, and the configuration shown in FIG. 4 can be obtained.

The operation of the embodiment shown in FIG. 3 is the same as that of the embodiment shown in FIG. 1 in terms of the voltage command calculator related to the present invention. That is, this voltage command calculation unit is a conversion of the orthogonal form of the operation form of the embodiment shown in FIG. 1 into a polar form and is equivalent. Therefore, if the constants r ₁ and (l ₁ + l ₂ ′) used in this calculation are corrected by using the identifier as shown in FIG. 4, the same effect as in the embodiment shown in FIG. Can be obtained.

Since the primary resistance r ₁ and the secondary resistance r ₂ of the induction motor are generally close to each other in the primary and secondary windings, their temperature rise and the resulting resistance change are
There is a substantially proportional relationship. Therefore, using Δr ₁ identified as described above, Δr ₂ can be estimated by the following equation.

[0040]

(Equation 6)

In (Equation 6), r ₁ * and r ₂ * are reference values of the primary and secondary resistances. By correcting the gain (proportional to r ₂ ) of the slip calculator in the above-described embodiment using Δr ₂ , it is possible to prevent the occurrence of slip frequency estimation error and speed control error due to secondary resistance change.

Further, when the variation of the primary resistance, which is the output of the identifier, exceeds a predetermined value, it can be determined that an abnormality such as overheating or disconnection of the induction motor has occurred, thereby protecting the induction motor. Can be done.

(Embodiment 3) A method of controlling an induction motor according to another embodiment of the present invention will be described with reference to FIG. 1
4 are the same as those in the above-mentioned embodiment. Frequency control unit 5 '
Is the output of the speed command ω _r * and the slip calculator 21, and the output of the differentiator 22 ′ for stable control of the current i _1q , and
The adder 23 adds the output of the current limiter 80 that operates during overload and prevents overcurrent of the induction motor, and outputs the frequency command ω ₁ *. The voltage command calculation unit 6 ′ is ω ₁ *
, I _1q, etc., the absolute value command V _1a * of the output voltage of the inverter and the induction motor internal phase difference angle command δ * are calculated and output. An expression expressing the content of the calculation is as follows.

[0044]

(Equation 7)

However, r ₁ = r ₁ * + Δr ₁ l ₁ + l ₂ ′ = (l ₁ * + l ₂ ′ *) + Δ (l ₁ + l ₂ ′) The coefficient unit 25 has a secondary interlinkage magnetic flux φ _2d * and ω. The product of ₁ * is output, and the adder 26 outputs the output r ₁ i from the coefficient unit 61.
_1q and the output ω ₁ * (l ₁ + l ₂ ′) i _1d from the multiplier 62 are added to obtain the coefficient of cos δ * in the arithmetic expression of V ₁ *. On the other hand, the output r ₁ i _1d of the coefficient unit 64 and the output ω ₁ * (l ₁ + l ₂ ′) i _1q of the multiplier 63 are subtracted by the subtractor 65 to obtain V
The coefficient of sin δ * in the calculation formula of _1a * is obtained. In the divider 28, the output of the adder 26 and the output of the subtractor 65 are divided, and the function unit 29 outputs the internal phase difference angle command δ * from the result.

Next, the function unit 33 and the multiplier 34 make V
The first term on the right side of _1a * is calculated, and the second term is calculated by the function unit 35 and the multiplier 66. And the adder 38
Outputs V _1a *. Using the above V _1a * and δ *, the voltage command of each phase is calculated in the coordinate converter 4, and the output voltage of the inverter 1 is controlled.

By the way, the primary resistance r ₁ is greatly affected by the temperature inside the induction motor, which varies depending on the operating state of the induction motor, and its value is large and fluctuates. Due to this fluctuation, the output characteristic of the induction motor, for example, the value of torque / i _1q changes greatly. This tendency is remarkable especially when the operating frequency is low. Therefore, the influence of the primary resistance fluctuation is removed as follows. That is, the low frequency detector 7
In the case of 0, the frequency command ω ₁ * is input, and when the absolute value of the command value is less than or equal to the predetermined value, the operation signal is output, and when it is more than that value, the operation signal is not output. The switch 71 is in the “closed” state when the operation signal is input, and the subtractor 73
The output from (i _1d * -i _1d ) is sent to the constant identifier 72.
The constant identifier 72 determines the constant variation amount Δr based on the input signal.
Identifies ₁ but keeps the value if there is no input. The identified value is sent to the coefficient units 61 and 64, and the coefficient value is adjusted. The coefficient of the slip calculator 21 is proportional to the secondary resistance value, and in the induction motor, the primary resistance and the secondary resistance change in substantially the same manner. Therefore, the identification value is also sent to the slip calculator 21 and used for adjusting the coefficient value. At that time, the signal passes through the coefficient unit 75, where
The primary resistance variation Δr ₁ is converted into the secondary resistance variation Δr ₂ .

By the way, in the high speed operation region, the influence of the leak reactance change is larger than that of the primary resistance change. However, the adverse effect of the leakage reactance variation on the operating characteristics is not so large in the high-speed operating range, and may be practically used without causing a problem. Therefore, it is sufficient to correct the primary resistance only during low speed operation.

As described above, according to the embodiments shown in FIGS. 1 to 5, it is possible to compensate for the influence of the variation of the motor constant of the induction motor, and the variation of the induction starting force (magnetic flux) and the accompanying speed. It is possible to prevent deterioration of control accuracy and torque reduction.

[0050]

According to the present invention, the influence of the fluctuation of the motor constant of the induction motor can be compensated, and the fluctuation of the induced electromotive force (magnetic flux) and the accompanying deterioration of the speed control accuracy and the reduction of the torque can be prevented. can do.

[Brief description of the drawings]

FIG. 1 is a configuration diagram of a control device for an induction motor according to a preferred embodiment of the present invention.

FIG. 2 is a configuration diagram of an identifier 215 shown in FIG.

FIG. 3 is a configuration diagram of a control device for an induction motor that is another embodiment of the present invention.

FIG. 4 is a block diagram of an identifier that can be used in FIG.

FIG. 5 is a configuration diagram of a control device for an induction motor according to another embodiment of the present invention.

[Explanation of symbols]

DESCRIPTION OF SYMBOLS 1 ... Voltage type PWM inverter, 2 ... Induction motor, 3 ... Integrator, 4 ... Current detector, 5, 5 '... Frequency control part, 6,
6 '... voltage command calculation unit, 7 ... three-phase voltage command unit.

 ─────────────────────────────────────────────────── ─── Continuation of the front page (72) Inventor Takayuki Matsui 4026 Kuji Town, Hitachi City, Ibaraki Prefecture Hitate Manufacturing Co., Ltd.Hitachi Laboratory Ltd. In Hitachi Research Laboratory (72) Inventor Hiroshi Fujii 7-1-1 Higashi Narashino, Narashino, Chiba Prefecture Hitachi Narashino Factory

Claims (5)

[Claims] 1. A method for controlling the speed of an induction motor by an inverter whose output voltage is controlled according to a voltage command signal, wherein a d-axis component (corresponding to an exciting current) of a primary current of the induction motor.
And q-axis component (corresponding to torque current) are detected, the motor constant is identified according to the variation of the d-axis component from the reference value, and the output voltage of the inverter is corrected based on this identified value. Induction motor control method. 2. The motor constant is identified by identifying a primary resistance when at least one of a frequency command value and a q-axis current component is a predetermined value or less, and the frequency command value and the q-axis current component are both predetermined values. The method for controlling an induction motor according to claim 1, wherein the leakage inductance is identified in the above case. 3. The motor constant is identified by identifying a primary resistance according to a variation of a d-axis component of the primary current from a reference value when a frequency command value is a predetermined value or less, and based on the identified value. 2. The method of controlling an induction motor according to claim 1, wherein the output voltage of the inverter is corrected. 4. The estimated value of the secondary resistance is calculated based on the identification value of the primary resistance.
A method for controlling an induction motor according to item 1. 5. The method of controlling an induction motor according to claim 3, wherein an abnormality of the induction motor is determined based on the identification value of the primary resistance.