JPH0131836B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION Field of the Invention The present invention relates to a television synchronization receiver that can be used in television receivers and VTR video tuners.

Conventional configurations and their problems In recent years, so-called electronic tuners that use variable capacitance diodes as tuning elements have been widely used in television receivers and VTR video tuners. Electronic tuners have advantages such as being non-contact, so there is no problem of contact failure, and being electronically controllable, making it convenient for multi-functions such as remote control. However, due to variations in the characteristics of variable capacitance diodes and the need for an inductor for tuning, it is difficult to manufacture them without adjustment or to automate them.

Therefore, in order to configure a receiver that is easy to integrate without using a tuning circuit using a variable capacitance diode and an inductor, it is possible to use a synchronous reception method. Although there are various synchronous reception methods, a synchronous carrier regeneration method is suitable for phase-synchronizing a synchronous carrier wave with a weak television signal. This method is called Costas loop.
known as the method.

FIG. 1 is a main part block diagram showing the configuration of a conventional synchronous carrier regeneration type synchronous receiver using a Costas loop. 1 is a first synchronous detector that synchronously detects the in-phase component of a modulated carrier input; 2 is a second synchronous detector that synchronously detects a quadrature component; 3 and 4 are each of these two synchronous detectors 1 and 2; A low-pass filter 5 low-pass filters the output, and 5 is a low-pass filter 3 for these two low-pass filters.
and a phase detector that detects the phase of the synchronized carrier wave with respect to the modulated carrier wave by multiplying the output of 4 by voltage, 6 is a low-pass filter that low-pass filters the output of this phase detector 5, and 7 is this low-pass filter. A voltage controlled oscillator 6 is controlled by the output thereof, and 8 is a 90° phase shifter which shifts the phase of the output of the voltage controlled oscillator 7 by 90°.

In this Costas loop type synchronous receiver, the first
The in-phase and quadrature component signals obtained from the second synchronous detectors 1 and 2 are applied to the phase detector 5, and from this phase detector 5, the receiver input, that is, the modulated carrier wave, and the output of the voltage controlled oscillator 7, that is, the synchronous By obtaining a voltage proportional to the phase error with the carrier wave and feeding this voltage back to the voltage controlled oscillator 7, the phase error is controlled to be zero.

If the conventional example shown in Fig. 1 is applied as is to a television receiver, the baseband video signal of the desired channel to be received can be obtained by synchronous detection, and the audio intermediate frequency signal can also be obtained. A carrier color signal and a carrier audio signal are generated. The carrier color signal and carrier audio signal of the lower adjacent channel are mixed into the synchronously detected baseband video signal as an interference signal.

As a countermeasure, it is possible to remove the lower adjacent channel by installing a tuning circuit using a variable capacitance diode and an inductor in the high frequency input section.
This deviates from the original purpose of constructing a receiver without using these elements.

Therefore, the present inventor has proposed a television synchronization receiver that can eliminate the interference of the carrier color signal and carrier audio signal of the lower adjacent channel to the baseband video signal of the channel desired to be received. This television synchronous receiver includes a voltage controlled oscillator, a 90° phase shifter that shifts the phase of the output of the voltage controlled oscillator by 90°, and an output of the voltage controlled oscillator and an output of the 90° phase shifter. first and second synchronous detectors that synchronously detect in-phase and quadrature components of a video carrier signal using the respective synchronous carrier waves, and outputs of the first and second synchronous detectors as a video signal base. first and second low pass filters for low pass filtering in the frequency range of the audio intermediate frequency signal and the video carrier signal by multiplying the outputs of the first and second low pass filters; a phase detector for detecting a phase difference between the outputs of the voltage controlled oscillator; means for feeding back the output of the phase detector to the voltage controlled oscillator; and a signal amplifier for amplifying the output of the first low pass filter;
An A/D converter converts the baseband video signal in the output of this signal amplifier from analog to digital, and a television synchronization signal or a color burst signal is separated from the output of the same signal amplifier. A clock generator that generates a clock signal by controlling it, and a time-direction (interframe) low-frequency circuit that receives the signal output from the A/D converter as input and operates using the clock signal output from the clock generator as a clock. The first low-pass filter is configured to include a filter and a D/A converter that converts the output of the temporal low-pass filter from digital to analog. The main part of the energy of the video signal is removed, and the temporal low-pass filter reduces the interference of the carrier color signal and audio intermediate frequency signal of the lower adjacent channel with the baseband video signal. .

The previously proposed television synchronization receiver will be explained with reference to FIGS. 2 to 9. FIG. 2 shows a block diagram of the main parts of the television synchronization receiver. In FIG. 2, 9 is a high frequency input section, 10 is a first synchronous detector, 11 is a second synchronous detector, 12 and 13 are first and second low pass filters, and 14 and 15 are signal amplifiers. , 16 and 17 are third and fourth low-pass filters, 18 is a phase detector, 19 is a Costas loop low-pass filter, 20 is a voltage controlled oscillator, 2
1 is a 90° phase shifter, and these constitute a Costas loop. 22 is an audio intermediate frequency amplifier;
3 is a frequency discriminator, 24 is a voltage subtractor, and 25 is a low-pass filter, these constitute a frequency pull-in circuit, and the output thereof is added to the output of the low-pass filter 19 of the Costas loop by a voltage adder 26. be done. 2
7 is a voltage storage device, 28 is a voltage selector, and 29 is a control input device, which constitute a channel selection voltage generation circuit. The output voltage of the voltage selector 28 is also summed in a voltage adder 26 with the output of the low-pass filter 19 of the Costas loop. 30 is an A/D converter that converts the output of the signal amplifier 14 from analog to digital;
Reference numeral 31 denotes a clock generator which separates a television synchronization signal or a color burst signal from the output of the signal amplifier 14 and generates a clock signal based on either signal. 32 is above A/
An arithmetic unit that adds a signal obtained by multiplying the output of the D converter 30 by 1-K and a signal obtained by multiplying the output of a frame memory 33 (to be described later) by K, and 33 is a frame memory that stores the output of this arithmetic unit 32 for each frame. ,3
4 is a chroma inverter that inverts the phase of the color signal of the color television signal stored in the frame memory 33 for each frame; 35 is the output of this chroma inverter 34 and the A/D converter 3;
36 is a coefficient generator that determines the coefficient K based on the output of the motion detector 35; 38 is a memory controller that writes and erases the frame memory 33 according to the address determined by the address generator 37. Construct an adaptive temporal low-pass filter. 39 is a video signal filter, 4
0 is a D/A converter for digital-to-analog conversion of the output of the video signal filter, 41 is a video output circuit, and 42 is an audio output circuit.

The operation of the television synchronous receiver shown in FIG. 2 constructed as described above will be explained below. It is assumed that the video carrier signal of the channel to be received, which is input to the high frequency input unit 9, is υ _V (t), and the audio carrier signal is υ _S (t). Since υ _V (t) is subjected to residual sideband modulation, it can be expressed as follows.

υ _V (t)=Re{[I(t)+jQ(t)]expj[ω _V t+
_V ]}=I(t)cos( _ωVt + _V )−Q(t)sin(
ω _V t+ _V (1) Here, Re is the real part of the expression in { }.
I(t) is a signal having an in-phase component with respect to the carrier wave, and includes a video signal. Q(t) is a signal of orthogonal component to the carrier wave, ω _V is the angular frequency of the video carrier wave, _V
is the phase of the video carrier.

Furthermore, the narrowband Gaussian noise n(t) is set as n(t)=n _C (t) cos(ω _V t+ _V ) −n _S (t) sin(ω _V t+ _V ) ……(2), and the above υ _V (t) and n(t) are applied to one terminal of each of the first and second synchronous detectors 10 and 11.

Now, let the output of the voltage controlled oscillator 20 be υ ₀ (t)=A ₀ cos (ω ₀ t+ ₀ )...(3), and apply this to the other terminal of the first synchronous detector 10 consisting of a voltage multiplier. Then, its output υ _PV
(t) is υ _PV (t)=A ₀ [υ _V (t)+n(t)]cos(ω ₀ t+
₀ ) = A ₀ /2 [I (t) + n _C (t)] {cos [(ω _V +
ω ₀ )t+ _V + ₀ )+cos[( _ωV − _ω0 )t+ _V − ₀
]} −A ₀ /2[Q(t)+n _S (t)]{sin[(ω _V
+ω ₀ )t+ _V + ₀ ] +sin [(ω _V −ω ₀ )t+ _V − ₀ ]}...(4) When the voltage-controlled oscillator output is synchronized with the video carrier wave, ω ₀ =ω _V , so υ _PV (t) = A ₀ /2 [I (t) + n _C (t)] {cos (2ω
_Vt + _V + ₀ )+cos( _V - ₀ )}- _A0 /2[Q(t
) +n _S (t)〕{sin(2ω _V t+ _V + ₀ )+sin
( _V − ₀ )}...(5) When the 2ω _V signal is removed by the low-pass filter 12, υ _PV (t) = A ₀ /2 [I (t) + n _C (t)] cos −A ₀ / 2[Q(t)+n _S (t)] sin ...(6) Here, is _V − ₀ , which is the phase difference between the video carrier wave and the voltage-controlled oscillator output. If = 0, υ _PV (t) = A ₀ /2 [I (t) + n _C (t)] ...(7). In other words, the signal and noise of the in-phase component with respect to the video carrier wave are obtained as the detection output. However, orthogonal components are not detected. This detection output is passed through a low-pass filter 12 to a signal amplifier 14 as a video detection output.
The signal is amplified by the D/A converter 40 and output to the temporal low-pass filter. The filtering characteristics of the low-pass filter 12 are shown in FIG. The video signal is filtered at baseband as shown in FIG. When a television signal is received using the conventional superheterodyne reception method, the overall baseband frequency characteristic can be considered to be flat due to the Nyquist filtering characteristics of the intermediate frequency amplifier. In this method, it must be assumed that the system is as shown in Figure 4a. That is, the voltage gain in the low frequency range is twice the gain in the high frequency range.
Therefore, in the embodiment shown in FIG. 2, the frequency characteristics of the signal amplifier 14 are corrected as shown in FIG. 4b.

Since the audio carrier signal υ _S (t) of television broadcasting is frequency modulated, υ _S (t) = A _S cos [{ω _S + s (t)} t + _S ]...(8
). where A _S is the amplitude of the audio carrier signal,
ω _S is the angular frequency of the audio carrier signal, s(t) is the audio signal, and _S is the phase of the audio carrier signal.

This υ _S (t) and υ ₀ (t) of equation (3) are calculated by the synchronous detector 10
, the output is υ _PS (t)=A _S cos [{ω _S +s(t)}t+ _S ]A ₀ cos
(ω ₀ t+ ₀ )=A _S A ₀ /2cos [(ω _S +ω ₀ )t+s(t
)t + _S + ₀ ] = A _S A ₀ /2cos [(ω _S −ω ₀ ) t
+s(t)t+ _S − ₀ ]...(9) When the frequency component of ω _S +ω ₀ is removed by the low-pass filter 12, υ _PS (t)=A _S A ₀ /2cos [(ω _S −ω ₀ )t +s(t)t+ _S − ₀ ] ...(10) If ω _IF =ω _S −ω ₀・ω ₀ =ω _V , then υ _PS (t)=A _S A ₀ /2cos [｛ω _IF +s (t)}t+ _S
− ₀ ] ...(11) υ _PS (t) in equation (10) is nothing but the audio carrier signal shown in equation (8) converted into an audio intermediate frequency signal with an angular frequency ω _IF .

The filtering characteristics of the low-pass filter 12 are designed to cover the frequency ω _IF of the audio intermediate frequency signal, as shown in FIG. The audio intermediate frequency signal passes through this low-pass filter 12 and is amplified by a signal amplifier 14 and an audio intermediate frequency amplifier 22. The output is demodulated by a frequency discriminator 23 to obtain an audio signal s(t). s(t) is supplied to the audio output circuit 42.

The carrier television signal is made up of signals having a frequency relationship as shown in FIG. 5a. The desired channel to receive is shown on the right, and the lower adjacent channel is shown on the left. The television signal of the desired channel is synchronously detected by the synchronous detector 10, and a baseband video signal, a carrier color signal, and a carrier color signal as shown in FIG.
The television signal of the lower adjacent channel is also converted by the synchronous detector 10 into an adjacent carrier video signal, an adjacent carrier color signal, and an adjacent carrier audio signal as shown in FIG. 5c.

The shaded part in Fig. 5c is the synchronous detector 1.
The zero output is removed when it passes through the low pass filter 12. This portion contains most of the adjacent carrier video signal. However, signals other than the shaded portion in FIG. 5c are mixed into the baseband video signal in FIG. 5b.

Next, the operation of removing the adjacent carrier video signal and adjacent carrier audio signal mixed into the baseband video signal will be described. Since the output υ _Q (t) of the 90° phase shifter has a phase difference of 90° with the output of the voltage controlled oscillator 20, υ _Q (t)=A ₀ sin (ω ₀ t+ ₀ )...(12) This is added to the second synchronous detector 11 consisting of a voltage multiplier along with υ _V (t) in Equation 1, and its output υ _PQ
(t) is passed through the low-pass filter 13, υ _PQ (t)=-A ₀ /2 [I(t)+n _C (t)] cos − A ₀ /2 [Q (t) + n _S (t)] sin... (13
) However, let ω ₀ = ω _V. This υ _PQ (t) is amplified by a signal amplifier 15 and applied to a phase detector 18.

In the phase detector 18 consisting of a voltage multiplier, υ _PV
(t) and υ _PQ (t) are voltage multiplied, resulting in a control voltage υ _C (t).

υ _C (t)=υ _PV (t)・υ _PQ (t)=−A ₀ ² /8 {[I
(t)+n _C (t))] ² − [Q(t)+n _S (t)] ² }si
nθ −A ₀ ² /4 [I(t)+n _C (t)][Q(t
)+n _S (t)]cosθ...(14) Here, θ=2. However, the amplification degrees of the first and second signal amplifiers are assumed to be 1 here.

The video carrier signal υ _V (t) is transmitted through vestigial sideband transmission, but its transmission characteristics are different from normal vestigial sideband transmission and consist of a double sideband transmission part and a single sideband transmission part. It's on. That is, the residual sideband characteristic of the video carrier signal υ _V (t) shown in FIG. 6a is the superposition of the double sideband characteristic shown in FIG. 6b and the single sideband characteristic shown in FIG. 6c. .

A signal by double sideband transmission consists only of in-phase components with respect to the phase of the carrier wave, and a signal by single sideband transmission consists of in-phase components and orthogonal components. Now I _L (t)
is the in-phase component of the signal υ _V (t) due to double-sideband transmission,
I _U (t) is the in-phase component of the signal υ _V (t) by single sideband transmission, and Q _U (t) is the signal υ _V (t) by single sideband transmission.
(14) becomes υ _C (t)=−A ₀ ² /8{[I _L (t)+I _U (t)+n _C (t
)] ² − [Q _U (t) + n _S (t) ² }sinθ −A ₀ ² /4 [I _L (t) + I _U (t) + n _C (t)] [Q
_U (t) + n _S (t)] cos θ... (15).

If the low-pass filter characteristics of the low-pass filters ₁₆ and 17 are in _FIG . ⁷ or narrower than that in _{FIG .} ′(t)〕 ² −
[n _S ′(t)] ² } sinθ−A ₀ ² /4 [I _L (t) +n _C ′(t)] [n _S ′(t)] cosθ……(16
) where n _C ′(t) and n _S ′(t) are the in-phase and quadrature components of narrowband Gaussian noise n(t) after passing through the low-pass filter 16.

If I _L (t)≧n _C ′(t) and I _L (t)≧n _S ′(t), υ _C (t)=−A ₀ ² /8 [I _L (t)] ² sinθ −A ₀ ² /4 [I _L (t)] [n _S ′ (t)] cosθ ... (17) [I _L (t)] Since ² ≠ 0, the loop bandwidth is the second If it is narrow enough to remove the term component, then
The voltage controlled oscillator 20 is controlled so that θ=0. That is, due to the phase error between the video carrier signal υ _V (t) and the output υ ₀ (t) of the voltage controlled oscillator 20,
=0.

Here, even if the loop bandwidth is made narrow enough to be 0, the average value of will be 0, and a certain amount of the noise component represented by the second term of equation (17) will remain. This noise component is generated by the voltage controlled oscillator 2.
Gives fluctuation to the output phase and output frequency of 0.

However, when comparing the second term of equation (17) with the second term of equation (15), the difference in amplitude is significantly large.
Q _U (t)≫n _S (t), _S () ² > _S ′() ² ,
This is also because I _U (t) is not included in equation (17). However, _S () ² and _S ′ () ² are n _S (t
)
and the variance of n _S ′(t).

That is, by inserting the low-pass filters 16 and 17 as shown in FIG. 2, the influence of the noise component, that is, the second term in equation (15) or the second term in equation (14), can be significantly reduced. can.

Furthermore, if the bands of the low-pass filters 16 and 17 are made narrower, the dispersion _S '() ² of n _S '(t) becomes smaller in proportion to the band. The voltage controlled oscillator 20
The fluctuations in the output phase and output frequency become smaller.

However, this frequency fluctuation does not completely disappear, but remains, albeit slightly. This residual frequency fluctuation is detected by the synchronous detector 1.
0, frequency fluctuations are applied to the audio signal carrier of the channel desired to be received, and the video carrier and audio carrier of the lower adjacent channel. This is because the synchronous detector 10 consists of a voltage multiplier, and this voltage multiplier converts the frequency of the signal from the high frequency input section 9 using the output from the voltage controlled oscillator 20.

Television audio signals are frequency modulated and have a maximum frequency deviation of ±25KHz. If the fluctuation given to the audio signal carrier of the channel desired to receive is about 20 to 30 Hz, the signal-to-noise ratio of the demodulated audio signal is about 60 dB, and this level of signal-to-noise ratio is permissible. On the other hand, if frequency fluctuations of several Hz to 20 to 30 Hz are applied to the video carrier wave of the lower adjacent channel, the frequency spectrum of the carrier color signal of the lower adjacent channel will also fluctuate to the same extent, resulting in the baseband video signal of the desired channel being received. Also, unlike the spectrum of the carrier color signal, the spectrum does not have energy concentrated at each frame frequency (30Hz). Also, the audio carrier wave of the lower adjacent channel will fluctuate to the same extent,
Since the audio carrier wave is frequency modulated, the spectrum of the carrier audio signal originally has a frequency width of about ±100KHz.

Arithmetic unit 32, frame memory 33, chroma inverter 34, motion detector 35, coefficient generator 3
6, an address generator 37, and a memory controller 38. The temporal low-pass filter is known as a noise reducer. This temporal low-pass filter is a recursive filter having a delay element for one frame of the video signal, and is a circuit that temporally averages the video signal for each frame period. As shown in FIG. 8, its frequency characteristic is a comb shape that repeats at the frame frequency. Moreover, the depth of the valley in this frequency characteristic changes according to the coefficient K. This coefficient K is then a function of the interframe difference signal examined by the motion detector 35.

Part of the carrier color signal, carrier audio signal, and carrier video signal of the lower adjacent channel frequency-converted by the synchronous detector 10 has a fluctuating spectrum, so it is processed by the motion-adaptive temporal low-pass filter. Most of it is removed. In this case, since it is a motion adaptive type, blurring is reduced by setting K→0 in the moving image portion, and when the image is close to a still image, K is increased to increase the degree of interference signal removal. In this way, interference from the lower adjacent channel mixed into the baseband video signal of the channel desired to be received can be removed.

The channel selection voltage stored in the voltage storage device 27 is selected by the voltage selector 28 in accordance with the desired channel inputted from the control input device 29 and is applied to the voltage adder 26. The voltage controlled oscillator 20 is controlled by this channel selection voltage, and the synchronous carrier wave υ ₀ (t)
occurs. The audio carrier wave υ _S (t) and this synchronous carrier wave υ ₀ (t) are applied to a synchronous detector 10, resulting in the generation of an audio intermediate frequency signal υ _PS (t). The frequency of the audio intermediate frequency signal υ _PS (t) is broadcasted by the frequency pull-in circuit into a video carrier wave υ _V
(t) carrier frequency ω _V (t) and audio carrier wave υ _S (t)
The frequency of the synchronous carrier wave υ ₀ (t) is controlled so that it is equal to the difference in carrier frequency ω _S of ω _IF , that is, ω IF. When this frequency falls into the frequency pull range of the Costas loop, the Costas loop rapidly enters a state of phase locking. When the Costas loops are phase synchronized, the synchronous detector 10 obtains a video signal υ _PV (t) and an audio intermediate frequency signal υ _PS (t). These signals pass through a low-pass filter 12, etc., the video signal is sent to the video output circuit 41, and the audio intermediate frequency signal is sent to the frequency discriminator 2.
3, and an audio signal that is the demodulated signal is output to the audio output circuit 42.

In the configuration shown in Figure 2, which was clarified in the above explanation,
The baseband video signal synchronously detected by the Costas loop is low-pass filtered by a low-pass filter whose low-pass filtering range is the frequency range of the baseband video signal and the frequency of the audio intermediate frequency signal. Most of the energy of the carrier video signal of the channel is removed, and this baseband video signal is further subjected to frame frequency interval comb filtering using a temporal low-pass filter, so that the carrier chrominance signal and the carrier audio signal of the adjacent channel are removed. Most of the interference to the baseband video signal that you want to receive can be removed. Furthermore, since the temporal low-pass filter is of a motion-adaptive type, the interference removal effect is particularly great when the desired video signal to be received is close to a still image.

However, in the above configuration,
Since signals in the frequency range where no luminance signal spectrum exists in the baseband video signal are also filtered by the temporal low-pass filter, some signals remain that are not removed from the carrier color signal of the lower adjacent channel. There's a problem.

OBJECT OF THE INVENTION An object of the present invention is to provide a television synchronization system that enables a comb filter to remove signals in a frequency range where no luminance signal spectrum exists in a baseband video signal synchronously detected by a Costas loop. is to provide a receiver.

Structure of the Invention The television synchronized receiver of the present invention includes a voltage controlled oscillator and a phase adjustment of the output of the voltage controlled oscillator.
A 90° phase shifter that shifts the phase by 90°, the output of the voltage controlled oscillator, and the output of the 90° phase shifter are each used as synchronous carrier waves, and the in-phase and quadrature components of the video carrier signal are synchronized with each synchronous carrier wave. First and second synchronous detectors for detecting waves, and first and second low-pass filters for low-pass filtering the outputs of the first and second synchronous detectors in the frequency range of the video baseband and audio intermediate frequency signals. a phase detector that detects a phase difference between the video carrier signal and the output of the voltage controlled oscillator by multiplying the outputs of the first and second low-pass filters; and an output of the phase detector. a signal amplifier for amplifying the output of the first low-pass filter;
an A/D converter that converts the baseband video signal in the output of the signal amplifier from analog to digital; a means for detecting the frequency of the color subcarrier of the lower adjacent channel; a comb filter provided with means for frequency shifting the comb filter frequency by the frequency of the color subcarrier of the lower adjacent channel; and subtracting the output of the comb filter from the output of the A/D converter. The subtracter is equipped with a subtracter that performs digital-to-analog conversion of the output of the subtracter, and a D/A converter that converts the output of the subtracter from digital to analog.This makes it possible to remove signals in a frequency range where no luminance signal spectrum exists.

Here, the concept of removing signals in a frequency range where no luminance signal spectrum exists using a comb filter will be explained. In the example shown in FIG. 2, the output of the first synchronous detector 10 includes, in addition to the baseband video signal shown in FIG.
It has already been mentioned that the carrier color signal and the carrier audio signal of the lower adjacent channel shown in Figure c are included.
Among these, the carrier color signal has a frequency spectrum as shown in FIG. That is, Figure 9a
, the adjacent chrominance signal subcarrier and the adjacent video signal carrier are shown to correspond to the adjacent chrominance signal subcarrier and the adjacent video signal carrier of FIG. 5c, respectively, and the spectrum of the carrier chrominance signal of the lower adjacent channel is is indicated by a dotted line. However, the spectrum of the video signal of the lower adjacent channel is omitted to simplify the diagram.

The spectrum shown by the solid line in FIG. 9a is the spectrum of the video signal of the channel desired to be received. The interval is equal to the horizontal scanning frequency f _H of the television signal. However, the illustration of the carrier color signal of the desired reception channel is omitted. The spectrum of the video signal of the channel desired to receive has a frequency difference of 1/2 of f _H /2 from the spectrum of the carrier color signal of the lower adjacent channel, and no frequency is incorporated. However, if the video carrier frequencies of both channels are in accordance with the standard,
The difference between the frequency of the spectrum of the lower adjacent channel carrier color signal and the frequency of the spectrum of the desired channel video signal to be received is 2.6 KHz. Because (f
This is because _H /2)×273−6MHz=2.6KHz. Furthermore, this frequency difference is not always fixed. If the transmission frequency (video signal carrier frequency) of one television signal moves, the frequency difference between the two spectra will also increase or decrease by that amount.

Therefore, it is desired to realize a rectangular filter having the frequency characteristics shown in FIG. 9b. The characteristics of this comb filter are such that it blocks the spectrum of the carrier color signal of the lower adjacent channel shown by the dotted line in FIG. 9a, and passes the spectrum of the video signal of the channel desired to be received. Furthermore, the frequency of the comb filter is adapted to shift in accordance with the frequency difference between the video carrier waves of both channels.

DESCRIPTION OF EMBODIMENTS An embodiment of the present invention will be described below with reference to the drawings. FIG. 10 is a block diagram of a main part of a television synchronization receiver according to an embodiment of the present invention. In FIG. 10, 43 is a high frequency input section;
44 is a first synchronous detector, 45 is a second synchronous detector, 46 and 47 are first and second low-pass filters, 48 and 49 are signal amplifiers, 50 and 5
1 is a third and fourth low-pass filter, 52 is a phase detector, 53 is a Costas loop low-pass filter, 54
55 is a voltage controlled oscillator, and 55 is a 90° phase shifter, which constitute a Costas loop. 56 is an audio intermediate frequency amplifier, 57 is a frequency discriminator, 58 is a voltage subtractor, and 59 is a low-pass filter, which constitute a frequency pull-in circuit, the output of which is sent to the voltage adder 60 to filter the low frequency of the Costas loop. It is added to the output of filter 53. 61 is a voltage storage device, 62 is a voltage selector, and 63 is a control signal input device, which constitute a channel selection voltage generation circuit. Voltage selector 62
The output voltage is also summed in a voltage adder 60 with the output of the low-pass filter 53 of the Costas loop. 6
4 is a sampling pulse generator which separates a color burst signal from the output of the signal amplifier 48 and generates a sampling pulse using this signal; 65 is an A/D converter which converts the output of the signal amplifier 48 from analog to digital; It is a D converter. 66 is a bandpass filter that performs bandpass filtering on the output of this A/D converter 65 with the frequency of the color subcarrier of the lower adjacent channel as the center frequency; 67 is a lowpass filter that receives the output of this bandpass filter 66 as an input; 68, a phase detector forming a phase lock loop together with the local oscillator 69; 70 a frequency counter that counts the output frequency of the local oscillator 69; 71, which receives the output of the A/D converter 65 as an input and operates in the phase lock loop; A comb filter 72 includes means for shifting the detected comb filter frequency by the frequency of the color width carrier of the lower adjacent channel; 73 is a temporal low-pass filter that temporally filters the output of this subtracter; 74 is a video signal filter;
75 digitally converts the output of this video signal filter.
A D/A converter performs analog conversion, 76 is a video output circuit, and 77 is an audio output circuit.

The configurations and operations of the Costas loop, frequency pull-in circuit, tuning voltage generation circuit, and time-direction low-pass filter of the television synchronized receiver of this embodiment configured as described above are shown in FIG. Since these parts are the same as those corresponding to , a description thereof will be omitted. In this embodiment, means for detecting the frequency of the color subcarrier of the lower adjacent channel, a comb filter for shifting the filtering frequency of the comb filter by the detected frequency, and the comb filter The operation of separating lower adjacent channel carrier color signals using a subtractor and a subtractor will now be described.

The center frequency of bandpass filter 66 is chosen to be equal to the frequency of the chrominance subcarrier of the lower adjacent channel, and this frequency is designated f ₀ . A phase lock loop consisting of 67, 68, and 69 is phase locked to the color subcarrier of the lower adjacent channel having this frequency f ₀ . Therefore, local oscillator 69 oscillates at frequency f ₀ . The output of this local oscillator 69 is counted by a frequency counter 70, and the frequency f ₀ is obtained as data.

Figure 11a shows the configuration of a conventional 2H comb filter. 78 and 79 are one horizontal period (1H) delay elements, and 80 is an adder. If the impulse response of the filter is h _i , its transfer function H(z) is: H(z)= _N 〓 ^i=-N h _i z ^-i ……(18) Here, z is a complex number, z ^-i is a delay of i periods expressed using a unit delay operator z ^-1 .

Impulse response h _i of the comb filter in Figure 11a
is h _i = 1/4 for i = -N, h _i = 1/2 for i = 0, h _i = 1/4 for i = N, h i = 0 for i = ±N, and other than _i = 0. Therefore, the transfer function H ₁ (z) is H ₁ (z)=1/4(z ^N +2+z ^−N (19) where N is the number of samples in 1H.

The frequency characteristic H ₁ (f) can be calculated by substituting z ^-1 = e ^-j2 〓 ^fT into equation (19) and calculating H ₁ (f) = 1/4 (e ^j2 〓 ^fNT +2+e ^-j2 〓 ^fNT = 1/2 { 1+cos(2πfNT)} ...(20) This is illustrated in Figure 12a.

In order to shift this frequency characteristic by the frequency _f0 , a comb filter as shown in FIG. 11b is constructed. 81 and 82 are one horizontal period delay elements that delay the input signal by one horizontal period, 83 and 84 are sine multipliers that multiply the input signal by a sine function, and 85 is an adder.

The impulse response of this filter is i=-N and h _i =
cos(−2πf ₀ NT) = cos(2πf ₀ NT), i=0 and h _i =
1. Since i = N and h _i = cos (2πf ₀ NT), the transfer function H ₂ (z) is H ₂ (z) = 1/4 {cos (2πf ₀ NT) z ^N +2 + cos (2πf ₀ NT)z ^-N } ...(21) The frequency characteristic H ₂ (f) is H ₂ (f) = 1/4 {cos (2πf ₀ NT) e ^j2 〓 ^fNT +2 + cos (2πf ₀ NT) e ^{- j2} 〓 ^fNT } ...(22) = 1/4 [1 + cos {2π (f + f ₀ ) NT}] + 1/4 [1 + cos {2π (f - f ₀ ) NT}] = 1/2H ₁ (f + f ₀ ) + 1 /2H ₁ (f−f ₀ )……(23
) The first and second terms of equation (23) are H ₁ of equation (20)
(f) shifted in frequency by −f ₀ and f ₀ . This relationship is shown in FIG. 12b.

Since the carrier color signal of the lower adjacent channel exists in the vicinity of the color subcarrier frequency f ₀ , the comb filter 71 transmits the carrier color signal centered at the frequency f ₀ to the comb filter shown in FIG. 11b above. A configuration is used in which bandpass filters whose bandwidth is the band of the color signal are connected in series. The comb filter 71 configured in this manner filters and separates the carrier color signal of the lower adjacent channel from the output of the A/D converter 65. This separated lower adjacent channel carrier color signal is subtracted from the output of the A/D converter 65 by a subtracter 72. The output of subtractor 72 is the one with the lower adjacent channel carrier color signal removed. The output of the subtracter 72 is applied to a temporal low-pass filter 73, and the signal is processed in the same manner as described in the conventional example below.

In addition, in the above explanation, 2H is used as a comb filter.
By using the configuration shown in Figure 13b for the 1H-shaped comb filter shown in Figure 13a, the comb filter frequency can be shifted by f ₀ in the same way as above. I can do it. Here, 86 is 1H
Delay element, 87 is an adder, 88 is a sine multiplier, 8
9 is a 1H delay element.

Effects of the Invention As is clear from the above description, the present invention detects the frequency of a lower adjacent channel color subcarrier from a signal obtained by a synchronous detector of a television synchronous receiver using a Costas loop. and the means to
A comb filter is provided to shift the comb filter frequency by this frequency, and the output of the comb filter is subtracted from the output of the synchronous detector, so the Costas loop performs synchronous detection. An effect can be obtained in that signals in a frequency range in which no luminance signal spectrum exists in the baseband video signal can be removed.

Furthermore, by configuring the means for detecting the frequency of the color subcarrier of the lower adjacent channel by a phase lock loop that phase-locks the output of the signal amplifier that amplifies the output of the synchronous detector, a television that applies digital signal processing can be realized. This has the effect of further increasing the effectiveness of the application of the Jiyoung receiver.

Furthermore, a means for frequency shifting the comb filter frequency by the frequency of the color subcarrier of the lower adjacent channel is added to the impulse response of the comb filter, and a sine function with the color subcarrier frequency of the lower adjacent channel as a variable is added. By using a sine multiplier for multiplication, even if the frequency of the color subcarrier of the lower adjacent channel changes, the comb filter frequency changes by the changed frequency.

[Brief explanation of drawings]

Fig. 1 is a block diagram of the main part of the conventional example, Fig. 2 is a block diagram of the main part of the previously proposed television synchronous receiver, and Fig. 3 is a frequency characteristic diagram of the low-pass filter that filters the output of the synchronous detector. , Fig. 4a shows the baseband frequency characteristic diagram of the video signal, Fig. 4b shows the frequency characteristic diagram of the video signal filter, and Fig. 5a shows the frequency relationship between the channel desired to receive the television signal and the lower adjacent channel. FIG. 5b is a diagram showing the frequency conversion relationship of the channel desired to be received, FIG. 5c is a diagram showing the frequency relationship of the lower adjacent channel, and FIG. Figure 6b is a characteristic diagram showing double sideband transmission during vestigial sideband transmission of a television signal, and Figure 6c is a characteristic diagram showing single sideband transmission during vestigial sideband transmission of a television signal. FIG. 7 is a frequency characteristic diagram of the third and fourth low-pass filters, FIG. 8 is a frequency characteristic diagram of the motion-adaptive temporal low-pass filter, and FIG. 9a is a frequency characteristic diagram of the desired reception channel. FIG. 9b is a frequency characteristic diagram of the comb filter used in the present invention, and FIG. 10 is a diagram showing the spectrum of the video signal and the spectrum of the carrier color signal of the lower adjacent channel. A block diagram of the main parts of a synchronous receiver, Fig. 11a is a block diagram of a conventional 2H-type comb filter,
Figure 11b is a block diagram of the 2H comb filter of the present invention, Figure 12a is a frequency characteristic diagram of a conventional 2H comb filter, and Figure 12b is the 2H comb filter of the present invention. The frequency characteristic diagram of Fig. 13a is the conventional 1H
The configuration diagram of the comb filter, FIG.
FIG. 2 is a configuration diagram of a 1H-shaped comb filter. 44...First synchronous detector, 45...Second synchronous detector, 46...First low-pass filter, 47...Second low-pass filter, 48, 49...Signal amplifier, 52...Phase detector , 54... Voltage controlled oscillator, 55... 90° phase shifter, 64... Sampling pulse generator, 65... A/D converter, 66... Bandpass filter, 67... Phase detector, 6
8...Low pass filter, 69...Local oscillator, 70...Frequency counter, 71...Comb filter, 72...Subtractor.

Claims (1)

[Claims] 1. A voltage controlled oscillator, a 90° phase shifter that shifts the phase of the output of the voltage controlled oscillator by 90°, and an output of the voltage controlled oscillator and an output of the 90° phase shifter. first and second synchronous detectors each having a synchronous carrier wave and synchronously detecting in-phase and quadrature components of a video carrier signal using the respective synchronous carrier waves;
and first and second low-pass filters that low-pass filter the output of the second synchronous detector in the frequency range of the video signal baseband and the audio intermediate frequency signal; a phase detector for detecting a phase difference between the video carrier signal and the output of the voltage controlled oscillator by multiplying the output of the first voltage controlled oscillator; A signal amplifier that amplifies the output of the low-pass filter, an A/D converter that converts the baseband video signal in the output of this signal amplifier from analog to digital, and detects the frequency of the color subcarrier of the lower adjacent channel. a comb filter having an input of the output of the A/D converter and a means for shifting the comb filter frequency by the frequency of the color subcarrier of the lower adjacent channel; and the comb filter. A subtracter that subtracts the output of the A/D converter from the output of the A/D converter, and a D/A converter that converts the output of the subtracter from digital to analog. TV synchronized receiver. 2 means for detecting the frequency of the color subcarrier of the lower adjacent channel converts the output of the signal amplifier into an analog
2. The television synchronization receiver according to claim 1, comprising a phase lock loop for phase locking the output of an A/D converter for digital conversion. 3 means for frequency shifting the comb filter frequency by the frequency of the color subcarrier of the lower adjacent channel;
2. The television synchronization receiver according to claim 1, comprising a sine multiplier that multiplies the impulse response of the comb filter by a sine function with the color subcarrier frequency of the lower adjacent channel as a variable.