KR840002386B1 - Power source device - Google Patents

Abstract

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Description

Power supply

1 is a circuit diagram of a conventional example.

2 and 3a and 3b are perspective views showing the specific structure of the transformer 10 in FIG.

4 and 5 are graphs showing the relationship with the hysteresis curve in the device of FIG.

6 is a time chart for explaining the operation of FIG.

7 is a circuit diagram of an embodiment according to the present invention.

8 and 9 are perspective views showing different concrete structures of the transformer 10 in FIG.

10 and 11 are equivalent circuit diagrams of FIG.

12 and 13 are time charts for explaining the operation of FIG.

FIG. 14 is an equivalent circuit diagram of FIG. 11 further simplified.

FIG. 15 is a circular diagram showing the trajectory of the inductance vector of FIG.

FIG. 16 is a graph showing the anti-resonance curve in the equivalent circuit of FIG.

FIG. 17 is a graph showing the anti-resonance curve when the horizontal axis is the mutual inductance when changing parameters in FIG.

18 is an experimental circuit diagram imitating the circuit of FIG.

FIG. 19 is a graph showing the anti-resonance curve in the experimental circuit of FIG. 18 and the anti-resonance curve in the equivalent circuit of FIG.

20 and 21 are graphs showing antiresonance curves when the control current Ic in FIG. 7 is changed linearly.

22 is a graph showing the hysteresis curve in FIG.

23 is a graph showing conventional load characteristics.

24a to e are time charts showing the output waveforms for various parts of FIG.

25 is a graph showing the load characteristics of this embodiment.

26 is a circuit diagram of another embodiment of the present invention.

27 to 29 are equivalent circuit diagrams simplifying the circuit of FIG.

The present invention relates to a power supply apparatus using a transformer for controlling a magnetic flux to obtain a constant voltage output. As the prior art of the present invention, a power supply apparatus already proposed by the present invention will be briefly described with reference to the drawings.

In FIG. 1, the AC from the AC power supply 1 is rectified by the full-wave rectifier 2 and smoothed by the smoothing capacitor 3 to generate a DC output. At this output end, the choke coil 4 and the primary coil (first coil) coil N ₁ of the transformer 10 and the collector emitter of the switching transistor 5 are connected in series, and the collector and emitter of the transistor 5 are connected in series. The switching diode 6 and the resonance capacitor 7 are connected in parallel between them. In addition, a frequency pulse signal of, for example, about 15 to 20 Hz is supplied to the base of the transistor 5.

The secondary coil N ₂ of the transformer 10 is connected with a small capacitance capacitor 13 for para-electric oscillation or wave type and a rectifier circuit 14 composed of two diodes. The output terminal has a smoothing capacitor 15 and a load. 16 is connected. The control circuit 17 also detects the magnitude of the output voltage E ₀ from the rectifier circuit 14 and transfers the control current Ic to the control coil Nc of the transformer 10.

Here, the transformer 10 uses, for example, the structure shown in FIG. 2, and the pair of magnetic cores 11 and 12 includes, for example, a square or rectangular plate nose fishing unit 10E. ) Are formed in the direction orthogonal to the four corners and have the same cross-sectional area (10A) to (10D) and the cores (11) and (12) are the (10A) to (10D) And (10A) to (10D) are in contact with each other at the end and face each other, and are combined so as to be a cube or a cube as a whole. In addition, the cores 11 and 12 are formed of a ferrite material, for example.

In addition, the primary coil N ₁ is wound over the magnetic cores 10B and 10D of the core 11, and the secondary coil N ₂ is wound over the magnetic cores 10A and 10C of the core 11. The control coil Nc is wound around the core 10A, 10B of the core 12. As shown in FIG. Therefore, in this case, the coils N ₁ and N ₂ are trans-bonded so that the coils N ₁ , N _2, and Nc are orthogonally coupled, but the coupling coefficients of the coils N ₁ and N ₂ at this time are about 0.5 to 0.6.

The magnetic flux distribution state of the transformer 10 is also shown in the 3a and 3b. In other words, if the excitation current of the coil N ₁ is I ₁ , the oscillation current of the coil N ₂ is I ₂ , and the load current from the coil N ₂ is I _L , the electric magnetic force NI of the transformer 10 becomes as follows.

NI = N ₁ I ₁ + N ₂ I ₂ -N ₂ I _L ... (One)

The magnetic force generated in the positive half cycle period of the output voltage Eo is + øs (Fig. 3a) and the magnetic flux generated in the negative half cycle period is -øs by the magnetic force NI, and the control coil Nc and the control current through the magnetic force NI If the magnetic flux generated by Ic is øc, the magnetic flux øs and øc are subtracted from each other in the magnetic fluxes 10A and 10D in the positive half cycle period, and the magnetic flux øs and øc are mutually different in the magnetic flux 10B and 10C. In the negative half cycle period (FIG. 3B), an inverse relationship is established.

Thus, for example, in the case of the BH characteristic (magnetization characteristic) in FIG. 4, at the peak time of the positive half cycle period, the operating points of the subjects 10A and 10D become point a, and the subject 10B, The operating point of 10C becomes point b, and at the peak time of the negative half-cycle period, the operating point of awareness 10B, 10C becomes point c, and the operating point of awareness 10A, 10D. Is the point d. Therefore, the operating region of the awareness 10A, 10D becomes the section of arrow 1A, and the operating region of the awareness 10B, 10C becomes the section of arrow 1B, and the output voltage of positive half cycle period. Eo is determined by the magnetic flux density + Bs of the magnetic fluxes 10A and 10D of the point a, and the output voltage of the negative half cycle period is determined by the magnetic flux density -Bs of the magnetic fluxes 10B and 10C of the point c.

Points a and c change depending on the magnetic flux? C, and the magnetic flux? C changes according to the control current Ic. Therefore, when the current Ic is controlled, the output voltage Eo is controlled. This output voltage Eo (t) is

In this case, the first term is the voltage induced by the trans coupling and the second term is the voltage induced by the parametric coupling. That is, the output voltage Eo (t) includes a voltage due to trans coupling and a voltage due to parametric oscillation. In addition, the ratio of the positive voltage varies depending on the coupling coefficient between the coils N ₁ and N ₂ , that is, the shape of the core and the winding method of the coil.

Therefore, as shown in FIG. 5, the magnetic flux when Ic = O is ø ₁ , the magnetic flux in the combined state is ø ₂ , the magnetic flux in the subtracted state is ø ₃ , and the magnetic flux ø ₁ , ø _2, and ø _3. If the change between them is Δø ₂ and Δø ₃ , the output voltage at Ic = O is as follows.

In addition, the output voltage e _os when the magnetic flux? Of Ic ≠ O is in the nonlinear region is as follows.

And because of the nonlinearity of the B-H properties

Δø ₃ > Δø ₂

If point e, b is in saturation region,

It becomes Therefore, according to this equation, the output voltage Eo is controlled when the change Δø ₃ of the magnetic flux is controlled by the control current Ic.

According to the power supply device of FIG. 1 using such a transformer 10, according to the on / off switching operation of the transistor 5, the same operation as the horizontal deflection of the television receiver is performed, and the collector voltage of the transistor 5 6a claim 1 primary coil of the transformer 10 is changed, and as shown in Fig N ₁ flows in the exciting current I _1, as illustrated in FIG. 6b. The choke coil 4 here limits the collector current of the on-period of the transistor 5 to stabilize its switching operation. Since the transformer 10 is excited by the current I ₁ , in the parallel circuit composed of the secondary coil N ₂ and the condenser 13, the output voltage Eo and the current I ₂ of the waveforms shown in FIGS. 6c and 6d are obtained. This voltage Eo is supplied to the rectifier circuit 14, and the load 16 is supplied with a constant DC voltage. FIG. 6E shows the induced voltages of the perceptions 10A and 10D of the transformer 10, and FIG. 6F shows the induced voltages of the perceptions 10B and 10C, respectively. 6g shows the current flowing in the midpoint tap of the secondary coil N ₂ of the transformer 10. Since the current I ₁ is unbalanced in the positive half cycle period and the negative half cycle period, this current is also unbalanced.

The output voltage Eo from the rectifier circuit 14 is detected by the control circuit 17, and the control current Ic flows through the control coil Nc of the transformer 10 in accordance with the detection output. That is, for example, when the output voltage Eo increases, the control current Ic increases and the maximum magnetic flux density Bs decreases, so that the output voltage Eo decreases. On the contrary, when the output voltage Eo decreases, the output voltage Eo increases in the direction of increasing the output voltage Eo. By controlling, the output voltage Eo is constantly stabilized.

Therefore, in such a power supply device, when the operation mode of the transformer 10 is in the saturation region, the capacitor 13 is a parametric oscillation capacitor, and the output voltage Eo is obtained by the sum of the transformer coupling and the parametric coupling. At this time, iron loss of the core and the coil winding of the coil winding are generally great due to the high magnetic flux density operation, so that the heat generation of the transformer increases and power consumption increases. In the case where the operation mode is in the linear region, the capacitor 13 is merely a waveform capacitor and has a small capacity. In this case, the output voltage is Eo = (2 ° ₁ -Δ ° ₃ ) KN ₂ f and the control sensitivity is lowered, so that the control range for the change of the input voltage Ei and the load 16 is narrow. In addition, as the magnetic flux density decreases, the number of turns N ₂ of the left coil increases at the same operating frequency, resulting in a large transformer, and in the case of supplying power to the same load capacity, the primary excitation current increases, resulting in consumption. Power is increased. In addition, according to the prior art as described above, there is a drawback that the output voltage becomes unbalanced between positive and negative.

The object of the present invention is to improve such prior art again, thereby increasing the control range by always maintaining the control sensitivity, and at the same time miniaturizing and reducing the transformer, or supplying the maximum load for the transformer having the same size and weight. In addition, the present invention provides a power supply device with high conversion efficiency and low power consumption.

DESCRIPTION OF THE PREFERRED EMBODIMENTS A preferred embodiment of a power supply apparatus according to the present invention will be described below with reference to the drawings.

7 is a circuit diagram showing a first embodiment of the present invention, in which a resonant capacitor 20 is connected in parallel with a secondary coil N ₂ of a transformer 10. In FIG. 7, for example, the AC output from the corresponding AC power supply 1 of 100 V is rectified by the full-wave rectifier 2, and smoothed by the smoothing capacitor 3 to produce a DC output. Between the DC output terminals, the choke coil 4 and the collector of the primary coil N ₁ of the transformer 10 and the switching transistor 5. The emitters are connected in series. The diode 6 and the capacitor 7 are connected in parallel between the collector and the emitter of the transistor 5, respectively. The secondary coil N ₂ of the transformer 10 is connected to the resonant capacitor 20, which is a main part of the present invention, and transmits the resonant frequency fo between the coil N ₂ and the capacitor 20 (capacity Cs). It is set slightly higher than the operating frequency f of (10). The AC output from the parallel resonant circuit composed of the secondary coil N ₂ and the resonant capacitor 20 is rectified by a full-wave rectifier 21 composed of a diode bridge, for example, and smoothed by a smoothing capacitor 22. It is supplied to the load 23. The output voltage Eo from the full-wave rectifier 22 is also transmitted to the control circuit 24 and supplied to the control coil N ₂ of the transformer 10 as the control current Ic.

Here, as the transformer 10, the orthogonal coupling transformer 10 of FIG. 2 may be used, and as shown in FIG. 8, a pair of magnetic cores 11 and 12 having the same shape as that of FIG. It is also possible to use the transformer 10 'wound around the secondary coil N ₂ as well as the primary shaft coil N ₁ in the perception 10B, 10D. In addition, by using the magnetic core 11 ′ having the same shape as the magnetic core 11, the cross sections of each of the magnetic cores 10A to 10D are opposed to the core base 10E of the magnetic core 11 so that the magnetic core ( The choke coil and the orthogonal transformer may be combined together by winding the coil N ₃ for the choke coil over the perceptions 10B and 10D of 11 '). In addition, a coil 10 wound around each of the coils N ₁ , N ₂ , Nc is formed as a transformer 10 of FIG. 7 by combining an EI type (or EE type) magnetic core as shown in FIG. 9. Also good. At this time, the magnetic flux (solid arrow in Fig. 9) generated by the control coil Nc wound around the central magnetic field 10M is the magnetic field 10L at both ends 10L and 10R at either end. The magnetic flux (dashed arrow) generated by the primary and secondary coils N ₁ and N ₂ wound around) is added, and the other side acts to be subtracted. Accordingly, although the control coil Nc is not alternatingly coupled to the primary and secondary coils N ₁ and N ₂ , the control coil Nc may change the magnetic fluxes of the magnetic cores 10L and 10R of the core 10. Magnetic flux control as shown in Figs. And 8 may be performed.

Next, the relationship between the output from the secondary coil N ₂ and the input applied to the primary coil N ₁ of the transformer 10 will be described under the above parallel resonance conditions.

First, if the circuit in the vicinity of the transformer 10 of FIG. 7 is displayed equivalently, it becomes like FIG. In FIG. 10, the power supply section composed of the full-wave rectifier 20 and the smoothing capacitor 3 of FIG. 7 is indicated by a DC power supply Eo, the switching transistor 5 is a switch SW, and the capacitor 7 is a capacitor. In Cr, the resonance capacitor 20 is represented by the capacitor Cs, and the load side of the rear end of the capacitor 20 is AC, and the load resistance R _L is respectively indicated. In addition, transformer 10 is converted to a T-type equivalent circuit, the choke coil 4 of the inductance LS, 1 primary coil self-inductance L _1, 2 primary coil self-inductance L ₂ and these coil N ₂ of N ₁ N _1, This is represented by the mutual inductance M of N ₂ .

As the transformer of the equivalent circuit of FIG. 10, when an ideal transformer is used in which primary and secondary inductances L ₁ and L ₂ are the same and there is no inductance flowing out of a close coupling, M = L ₁ = L ₂ The circuit is simplified as shown in FIG. The voltage across e _ab between this first terminal ab 11 degrees is as help the 12a, the energy stored in the on-period of the switch SW T _ON is consumed in the OFF period T _OFF. Therefore, we can ignore the loss of the system, the time-12a (時間積) in view of a T _ON period shown by oblique lines in e _ab S _ON and T _OFF period of time between _ab e S is the _OFF

S _ON = S _OFF = S... (6)

The relation of is established. The current flowing through the terminal a (primary current of the transformer) i is as shown in Fig. 12b. This is the same as the operation of the horizontal deflection circuit of the television receiver. Moreover, the on-period T _ON in FIG. 12 does not have to coincide with the operating period of the transistor 5, but is the sum of the braking period and the on-operating period of the transistor.

라디안만큼 위상이 진행된 정현파 전압이며, 또한 반주기의 시간적이 상기(6)식의 시간적 S와 같아지는 정현파 전압 e'_ab을 기준벡터

로 하여Here, in the interpretation of the circuit, the voltage important component of the primary harmonic component of the current i e _ab is about ₁ in the cycle, such as the voltage e _ab

The reference vector is a sinusoidal voltage whose phase has been advanced in radians, and the sinusoidal voltage e ' _ab whose half-period temporal is equal to the temporal S of Equation (6).

By

e ' _ab = E _m sin wt... (7)

Ei이며 또한 (7)식의 반주기의 시간적 S(제13b)도의 사선부)은

이므로 E_mS_ON=S로 될 때의 (7)식의 정현파 전압 e'_ab는Place it. Here, for example, when T _ON = T _OFF , S _ON (diagonal portion) is substantially smaller than that in FIG. 13a.

Ei and the semi-period temporal S (the oblique line in Fig. 13b) of (7) is

Since the sinusoidal voltage e ' _ab in the equation (7) when E _m S _ON = S is

It becomes

When a constant voltage source that outputs such a sinusoidal voltage e ' _ab is connected between the terminals ab in FIG. 11, the constant voltage source is converted into a constant current source according to the Hosea-Milman's theorem, and simplified. You can get it. The inductance L of FIG. 14 is obtained by using L _S and M.

는, 상기 정현파 전압의 벡터표시

를 사용하면Is displayed. In addition, the current vector of the constant current source

Is a vector display of the sinusoidal voltage

Using

It can be represented as.

는Since this Fig. 14 is a parallel resonance circuit of L, Cs, and R _L , all impedances

Is

의 실수부인 저항성분을 R(W), 허수부인 리액턴스 성분을 X(W)로 놓으면It becomes This complex impedance

If you set the real component of R (W) and the imaginary component of reactance to X (W)

는 부하저항 R_L성분만으로 된다. 이 공진 각 주파수 Wo는Since a circle diagram as shown in FIG. 15 can be depicted according to the change of each frequency W. As also shown in this circle diagram, when W becomes the resonant angular frequency Wo, the inductive reactance and the capacitive reactance cancel each other out,

Becomes only the load resistance R _L component. This resonant angular frequency Wo is

Since the resonant frequency fo is Wo = 2πfo

It becomes

Therefore, the resonant frequency fo of the system is the capacitance of the condenser 20 connected in parallel to the inductance Ls of the choke coil 4, the mutual inductance M between the primary and secondary coils of the transformer 10 and the secondary coil N ₂ . The resonance capacitor 7 determined in Cs and connected in series with the primary shaft coil N ₁ may be considered to contribute only to the determination of the converter pulse width, that is, the off period T _OFF .

Next, in such a resonance mode, when the square root (effective value, rms) of the squared mean value of the output voltage is Eos,

로 할 때에는

=

/jWLs이므로For example, the sinusoidal voltage of the equation (7) is a reference vector.

When we do

=

/ jWLs

It becomes If we express this expression (13) as the coefficient of mutual inductance M,

｜는It becomes In addition, the absolute value of the impedance |

｜

It becomes

｜의 주파수 특성을 제16도에 도시하였다. 이 주파수 특성 그래프는 상기 Ls,M을 매개변수로 할 때의 반공진 곡선을 나타내며, 상기 Cs=0.1㎌로 하고 R=200Ω로한 때를 실선으로, R=375Ω로 할때를 점전으로 각기 도시하고 있다. 또 각각 3개씩의 곡선은 상기 매개변수 Ls,M이Absolute value of impedance obtained from this equation (15) |

The frequency characteristic of | is shown in FIG. This frequency characteristic graph shows the anti-resonance curve when Ls and M are used as parameters, and when Cs = 0.1 Hz and R = 200 Hz, the solid lines are shown as solid lines, respectively. have. Each of the three curves has the parameters Ls and M

I: Ls = 0.94mH, M = 1mH

II: Ls = 2.5mH, M = 1mH

III: Ls = 2.5mH, M = 5mH

It shows when it has the value of.

｜의 변화율이 M<2mH인 경우에 비해 크다. 이와 같이 상호 인덕턴스 M를 변화시킬 때, 시스템의 임피던스의 절대값 ｜

｜은 공진곡선을 따라 변화하며, 한편 공진곡선의 좌우에서 변화율이 다르다.Next, Fig. 17 shows the impedance change when the mutual inductance M is changed with the switching frequency fH = 15.75 kHz fixed and Cs = 0.1 kHz and Ls = 2.5 mH. As can be seen from FIG. 17, when M = 2mH |

It is larger than the case where the change rate of | is M <2mH. Thus, when the mutual inductance M is changed, the absolute value of the impedance of the system |

Is changed along the resonance curve, while the rate of change is different on the left and right of the resonance curve.

Next, the theoretical values calculated based on the circuit analysis are compared with the measured values obtained by using an experimental circuit (Fig. 8) that simulates the actual power supply circuit. FIG. 18 is the same as that of FIG. 11, wherein the wave rectifier 21 of FIG. 7 is the AC load resistance R _L , and the transformer 10 is a transformer as long as L ₁ = L ₂ and the coupling coefficient K = 1. In addition, by changing the tap to change the mutual inductance M, the change in the magnetic flux caused by the change of the control current Ic described above is shown.

In such an experimental circuit, the mutual inductance M and the output voltage when the switching frequency fs = 15.75 Hz of the transistor Q are fixed and the choke coil Ls = 2.5 mH, the resonance capacitance Cs = 0.1 Hz, and the input DC voltage Ei = 120V. The relationship with Eo (rms) is shown in FIG. In FIG. 19, the solid line indicates when R = 200 Hz, the dotted line indicates when R = 375 Hz, and curves A and B represent measured values according to the experimental circuit of FIG. 18, and curves C and D are described above. The theoretical value based on circuit analysis is shown.

Here, the difference between the theoretical value and the measured value is because the output voltage E _OT due to the transformer coupling exists, and if the current flowing through M is iM (t)

Is displayed. Therefore, the measured value E of the total output voltage obtained by the experiment is the sum of the output voltage Eos of the theoretical value based on the circuit analysis and the voltage component by the transformer coupling of the formula (16).

이 대략 전출력전압 Eo의 10% 정도인 것으로 나타나 있다. 또한 제19도로부터, 상호 인덕턴스 M의 값이 작아짐에 따라, 상기 트랜스 결합분이 작아지고 이론값은 측정값에 접근하지만, 이것은 상기 (17)식으로부터 알 수 있다.In FIG. 19, the output voltage by the transformer coupling is shown in FIG.

This is approximately 10% of the total output voltage Eo. Also from Fig. 19, as the value of the mutual inductance M decreases, the trans coupling fraction decreases and the theoretical value approaches the measured value, but this can be seen from the above equation (17).

Although the above has considered the case where the mutual inductance M is changed linearly, the change in the mutual inductance M is nonlinear when the control current Ic changes in the actual power supply device like the seventh.

Next, as the actual control sensitivity, the changes in the output voltages Eo and Eos with respect to the control current Ic are shown in FIGS. 20 and 21. FIG. Here, FIG. 20 shows the time when the direct current component I = OA of the current supplied to the mutual inductance M, and FIG. 21 shows the time when I = 0.25A, as described above for the coil for obtaining such mutual inductance M. The control coil Nc for controlling magnetic flux is arrange | positioned, and Ic is applied to this control coil Nc. The solid lines of Figs. 20 and 21 are fused with R _L = 200 Hz, the dashed lines are fused with R _L = 375 Hz, and curves A and B are measured values Eo, and curves C and D are theoretical values Eos. Each one is composed.

를 각각 S_L,S_R로 하면 평균 경사로부터 구한 값은From these graphs, the output voltage for the change of the control current Ic in the region of the left L of the peak point of the anti-resonance curve, that is, the control current Ic is approximately 9.4 mA or more in FIG. 20 and Ic is approximately 4.2 mA or more in FIG. Is large and control sensitivity is good. Control sensitivity of left L and right R of the peak points of these graphs

Let S _L and S _R be the values of the average slope

S _L ≒ 33V / mA

S _R ≒ 22V / mA

It becomes Therefore, it is preferable to use the left L of the anti-resonance curve. If the resonance frequency at any M value is fo (M), the switching frequency fsw is

fo(M) …(18)fsw

fo (M)... (18)

It is preferable to set as follows.

In addition, in the above-described circuit analysis of FIG. 10 and below, N ₁ = N ₂ and L ₁ = L ₂ = M, but in general, N ₂ = nN ₁ , the results are almost the same. The output voltage Eos at this time

It becomes The mutual inductance M in this equation (19) is

Where L ₁ and L ₂ are the inductances of the coils N ₁ and N ₂ respectively, k is the coupling coefficient of the transformer 10, μo is the permeability of the vacuum μe is the specific permeability, A is the cross-sectional area of the core, Length. Also, the output voltage Eo _T due to the transformer coupling

This corresponds to the trans bond of formula (4) described above.

As described above, according to the embodiment of the present invention, when the resonant capacitor 20 is connected to the secondary shaft coil N ₂ of the transformer 10, the output voltage Eo is determined by the induced voltage Eo _T by the transformer coupling and the parallel resonance. Is taken as the sum of the resonance voltage Eos,

Eo = Eo _T + Eos... (21)

to be. In addition, the resonance voltage Eos is about 90% of the total output voltage Eo, and the remaining 10% is the output voltage by the transformer coupling. Next, the magnetic flux of the magnetic core of the transformer 10 can be controlled by changing the control current Ic, and the mutual inductance M at this time is also controlled at the same time. However, the relationship between the control current Ic and the mutual inductance M is nonlinear, and when the control current Ic increases, the mutual inductance M decreases, and when the left side of the anti-resonance curve described above is used, the output voltage also decreases. Next, in the case where the frequency is fixed, when the control current Ic is changed, the resonance voltage Eos changes along the anti-resonance curve, and together with this, the output voltage Eo _T by the transformer coupling changes along the upper part of the hysteresis curve. By inserting the resonant capacitor 20 in the secondary, the sensitivity is higher than that of simple transformer coupling.

That is, the induced voltage Eo _T by transfection's coupling is controlled according to the change of magnetic flux ø, as shown in equation (21), for example because it is Δø ₃ »Δø ₂ according to the hysteresis curve illustrated in Figure 22, It is shown almost linearly by the magnetic flux change of the magnetic flux in which the magnetic flux is reduced. Therefore, although the resonance voltage Eos is controlled by the change in the tangential slope of the hysteresis curve in FIG. 22, it is Δμe ₃ »Δμe _2, so that the change in permeability of the subjective magnetic force is added, that is, the slope of the nonlinear hysteresis phase of FIG. Controlled.

Here, for example, when the magnetic core is wettite, it is nonlinearly changed in the range of Δμe = 2500 to 200, and the non-rotation of the change is larger than that of the linearly varying flux, so that the output by trans coupling The output voltage Eos by resonance rather than the voltage Eo _T indicates that the control sensitivity is higher for the same control current.

Therefore, in the case where the output voltage is drawn out only by the induced voltage by the transformer coupling, when the resonant capacitor like the present invention is connected to the secondary shaft coil of the transformer, the control range for the change of the input voltage and the load is expanded. The effect that the power supply ripple suppression and the DC output resistance are improved is obtained. In addition, since the resonance voltage becomes an output voltage superimposed on the transformer coupling voltage, it is not necessary to increase the number of turns of the secondary shaft coil, and the output voltage increases with the same magnetic flux density, thereby miniaturizing the transformer. In the same manner, when a heavy transformer is used, electric power is supplied from the discharge energy of the capacitor 10 by parallel resonance, and the maximum supply power greatly increases. Since this can be done without increasing the excitation current or increasing the magnetic flux density, the efficiency can be increased and the power consumption can be reduced. In addition, since the output voltage waveform of the secondary shaft coil is substantially sinusoidal by resonance with the resonance capacitor, even if the excitation current of the primary coil is an asymmetrical AC power waveform such as a stationary wave current, the load current of the secondary rectifier circuit It is a rectified symmetrical waveform of positive symmetry, and the balance of the DC current circuit is maintained.

Next, in order to compare the present invention having such a feature with the conventional example, how the output characteristic varies depending on the presence or absence of the capacitor 20 for resonance in the configuration of FIG. Furthermore, in the case of the prior art example of FIG. 1, the capacitor 13 connected to the secondary coil N ₂ has a small capacity and has a different purpose from the present invention, and the operation mode of the circuit of FIG. 1 is almost similar to that of the capacitor 20 in FIG. If there is no.

First, as a conventional example, in the absence of the capacitor 20 of FIG. 7, the output waveforms of the respective parts as shown in FIG. 6 are obtained. In the case of obtaining the output voltage Eo of FIG. When the ferrite magnetic core is used as the perception of the transformer 10 and the input voltage is 100 V, the primary current in Fig. 6b is ± 2.5 A (5 A _PP ). The maximum magnetic flux density at this time is 1680 gauss, and the load characteristics are as shown in FIG. In FIG. 23, the solid line shows the output voltage, and the dotted line shows the ripple voltage, respectively. As can be seen from FIG. 23, when the output current I _L is in the range of about 0.16A to 0.4A, a constant voltage output of 115V is obtained, thus the minimum load is about 18.5W and the maximum load is 46W. The change range of the control current is in the range of 19.2 mA between 44.2 mA and 25 mA, and DELTA Ic = 19.2 mA / 46 W to 18.5 W. At this time, the conversion efficiency from AC on the primary side to DC on the secondary side is 46W / 60.5W = 76.1%.

Next, when the resonant capacitor 20 of the present invention is connected, the output waveforms of the respective parts shown in Figs. 24A to 24E are obtained. 24a shows the current waveform on the primary side and 24b shows the collector of the transistor (5). Rectifier a voltage V _CE between the emitter and the 24c to turn the current waveform of the transformer 2, the axle coil N ₂ of the terminal voltage, the 24d resonant capacitor 20 for the turning of the 10, and, the 24e turns 21 Each of the load current waveforms is shown. 25 shows load characteristics, the solid line shows the output voltage and the dotted line shows the voltage ripple, respectively. As described above, in the case of controlling the resonance voltage, when the AC input voltage is 100 V and the DC output voltage is 115 V, the primary side current in FIG. 24A is approximately 1.5 A, and 2.5 to 3.0 A _PP flows up and down. Since the output current I _L from which the constant voltage 115 V is obtained is OA to 1.7 A, the minimum load is OW at no load and the maximum load is about 80.5 W. The change range of the control current at this time is 36 mA to 25 mA, and ΔIc = 11 mA / 80.5 W to OW. AC to DC conversion efficiency is 80.5W / 97W = 83%.

As confirmed in the above experiments, the control sensitivity is improved, the maximum supply load is increased, and the conversion efficiency is improved.

Next, as another embodiment of the present invention, as shown in FIG. 26, the same effect as described above even when the resonant capacitor 30 is connected in series with N ₂ to the secondary coil of the transformer 10. Can be obtained. Since the other configuration of FIG. 26 is the same as that of FIG. 7, the same reference numerals are attached to the same parts, and description thereof is omitted.

The equivalent circuit of FIG. 26 is subjected to the equivalent conversion of the equivalent circuit of FIG. 10 described above to obtain an equivalent circuit of FIG. The symbol of each circuit element for the equivalent circuit of FIG. 27 is the same as that of FIG. 10, and the equivalent circuit of FIG. 28 can be obtained by substituting the input source ab terminal with the voltage source Ei for generating a sinusoidal wave emsinwt. The equivalent circuit of FIG. 29 is obtained by applying Ho-Febnan's theorem to the power supply side by looking at the power supply side at the terminal cd of 28 degrees. The voltage in FIG. 29 is Ei '

The inductance L summarized by the dashed line in FIG. 29 is

to be. If we add this inductance L and the inductance L ₂ -K of the 29th degree and set it to Lo

는Is displayed. FIG. 29 is a series resonance circuit of such an inductance Lo, a resistor R _L , and a resonance capacitor Cs.

Is

(=l/Z)는Become admittance

(= l / Z) is

It becomes Resonance angle frequency Wo and resonance frequency fo

이므로, 상기 (23)식에 의하여 입력의 실효값 전압 Ei'(rms)는to be. In this case, I = Ei

Since the effective value voltage Ei '(rms) of the input is

는The effective value of the current

Is

이며,It becomes Therefore, output voltage Eo is R _L

Is,

Is displayed.

Therefore, as in the case of parallel resonance, when the series resonance frequency fo is selected near the operating frequency and the M, L ₁ , L ₂ is changed with the control coil N ₂ , the output voltage is controlled. In addition, the same effects as in the case of the parallel resonance can be obtained, of course.

As can be seen from the above description, according to the power supply apparatus according to the present invention, a resonant capacitor is connected in parallel or in series with a secondary shaft coil, which is an output shaft of a transformer, and the resonance frequency is slightly higher than the operating frequency of the transformer. By controlling the mutual inductance value along the lower left curve with high frequency sensitivity of the resonance curve of the output voltage, the control sensitivity is improved, the constant voltage area is enlarged, and the power supply ripple is at the same time. This can suppress the reduction of the DC output resistance. In addition, since the maximum load supply capability of the transformer 10 is improved and the number of turns of the transformer 10 is not increased, the size and weight of the transformer can be reduced.

Claims (1)

The primary coil N ₁ and the secondary coil N ₂ are wound around a magnetic core having a nonlinear region, and at the same time, the magnetic flux in the magnetic core is not alternating with the primary coil N ₁ and the secondary coil N ₂ . A transformer 10 in which a control coil N ₂ to be controlled is wound; A switching circuit connected in series with one end of the primary coil N ₁ of the transformer 10, A direct current power source connected to the other end of the primary coil N ₁ of the transformer 10, An oscillation circuit for controlling the switching circuit; A rectifier circuit connected between the secondary coil N ₂ and an output terminal; In the power supply circuit having a control circuit for varying the control coil N ₂ in accordance with the voltage of the output terminal to maintain a substantially constant voltage of the output terminal, The secondary winding N _2, and connecting the capacitor 20 in parallel, and constitutes a resonant circuit as an inductance of the capacitor 20 and the secondary winding N _2, and the oscillation of the resonance frequency of this resonant circuit, the oscillator circuit Power supply, characterized in that set to a value slightly higher than the frequency.

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