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US20150280727A1 - Multiplying digital-to-analog converter and pipeline analog-to-digital converter using the same - Google Patents

Multiplying digital-to-analog converter and pipeline analog-to-digital converter using the same Download PDF

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US20150280727A1
US20150280727A1 US14/508,208 US201414508208A US2015280727A1 US 20150280727 A1 US20150280727 A1 US 20150280727A1 US 201414508208 A US201414508208 A US 201414508208A US 2015280727 A1 US2015280727 A1 US 2015280727A1
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analog
capacitive load
load cells
digital
switched
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US9160360B1 (en
Inventor
Yuan-Ching Lien
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MediaTek Inc
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MediaTek Inc
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Priority to EP15152823.9A priority patent/EP2924880A1/en
Priority to CN201510130508.8A priority patent/CN104954019B/en
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    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03M—CODING; DECODING; CODE CONVERSION IN GENERAL
    • H03M1/00—Analogue/digital conversion; Digital/analogue conversion
    • H03M1/12—Analogue/digital converters
    • H03M1/1205—Multiplexed conversion systems
    • H03M1/121—Interleaved, i.e. using multiple converters or converter parts for one channel
    • H03M1/1215—Interleaved, i.e. using multiple converters or converter parts for one channel using time-division multiplexing
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03M—CODING; DECODING; CODE CONVERSION IN GENERAL
    • H03M1/00—Analogue/digital conversion; Digital/analogue conversion
    • H03M1/12—Analogue/digital converters
    • H03M1/34—Analogue value compared with reference values
    • H03M1/38—Analogue value compared with reference values sequentially only, e.g. successive approximation type
    • H03M1/44—Sequential comparisons in series-connected stages with change in value of analogue signal
    • H03M1/442—Sequential comparisons in series-connected stages with change in value of analogue signal using switched capacitors
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03M—CODING; DECODING; CODE CONVERSION IN GENERAL
    • H03M1/00—Analogue/digital conversion; Digital/analogue conversion
    • H03M1/06—Continuously compensating for, or preventing, undesired influence of physical parameters
    • H03M1/0617—Continuously compensating for, or preventing, undesired influence of physical parameters characterised by the use of methods or means not specific to a particular type of detrimental influence
    • H03M1/0675—Continuously compensating for, or preventing, undesired influence of physical parameters characterised by the use of methods or means not specific to a particular type of detrimental influence using redundancy
    • H03M1/0697—Continuously compensating for, or preventing, undesired influence of physical parameters characterised by the use of methods or means not specific to a particular type of detrimental influence using redundancy in time, e.g. using additional comparison cycles
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03M—CODING; DECODING; CODE CONVERSION IN GENERAL
    • H03M1/00—Analogue/digital conversion; Digital/analogue conversion
    • H03M1/12—Analogue/digital converters
    • H03M1/14—Conversion in steps with each step involving the same or a different conversion means and delivering more than one bit
    • H03M1/16—Conversion in steps with each step involving the same or a different conversion means and delivering more than one bit with scale factor modification, i.e. by changing the amplification between the steps
    • H03M1/164—Conversion in steps with each step involving the same or a different conversion means and delivering more than one bit with scale factor modification, i.e. by changing the amplification between the steps the steps being performed sequentially in series-connected stages
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03M—CODING; DECODING; CODE CONVERSION IN GENERAL
    • H03M1/00—Analogue/digital conversion; Digital/analogue conversion
    • H03M1/12—Analogue/digital converters
    • H03M1/34—Analogue value compared with reference values
    • H03M1/38—Analogue value compared with reference values sequentially only, e.g. successive approximation type
    • H03M1/46—Analogue value compared with reference values sequentially only, e.g. successive approximation type with digital/analogue converter for supplying reference values to converter
    • H03M1/466—Analogue value compared with reference values sequentially only, e.g. successive approximation type with digital/analogue converter for supplying reference values to converter using switched capacitors
    • H03M1/468—Analogue value compared with reference values sequentially only, e.g. successive approximation type with digital/analogue converter for supplying reference values to converter using switched capacitors in which the input S/H circuit is merged with the feedback DAC array
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03M—CODING; DECODING; CODE CONVERSION IN GENERAL
    • H03M1/00—Analogue/digital conversion; Digital/analogue conversion
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03M—CODING; DECODING; CODE CONVERSION IN GENERAL
    • H03M1/00—Analogue/digital conversion; Digital/analogue conversion
    • H03M1/12—Analogue/digital converters

Definitions

  • the present invention relates to a multiplying digital-to-analog converter (MDAC) with capacitive load reset on an operational amplifier and a pipeline analog-to-digital converter (pipeline ADC) using the MDAC.
  • MDAC digital-to-analog converter
  • pipeline ADC pipeline analog-to-digital converter
  • Pipeline ADC has become a popular ADC architecture, which is used in a wide range of applications, including CCD imaging, ultrasonic medical imaging, digital receivers, base stations, digital video (for example, HDTV), xDSL, cable modems, fast Ethernet, and so on.
  • a pipeline ADC generally includes operational amplifiers (op amps) and the settling time of the pipeline ADC depends on the design of the op amps.
  • op amps operational amplifiers
  • a pipeline ADC with a fast settling time is called for.
  • a multiplying digital-to-analog converter (MDAC) and a pipeline analog-to-digital converter (pipeline ADC) using the MDAC are disclosed.
  • An MDAC in accordance with an exemplary embodiment of the disclosure includes an operational amplifier, a first switched-capacitor network and a second switched capacitor network.
  • the MDAC generates an analog output in accordance with digital input bits.
  • the first switched-capacitor network and a second switched capacitor network share the operational amplifier.
  • the first switched-capacitor network samples an analog signal when the first switched-capacitor network is disconnected from the operational amplifier.
  • the first switched-capacitor network is coupled to the operational amplifier for sampled signal amplification based on the digital input bits when the second switched-capacitor network is disconnected from the operational amplifier.
  • the second switched-capacitor network samples the analog signal when the second switched-capacitor network is disconnected from the operational amplifier.
  • the second switched-capacitor network is coupled to the operational amplifier for sampled signal amplification based on the digital input bits when the first switched-capacitor network is disconnected from the operational amplifier.
  • the operational amplifier is further coupled with first capacitive load cells when the first switched-capacitor network is coupled to the operational amplifier, and the first capacitive load cells are reset when the first switched-capacitor network is disconnected from the operational amplifier.
  • the operational amplifier is further coupled with second capacitive load cells when the second switched-capacitor network is coupled to the operational amplifier, and the second capacitive load cells are reset when the second switched-capacitor network is disconnected from the operational amplifier.
  • a pipeline ADC is shown.
  • the pipeline ADC further includes first and second analog-to-digital converters (ADCs) and a processing unit.
  • the processing unit is for time alignment and digital error correction.
  • the MDAC is coupled to an analog input of the pipeline ADC to receive the analog signal that is sampled by the first or second switched-capacitor networks.
  • the analog signal is further provided to the first ADC to generate the digital input bits for the MDAC.
  • the analog output of the MDAC is coupled to the second ADC and the second ADC outputs a digital representation of a signal received by the second ADC.
  • the processing unit receives the digital input bits generated by the first ADC and the digital representation generated by the second ADC for generation of a digital output of the pipeline ADC.
  • FIG. 1 illustrates a pipeline ADC 100 using an MDAC 102 in accordance with an exemplary embodiment of the disclosure
  • FIG. 2 illustrates the MDAC 102 in detail in accordance with an exemplary embodiment of the disclosure
  • FIG. 3 shows waveforms for operating the opamp-sharing architecture of FIG. 2 ;
  • FIG. 4 depicts an adjustment technique for the reset duration of the sampling capacitor cells Csar 1 of SAR 1 (or Csar 2 of SAR 2 ) in accordance with an exemplary embodiment of the disclosure.
  • FIG. 1 illustrates a pipeline ADC 100 using an MDAC 102 in accordance with an exemplary embodiment of the disclosure.
  • the pipeline ADC 100 includes N stages numbered from Stage 1 to StageN and a processing unit 104 .
  • the last stage StageN is an analog-to-digital converter (ADC), e.g. a direct conversion ADC or a successive approximation register ADC (abbreviated to SAR ADC).
  • ADC analog-to-digital converter
  • ADC analog-to-digital converter
  • SAR ADC successive approximation register ADC
  • an analog input of the pipeline ADC 100 is coupled to the circuit 106 in the first stage Stage 1 as an analog signal Vi.
  • the ADC 108 performs an analog-to-digital conversion on the analog signal Vi and generates digital input bits 110 (e.g. K-bits) for the MDAC 102 .
  • the MDAC 102 receives the digital input bits 110 and the analog signal Vi and generates an analog output Vo for the next stage Stage 2 .
  • the digital input bits 110 are further received by the processing unit 104 . Similar to the circuit 106 , stages Stage 2 to Stage(N ⁇ 1 ) are provided and cascaded after the first stage Stage 1 , each providing K-bits to the processing unit 104 .
  • the last stage StageN an ADC, converts the analog output of the previous stage Stage(N ⁇ 1 ) to a digital representation 112 (e.g. J-bits).
  • the processing unit 104 for time alignment and error correction generates the digital output of the pipeline ADC 100 .
  • the MDAC 102 is in an opamp-sharing architecture. There is one operational amplifier in the MDAC 102 and the operational amplifier is shared by several signal processing channels. When one channel is using the operational amplifier, a capacitive-load reset operation is performed for another channel that is disconnected from the operation amplifier. In this manner, the settling time of the pipeline ADC 100 is effectively reduced without sacrificing the normal operation phases of the pipeline ADC 100 .
  • FIG. 2 illustrates the MDAC 102 in detail in accordance with an exemplary embodiment of the disclosure, which is in the stage followed by an ADC 202 (e.g. the ADC in StageN of FIG. 1 ).
  • the MDAC 102 includes an operational amplifier OP and switched-capacitor networks DAC 1 and DAC 2 sharing the operational amplifier OP.
  • the switched-capacitor networks DAC 1 and DAC 2 correspond to two different signal processing channels, respectively.
  • the ADC 202 may be implemented by two time-interleaved SAR ADCs SAR 1 and SAR 2 corresponding to the two channels, respectively.
  • channel 1 The signal processing of channel 1 is discussed in this paragraph.
  • a sampling phase for channel 1 control signal ⁇ 1 off
  • the switched-capacitor network DAC 1 is disconnected from the operational amplifier OP and the analog signal Vi is sampled by the capacitors CS and Cf of the switched-capacitor network DAC 1 .
  • the switched-capacitor network DAC 1 is coupled to the operational amplifier OP for sampled signal amplification based on the digital input bits 110 provided from the ADC 108 .
  • the feedback capacitor Cf of DAC 1 is coupled between the input and output terminals of the operational amplifier OP and the remaining capacitors Cs of DAC 1 are controlled by the digital input bits 110 , respectively, to connect the input terminal of the operational amplifier OP to ground or a reference voltage source Vr.
  • the switched-capacitor network DAC 1 is coupled to the operation amplifier OP, the analog output Vo of the MDAC 102 is sampled into the sampling capacitor cells of the SAR ADC SAR 1 .
  • the SAR ADC SAR 1 determines a digital representation of the sampled analog output in successive approximations.
  • channel 2 The signal processing of channel 2 is discussed in this paragraph.
  • a sampling phase for channel 2 control signal ⁇ 2 off
  • the switched-capacitor network DAC 2 is disconnected from the operational amplifier OP and the analog signal Vi is sampled by the capacitors CS and Cf of the switched-capacitor network DAC 2 .
  • the switched-capacitor network DAC 2 is coupled to the operational amplifier OP for sampled signal amplification based on the digital input bits 110 provided from the ADC 108 .
  • the feedback capacitor Cf of DAC 2 is coupled between the input and output terminals of the operational amplifier OP and the remaining capacitors Cs of DAC 2 are controlled by the digital input bits 110 , respectively, to connect the input terminal of the operational amplifier OP to ground or a reference voltage source Vr.
  • the switched-capacitor network DAC 2 is coupled to the operation amplifier OP, the analog output Vo of the MDAC 102 is sampled into the sampling capacitor cells of the SAR ADC SAR 2 .
  • the SAR ADC SAR 2 determines a digital representation of the sampled analog output in successive approximations.
  • the operational amplifier OP is shared by the two signal processing channels.
  • the control signals ⁇ 1 and ⁇ 2 are in opposite phases.
  • the switched-capacitor network DAC 1 is disconnected from the operational amplifier OP for the sampling phase (control signal ⁇ 1 off) when the switched-capacitor network DAC 2 is coupled to the operational amplifier OP for the amplification phase (control signal ⁇ 2 on).
  • the switched-capacitor network DAC 1 is coupled to the operational amplifier OP for the amplification phase (control signal ⁇ 1 on) when the switched-capacitor network DAC 2 is disconnected from the operational amplifier OP for the sampling phase (control signal ⁇ 2 off).
  • a proposed architecture 204 of the operational amplifier OP and the sampling capacitor cells Csar 1 of the SAR ADC (SAR 1 ) and the sampling capacitor cells Csar 2 of the SAR ADC (SAR 2 ) are also shown in FIG. 2 .
  • the operational amplifier OP amplifies an input signal OPin by two gain stages gm 1 and gm 2 and generates an output signal OPout as the analog output Vo of the MDAC 102 .
  • compensation capacitor cells Cc 1 for loop compensation of the signal processing loop formed by the switched-capacitor network DAC 1 and the operational amplifier OP and compensation capacitor cells Cc 2 for loop compensation of the signal processing loop formed by the switched-capacitor network DAC 2 and the operational amplifier OP are provided.
  • the sampling capacitor cells Csar 1 and/or the compensation capacitor cells Cc 1 are/is regarded as capacitive load cells of channel 1 .
  • the sampling capacitor cells Csar 2 and/or the compensation capacitor cells Cc 2 are/is regarded as capacitive load cells of channel 2 .
  • the compensation capacitor cells Cc 1 for channel 1 and the compensation capacitor cells Cc 2 for channel 2 are discussed in this paragraph and are operated according to the control signals (p 1 and (p 2 .
  • the compensation capacitor cells Cc 1 are coupled in parallel between the input and output terminals of the second gain stage gm 2 for loop compensation when operational amplifier OP is coupled to the switched-capacitor network DAC 1 for the amplification phase of channel 1 , and the compensation capacitor cells Cc 1 are reset when the operational amplifier OP is disconnected from the switched-capacitor network DAC 1 and removed from the signal processing path of channel 1 . In this manner, the compensation capacitor cells Cc 1 are reset without sacrificing the normal operation phases of channel 1 .
  • the compensation capacitor cells Cc 2 are coupled in parallel between the input and output terminals of the second gain stage gm 2 for loop compensation when operational amplifier OP is coupled to the switched-capacitor network DAC 2 for the amplification phase of channel 2 , and the compensation capacitor cells Cc 2 are reset when the operational amplifier OP is disconnected from the switched-capacitor network DAC 2 and removed from the signal processing path of channel 2 . In this manner, the compensation capacitor cells Cc 2 are reset without sacrificing the normal operation phases of channel 2 .
  • the aforementioned capacitive load reset operation results in a fast settling time of the entire pipeline ADC 100 of FIG. 1 .
  • sampling capacitor cells Csar 1 of SAR 1 and the sampling capacitor cells Csar 2 of SAR 2 are discussed in this paragraph and are operated according to the control signals ⁇ 1 , ⁇ 2 , ⁇ 1 R and ⁇ 2 R.
  • the sampling capacitor cells Csar 1 of SAR 1 are coupled to the operational amplifier OP to sample the analog output Vo of the MDAC 102 when the switched-capacitor network DAC 1 is coupled to the operational amplifier OP.
  • the sampling capacitor cells Csar 1 of SAR 1 are reset after the successive approximation cycles of the SAR ADC SAR 1 and, at that moment, the operational amplifier OP is disconnected from the switched-capacitor network DAC 1 and the sampling capacitor cells Csar 1 of SAR 1 are disconnected from the operation amplifier OP. In this manner, the sampling capacitor cells Csar 1 are reset without sacrificing the normal operation phases of the SAR ADC SAR 1 .
  • the sampling capacitor cells Csar 2 of SAR 2 are coupled to the operational amplifier OP to sample the analog output Vo of the MDAC 102 when the switched-capacitor network DAC 2 is coupled to the operational amplifier OP.
  • the sampling capacitor cells Csar 2 of SAR 2 are reset after the successive approximation cycles of the SAR ADC SAR 2 and, at that moment, the operational amplifier OP is disconnected from the switched-capacitor network DAC 2 and the sampling capacitor cells Csar 2 of SAR 2 are disconnected from the operation amplifier OP. In this manner, the sampling capacitor cells Csar 2 are reset without sacrificing the normal operation phases of the SAR ADC SAR 2 .
  • the aforementioned capacitive load reset operation results in a fast settling time of the entire pipeline ADC 100 of FIG. 1 .
  • the capacitive load cells (e.g., Cc 1 , Cc 2 , Csar 1 or Csar 2 ) are reset by connecting all top plates and bottom plates to a common mode voltage level (e.g., a ground level).
  • the MDAC 102 is in a differential architecture and the capacitive load cells (e.g., Cc 1 , Cc 2 , Csar 1 or Csar 2 ) may be reset by connecting the positive capacitor cells (provided from the capacitive load cells for the positive signal path) with the negative capacitor cells (provided from the capacitive load cells for the negative signal path) for charge neutralization.
  • switched-capacitor network architecture (containing capacitors Cs and Cf and operated by a reference voltage source Vr) shown in FIG. 2 is not intended to limit the structure of the switched-capacitor networks.
  • the switched-capacitor network architecture shown in FIG. 2 may be replaced by another switched-capacitor network circuit.
  • FIG. 3 shows waveforms for operating the op-sharing architecture of FIG. 2 .
  • the compensation capacitor cells Cc 1 and the sampling capacitor cells Csar 1 specially designed for channel 1 are reset when the operational amplifier OP is switched to the signal amplification of channel 2 , without sacrificing the normal operation phases of channel 1 .
  • the compensation capacitor cells Cc 2 and the sampling capacitor cells Csar 2 specially designed for channel 2 are reset when the operational amplifier OP is switched to the signal amplification of channel 1 , without sacrificing the normal operation phases of channel 2 .
  • sampling capacitor cells Csar 1 are reset after the successive approximations provided for the analog output Vo sampled in the sampling capacitor cells Csar 1
  • sampling capacitor cells Csar 2 are reset after the successive approximations provided for the analog output Vo sampled in the sampling capacitor cells Csar 2 .
  • FIG. 4 depicts an adjustment technique for the reset duration of the sampling capacitor cells Csar 1 of SAR 1 and Csar 2 of SAR 2 in accordance with an exemplary embodiment of the disclosure.
  • Signal CK is a clock signal.
  • the signal CK is the asynchronous ADC clock generated by the asynchronous SAR ADC itself.
  • Each SAR ADC conversion is provided with a fixed number of clock cycles.
  • the reset duration for sampling capacitor cells of an SAR ADC depends on the metastability of successive approximations provided for a signal sampled in the sampling capacitor cells.
  • waveform 402 when a digital representation of a signal sampled in the sampling capacitor cells is obtained during a proper number of successive approximation cycles, no metastability error occurs and therefore the sampling capacitor cells are reset for a pre-determined reset duration T.
  • waveform 404 when a metastability error occurs to indicate that more conversion time are required for a fixed number of clock cycles, the reset duration for the sampling capacitor cells are reduced to T′ or even zero.

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Abstract

A multiplying digital-to-analog converter (MDAC) with capacitive load reset on an operational amplifier and a pipeline analog-to-digital converter using the MDAC are disclosed. The MDAC includes an operational amplifier and first and second switched-capacitor networks sharing the operational amplifier. The operational amplifier is further coupled with first capacitive load cells when the first switched-capacitor network is coupled to the operational amplifier, and the first capacitive load cells are reset when the first switched-capacitor network is disconnected from the operational amplifier. The operational amplifier is further coupled with second capacitive load cells when the second switched-capacitor network is coupled to the operational amplifier, and the second capacitive load cells are reset when the second switched-capacitor network is disconnected from the operational amplifier.

Description

    CROSS REFERENCE TO RELATED APPLICATIONS
  • This application claims the benefit of U.S. Provisional Application No. 61/971,043 filed Mar. 27, 2014, the entirety of which is incorporated by reference herein.
    • BACKGROUND OF THE INVENTION
  • 1. Field of the Invention
  • The present invention relates to a multiplying digital-to-analog converter (MDAC) with capacitive load reset on an operational amplifier and a pipeline analog-to-digital converter (pipeline ADC) using the MDAC.
  • 2. Description of the Related Art
  • Pipeline ADC has become a popular ADC architecture, which is used in a wide range of applications, including CCD imaging, ultrasonic medical imaging, digital receivers, base stations, digital video (for example, HDTV), xDSL, cable modems, fast Ethernet, and so on.
  • A pipeline ADC generally includes operational amplifiers (op amps) and the settling time of the pipeline ADC depends on the design of the op amps. A pipeline ADC with a fast settling time is called for.
    • BRIEF SUMMARY OF THE INVENTION
  • A multiplying digital-to-analog converter (MDAC) and a pipeline analog-to-digital converter (pipeline ADC) using the MDAC are disclosed.
  • An MDAC in accordance with an exemplary embodiment of the disclosure includes an operational amplifier, a first switched-capacitor network and a second switched capacitor network. The MDAC generates an analog output in accordance with digital input bits. The first switched-capacitor network and a second switched capacitor network share the operational amplifier. The first switched-capacitor network samples an analog signal when the first switched-capacitor network is disconnected from the operational amplifier. The first switched-capacitor network is coupled to the operational amplifier for sampled signal amplification based on the digital input bits when the second switched-capacitor network is disconnected from the operational amplifier. The second switched-capacitor network samples the analog signal when the second switched-capacitor network is disconnected from the operational amplifier. The second switched-capacitor network is coupled to the operational amplifier for sampled signal amplification based on the digital input bits when the first switched-capacitor network is disconnected from the operational amplifier. The operational amplifier is further coupled with first capacitive load cells when the first switched-capacitor network is coupled to the operational amplifier, and the first capacitive load cells are reset when the first switched-capacitor network is disconnected from the operational amplifier. The operational amplifier is further coupled with second capacitive load cells when the second switched-capacitor network is coupled to the operational amplifier, and the second capacitive load cells are reset when the second switched-capacitor network is disconnected from the operational amplifier.
  • In another exemplary embodiment, a pipeline ADC is shown. In addition to the aforementioned MDAC, the pipeline ADC further includes first and second analog-to-digital converters (ADCs) and a processing unit. The processing unit is for time alignment and digital error correction. The MDAC is coupled to an analog input of the pipeline ADC to receive the analog signal that is sampled by the first or second switched-capacitor networks. Furthermore, the analog signal is further provided to the first ADC to generate the digital input bits for the MDAC. The analog output of the MDAC is coupled to the second ADC and the second ADC outputs a digital representation of a signal received by the second ADC. The processing unit receives the digital input bits generated by the first ADC and the digital representation generated by the second ADC for generation of a digital output of the pipeline ADC.
  • A detailed description is given in the following embodiments with reference to the accompanying drawings.
    • BRIEF DESCRIPTION OF THE DRAWINGS
  • The present invention can be more fully understood by reading the subsequent detailed description and examples with references made to the accompanying drawings, wherein:
  • FIG. 1 illustrates a pipeline ADC 100 using an MDAC 102 in accordance with an exemplary embodiment of the disclosure;
  • FIG. 2 illustrates the MDAC 102 in detail in accordance with an exemplary embodiment of the disclosure;
  • FIG. 3 shows waveforms for operating the opamp-sharing architecture of FIG. 2; and
  • FIG. 4 depicts an adjustment technique for the reset duration of the sampling capacitor cells Csar1 of SAR1 (or Csar2 of SAR2) in accordance with an exemplary embodiment of the disclosure.
    •  DETAILED DESCRIPTION OF THE INVENTION
  • The following description is of the best-contemplated mode of carrying out the invention. This description is made for the purpose of illustrating the general principles of the invention and should not be taken in a limiting sense. The scope of the invention is best determined by reference to the appended claims.
  • FIG. 1 illustrates a pipeline ADC 100 using an MDAC 102 in accordance with an exemplary embodiment of the disclosure. The pipeline ADC 100 includes N stages numbered from Stage1 to StageN and a processing unit 104. The last stage StageN is an analog-to-digital converter (ADC), e.g. a direct conversion ADC or a successive approximation register ADC (abbreviated to SAR ADC). A proposed circuit, e.g. circuit 106, for any of the stages Stage1 to Stage(N-1) is shown and discussed in the following paragraphs, which includes an ADC 108 as well as the MDAC 102.
  • As shown, an analog input of the pipeline ADC 100 is coupled to the circuit 106 in the first stage Stage1 as an analog signal Vi. The ADC 108 performs an analog-to-digital conversion on the analog signal Vi and generates digital input bits 110 (e.g. K-bits) for the MDAC 102. The MDAC 102 receives the digital input bits 110 and the analog signal Vi and generates an analog output Vo for the next stage Stage2. The digital input bits 110 are further received by the processing unit 104. Similar to the circuit 106, stages Stage2 to Stage(N−1) are provided and cascaded after the first stage Stage1, each providing K-bits to the processing unit 104. The last stage StageN, an ADC, converts the analog output of the previous stage Stage(N−1) to a digital representation 112 (e.g. J-bits). In accordance with the digital bits received from the stages Stage1 to StageN, the processing unit 104 for time alignment and error correction generates the digital output of the pipeline ADC 100.
  • Note that the MDAC 102 is in an opamp-sharing architecture. There is one operational amplifier in the MDAC 102 and the operational amplifier is shared by several signal processing channels. When one channel is using the operational amplifier, a capacitive-load reset operation is performed for another channel that is disconnected from the operation amplifier. In this manner, the settling time of the pipeline ADC 100 is effectively reduced without sacrificing the normal operation phases of the pipeline ADC 100.
  • FIG. 2 illustrates the MDAC 102 in detail in accordance with an exemplary embodiment of the disclosure, which is in the stage followed by an ADC 202 (e.g. the ADC in StageN of FIG. 1). As shown in FIG. 2, the MDAC 102 includes an operational amplifier OP and switched-capacitor networks DAC1 and DAC2 sharing the operational amplifier OP. The switched-capacitor networks DAC1 and DAC2 correspond to two different signal processing channels, respectively. The ADC 202 may be implemented by two time-interleaved SAR ADCs SAR1 and SAR2 corresponding to the two channels, respectively.
  • The signal processing of channel 1 is discussed in this paragraph. In a sampling phase for channel 1 (control signal φ1 off), the switched-capacitor network DAC1 is disconnected from the operational amplifier OP and the analog signal Vi is sampled by the capacitors CS and Cf of the switched-capacitor network DAC1. In an amplification phase for channel 1 (control signal φ1 on), the switched-capacitor network DAC1 is coupled to the operational amplifier OP for sampled signal amplification based on the digital input bits 110 provided from the ADC 108. For signal amplification, the feedback capacitor Cf of DAC1 is coupled between the input and output terminals of the operational amplifier OP and the remaining capacitors Cs of DAC1 are controlled by the digital input bits 110, respectively, to connect the input terminal of the operational amplifier OP to ground or a reference voltage source Vr. When the switched-capacitor network DAC1 is coupled to the operation amplifier OP, the analog output Vo of the MDAC 102 is sampled into the sampling capacitor cells of the SAR ADC SAR1. When the switched-capacitor network DAC1 is disconnected from the operation amplifier OP, the SAR ADC SAR1 determines a digital representation of the sampled analog output in successive approximations.
  • The signal processing of channel 2 is discussed in this paragraph. In a sampling phase for channel 2 (control signal φ2 off), the switched-capacitor network DAC2 is disconnected from the operational amplifier OP and the analog signal Vi is sampled by the capacitors CS and Cf of the switched-capacitor network DAC2. In an amplification phase for channel 2 (control signal φ2 on), the switched-capacitor network DAC2 is coupled to the operational amplifier OP for sampled signal amplification based on the digital input bits 110 provided from the ADC 108. For signal amplification, the feedback capacitor Cf of DAC2 is coupled between the input and output terminals of the operational amplifier OP and the remaining capacitors Cs of DAC2 are controlled by the digital input bits 110, respectively, to connect the input terminal of the operational amplifier OP to ground or a reference voltage source Vr. When the switched-capacitor network DAC2 is coupled to the operation amplifier OP, the analog output Vo of the MDAC 102 is sampled into the sampling capacitor cells of the SAR ADC SAR2. When the switched-capacitor network DAC2 is disconnected from the operation amplifier OP, the SAR ADC SAR2 determines a digital representation of the sampled analog output in successive approximations.
  • As described above, the operational amplifier OP is shared by the two signal processing channels. In an exemplary embodiment, the control signals φ1 and φ2 are in opposite phases. The switched-capacitor network DAC1 is disconnected from the operational amplifier OP for the sampling phase (control signal φ1 off) when the switched-capacitor network DAC2 is coupled to the operational amplifier OP for the amplification phase (control signal φ2 on). The switched-capacitor network DAC1 is coupled to the operational amplifier OP for the amplification phase (control signal φ1 on) when the switched-capacitor network DAC2 is disconnected from the operational amplifier OP for the sampling phase (control signal φ2 off).
  • A proposed architecture 204 of the operational amplifier OP and the sampling capacitor cells Csar1 of the SAR ADC (SAR1) and the sampling capacitor cells Csar2 of the SAR ADC (SAR2) are also shown in FIG. 2. Referring to the proposed architecture 204, the operational amplifier OP amplifies an input signal OPin by two gain stages gm1 and gm2 and generates an output signal OPout as the analog output Vo of the MDAC 102. Furthermore, compensation capacitor cells Cc1 for loop compensation of the signal processing loop formed by the switched-capacitor network DAC1 and the operational amplifier OP and compensation capacitor cells Cc2 for loop compensation of the signal processing loop formed by the switched-capacitor network DAC2 and the operational amplifier OP are provided. The sampling capacitor cells Csar1 and/or the compensation capacitor cells Cc1 are/is regarded as capacitive load cells of channel 1. The sampling capacitor cells Csar2 and/or the compensation capacitor cells Cc2 are/is regarded as capacitive load cells of channel 2.
  • The compensation capacitor cells Cc1 for channel 1 and the compensation capacitor cells Cc2 for channel 2 are discussed in this paragraph and are operated according to the control signals (p1 and (p2. As shown, the compensation capacitor cells Cc1 are coupled in parallel between the input and output terminals of the second gain stage gm2 for loop compensation when operational amplifier OP is coupled to the switched-capacitor network DAC1 for the amplification phase of channel 1, and the compensation capacitor cells Cc1 are reset when the operational amplifier OP is disconnected from the switched-capacitor network DAC1 and removed from the signal processing path of channel 1. In this manner, the compensation capacitor cells Cc1 are reset without sacrificing the normal operation phases of channel 1. Furthermore, the compensation capacitor cells Cc2 are coupled in parallel between the input and output terminals of the second gain stage gm2 for loop compensation when operational amplifier OP is coupled to the switched-capacitor network DAC2 for the amplification phase of channel 2, and the compensation capacitor cells Cc2 are reset when the operational amplifier OP is disconnected from the switched-capacitor network DAC2 and removed from the signal processing path of channel 2. In this manner, the compensation capacitor cells Cc2 are reset without sacrificing the normal operation phases of channel 2. The aforementioned capacitive load reset operation results in a fast settling time of the entire pipeline ADC 100 of FIG. 1.
  • The sampling capacitor cells Csar1 of SAR1 and the sampling capacitor cells Csar2 of SAR2 are discussed in this paragraph and are operated according to the control signals φ1, φ2, φ1R and φ2R. The sampling capacitor cells Csar1 of SAR1 are coupled to the operational amplifier OP to sample the analog output Vo of the MDAC 102 when the switched-capacitor network DAC1 is coupled to the operational amplifier OP. According to the control signal φ2R, the sampling capacitor cells Csar1 of SAR1 are reset after the successive approximation cycles of the SAR ADC SAR1 and, at that moment, the operational amplifier OP is disconnected from the switched-capacitor network DAC1 and the sampling capacitor cells Csar1 of SAR1 are disconnected from the operation amplifier OP. In this manner, the sampling capacitor cells Csar1 are reset without sacrificing the normal operation phases of the SAR ADC SAR1. The sampling capacitor cells Csar2 of SAR2 are coupled to the operational amplifier OP to sample the analog output Vo of the MDAC 102 when the switched-capacitor network DAC2 is coupled to the operational amplifier OP. According to the control signal φ1R, the sampling capacitor cells Csar2 of SAR2 are reset after the successive approximation cycles of the SAR ADC SAR2 and, at that moment, the operational amplifier OP is disconnected from the switched-capacitor network DAC2 and the sampling capacitor cells Csar2 of SAR2 are disconnected from the operation amplifier OP. In this manner, the sampling capacitor cells Csar2 are reset without sacrificing the normal operation phases of the SAR ADC SAR2. The aforementioned capacitive load reset operation results in a fast settling time of the entire pipeline ADC 100 of FIG. 1.
  • In FIG. 2, the capacitive load cells (e.g., Cc1, Cc2, Csar1 or Csar2) are reset by connecting all top plates and bottom plates to a common mode voltage level (e.g., a ground level). In another exemplary embodiment, the MDAC 102 is in a differential architecture and the capacitive load cells (e.g., Cc1, Cc2, Csar1 or Csar2) may be reset by connecting the positive capacitor cells (provided from the capacitive load cells for the positive signal path) with the negative capacitor cells (provided from the capacitive load cells for the negative signal path) for charge neutralization.
  • Note that the switched-capacitor network architecture (containing capacitors Cs and Cf and operated by a reference voltage source Vr) shown in FIG. 2 is not intended to limit the structure of the switched-capacitor networks. The switched-capacitor network architecture shown in FIG. 2 may be replaced by another switched-capacitor network circuit.
  • Furthermore, it is not intended to limit the operational amplifier OP to the 2-stage architecture 204 and any capacitive device coupled to the operational amplifier OP as a capacitive loading may be reset by the aforementioned technique.
  • FIG. 3 shows waveforms for operating the op-sharing architecture of FIG. 2. The compensation capacitor cells Cc1 and the sampling capacitor cells Csar1 specially designed for channel 1 are reset when the operational amplifier OP is switched to the signal amplification of channel 2, without sacrificing the normal operation phases of channel 1. The compensation capacitor cells Cc2 and the sampling capacitor cells Csar2 specially designed for channel 2 are reset when the operational amplifier OP is switched to the signal amplification of channel 1, without sacrificing the normal operation phases of channel 2. Note that the sampling capacitor cells Csar1 are reset after the successive approximations provided for the analog output Vo sampled in the sampling capacitor cells Csar1, and the sampling capacitor cells Csar2 are reset after the successive approximations provided for the analog output Vo sampled in the sampling capacitor cells Csar2.
  • FIG. 4 depicts an adjustment technique for the reset duration of the sampling capacitor cells Csar1 of SAR1 and Csar2 of SAR2 in accordance with an exemplary embodiment of the disclosure. Signal CK is a clock signal. In a case wherein the SAR ADC SAR1/SAR2 is in an asynchronous architecture, the signal CK is the asynchronous ADC clock generated by the asynchronous SAR ADC itself. Each SAR ADC conversion is provided with a fixed number of clock cycles. The reset duration for sampling capacitor cells of an SAR ADC depends on the metastability of successive approximations provided for a signal sampled in the sampling capacitor cells. Referring to waveform 402, when a digital representation of a signal sampled in the sampling capacitor cells is obtained during a proper number of successive approximation cycles, no metastability error occurs and therefore the sampling capacitor cells are reset for a pre-determined reset duration T. Referring to waveform 404, when a metastability error occurs to indicate that more conversion time are required for a fixed number of clock cycles, the reset duration for the sampling capacitor cells are reduced to T′ or even zero.
  • While the invention has been described by way of example and in terms of the preferred embodiments, it is to be understood that the invention is not limited to the disclosed embodiments. On the contrary, it is intended to cover various modifications and similar arrangements (as would be apparent to those skilled in the art). Therefore, the scope of the appended claims should be accorded the broadest interpretation so as to encompass all such modifications and similar arrangements.

Claims (13)

What is claimed is:
1. A multiplying digital-to-analog converter, generating an analog output in accordance with digital input bits, comprising:
an operational amplifier; and
a first switched-capacitor network and a second switched capacitor network sharing the operational amplifier,
wherein:
the first switched-capacitor network samples an analog signal when the first switched-capacitor network is disconnected from the operational amplifier;
the first switched-capacitor network is coupled to the operational amplifier for sampled signal amplification based on the digital input bits when the second switched-capacitor network is disconnected from the operational amplifier;
the second switched-capacitor network samples the analog signal when the second switched-capacitor network is disconnected from the operational amplifier;
the second switched-capacitor network is coupled to the operational amplifier for sampled signal amplification based on the digital input bits when the first switched-capacitor network is disconnected from the operational amplifier;
the operational amplifier is further coupled with first capacitive load cells when the first switched-capacitor network is coupled to the operational amplifier, and the first capacitive load cells are reset when the first switched-capacitor network is disconnected from the operational
the operational amplifier is further coupled with second capacitive load cells when the second switched-capacitor network is coupled to the operational amplifier, and the second capacitive load cells are reset when the second switched-capacitor network is disconnected from the operational amplifier.
2. The multiplying digital-to-analog converter as claimed in claim 1, wherein:
the first capacitive load cells are provided for the first switched-capacitor network and the operational amplifier for loop compensation; and
the second capacitive load cells are provided for the second switched-capacitor network and the operational amplifier for loop compensation.
3. The multiplying digital-to-analog converter as claimed in claim 2, wherein:
the first capacitive load cells are reset by connecting top plates and bottom plates of the first capacitive load cells to a common mode voltage level; and
the second capacitive load cells are reset by connecting top plates and bottom plates of the second capacitive load cells to a common mode voltage level.
4. The multiplying digital-to-analog converter as claimed in claim 2, wherein:
the multiplying digital-to-analog converter is in a differential architecture;
the first capacitive load cells are reset by connecting positive capacitor cells between the first capacitive load cells with negative capacitor cells between the first capacitive load cells for charge neutralization; and
the second capacitive load cells are reset by connecting positive capacitor cells between the second capacitive load cells with negative capacitor cells between the second capacitive load cells for charge neutralization.
5. The multiplying digital-to-analog converter as claimed in claim 1, wherein:
the analog output of the multiplying digital-to-analog converter is further coupled to an analog-to-digital converter comprising a first successive approximation register analog-to-digital converter and a second successive approximation register analog-to-digital converter;
the first capacitive load cells are provided by the first successive approximation register analog-to-digital converter, which sample the analog output of the multiplying digital-to-analog converter when the first switched-capacitor network is coupled to the operational amplifier; and
the second capacitive load cells are provided by the second successive approximation register analog-to-digital converter, which sample the analog output of the multiplying digital-to-analog converter when the second switched-capacitor network is coupled to the operational amplifier.
6. The multiplying digital-to-analog converter as claimed in claim 5, wherein:
when the first switched-capacitor network is disconnected from the operational amplifier, the first successive approximation register analog-to-digital converter determines a digital representation of the analog output sampled in the first capacitive load cells in successive approximations; and
when the second switched-capacitor network is disconnected from the operational amplifier, the second successive approximation register analog-to-digital converter determines a digital representation of the analog output sampled in the second capacitive load cells in successive approximations.
7. The multiplying digital-to-analog converter as claimed in claim 6, wherein:
the first capacitive load cells are reset after successive approximations provided
the second capacitive load cells are reset after successive approximations provided for the analog output sampled in the second capacitive load cells.
8. The multiplying digital-to-analog converter as claimed in claim 7, wherein:
a reset duration for the first capacitive load cells depends on metastability of the successive approximations provided for the analog output sampled in the first capacitive load cells; and
a reset duration for the second capacitive load cells depends on metastability of the successive approximations provided for the analog output sampled in the second capacitive load cells.
9. The multiplying digital-to-analog converter as claimed in claim 8, wherein:
the reset duration for the first capacitive load cells is reduced when a metastability error occurs to indicate that more cycles are required for the successive approximations provided for the analog output sampled in the first capacitive load cells; and
the reset duration for the second capacitive load cells is reduced to zero when a metastability error occurs to indicate that more cycles are required for the successive approximations provided for the analog output sampled in the second capacitive load cells.
10. The multiplying digital-to-analog converter as claimed in claim 5, wherein:
the first capacitive load cells are reset by connecting top plates and bottom plates of the first capacitive load cells to a common mode voltage level; and
the second capacitive load cells are reset by connecting top plates and bottom plates of the second capacitive load cells to a common mode voltage level.
11. The multiplying digital-to-analog converter as claimed in claim 5, wherein:
the multiplying digital-to-analog converter is in a differential architecture;
the first capacitive load cells are reset by connecting positive capacitor cells between the first capacitive load cells with negative capacitor cells between the first capacitive load cells for charge neutralization; and
the second capacitive load cells are reset by connecting positive capacitor cells between the second capacitive load cells with negative capacitor cells between the second capacitive load cells for charge neutralization.
12. The multiplying digital-to-analog converter as claimed in claim 1, wherein:
the first switched-capacitor network samples the analog signal when the second switched-capacitor network is coupled to the operational amplifier for sampled signal amplification based on the digital input bits; and
the second switched-capacitor network samples the analog signal when the first switched-capacitor network is coupled to the operational amplifier for sampled signal amplification based on the digital input bits.
13. A pipeline analog-to-digital converter, comprising:
the multiplying digital-to-analog converter as claimed in claim 1 and coupled to an analog input of the pipeline analog-to-digital converter to receive the analog signal;
a first analog-to-digital converter that provides the multiplying digital-to-analog converter with the digital input bits based on the analog signal;
a second analog-to-digital converter, coupled to the analog output of the multiplying digital-to-analog converter for generation of a digital representation; and
a processing unit for time alignment and digital error correction, receiving the digital input bits generated by the first analog-to-digital converter and the for generation of a digital output of the pipeline analog-to-digital converter.
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* Cited by examiner, † Cited by third party
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US20150263757A1 (en) * 2014-03-11 2015-09-17 Texas Instruments Incorporated Digital to analog converter discharge circuit and associated method for analog to digital converter circuits
US9386240B1 (en) * 2015-03-12 2016-07-05 Omnivision Technologies, Inc. Compensation for dual conversion gain high dynamic range sensor
US20160241254A1 (en) * 2014-07-31 2016-08-18 Chung-Ang University Industry-Academy Cooperation Foundation Asynchronous successive approximation register analog-to-digital converter and internal clock generator included therein
US10749542B2 (en) * 2018-06-29 2020-08-18 Luxtera Llc. Method and system for an asynchronous successive approximation register analog-to-digital converter with word completion algorithm
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US11811419B2 (en) 2018-06-29 2023-11-07 Cisco Technology, Inc. Method and system for an asynchronous successive approximation register analog-to-digital converter with word completion algorithm

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* Cited by examiner, † Cited by third party
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US10382050B1 (en) * 2018-05-25 2019-08-13 Dialog Semiconductor (Uk) Limited Integrated multiplying successive approximation analog to digital converter
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US20250141461A1 (en) * 2023-10-27 2025-05-01 Avago Technologies International Sales Pte. Limited System and method for reconfigurable hybrid analog-to-digital converter (adc)

Citations (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US7304598B1 (en) * 2006-08-30 2007-12-04 Infineon Technologies Ag Shared amplifier circuit

Family Cites Families (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US7847720B2 (en) * 2009-01-16 2010-12-07 Mediatek Inc. Pipelined analog-to-digital converter
US7911370B2 (en) * 2009-06-25 2011-03-22 Mediatek Inc. Pipeline analog-to-digital converter with programmable gain function
US8471753B1 (en) * 2010-08-24 2013-06-25 Mediatek Inc. Pipelined analog-to-digital converter and method for converting analog signal to digital signal
TWI489785B (en) * 2011-08-31 2015-06-21 Pixart Imaging Inc Pipeline analog-to-digital converter and method thereof
US8659461B1 (en) 2012-11-13 2014-02-25 University Of Macau Analog to digital converter circuit

Patent Citations (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US7304598B1 (en) * 2006-08-30 2007-12-04 Infineon Technologies Ag Shared amplifier circuit

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US9294116B2 (en) * 2014-03-11 2016-03-22 Texas Instruments Incorporated Digital to analog converter discharge circuit and associated method for analog to digital converter circuits
US20160241254A1 (en) * 2014-07-31 2016-08-18 Chung-Ang University Industry-Academy Cooperation Foundation Asynchronous successive approximation register analog-to-digital converter and internal clock generator included therein
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US9386240B1 (en) * 2015-03-12 2016-07-05 Omnivision Technologies, Inc. Compensation for dual conversion gain high dynamic range sensor
US10749542B2 (en) * 2018-06-29 2020-08-18 Luxtera Llc. Method and system for an asynchronous successive approximation register analog-to-digital converter with word completion algorithm
US11290122B2 (en) 2018-06-29 2022-03-29 Luxtera Llc Method and system for an asynchronous successive approximation register analog-to-digital converter with word completion algorithm
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